COMMITTEE T1 – TELECOMMUNICATIONS
Working Group T1E1.4
November 28, 2000
T1E1.4/2000-002R6
CONTRIBUTION
TITLE:
Draft proposed American National Standard, Spectrum Management for Loop
Transmission Systems
SOURCE*: Editor
PROJECT: T1E1-38, Spectral Compatibility Aspects for Facilities between a Central Office and the
Network-to-Customer Interface (Twisted Pair Transmission System)
ABSTRACT
Attached is year 2000 R6 version of the draft standard for Spectrum Compatibility T1.417. It reflects
final draft status after changes resulting from agreements made during the November 13 – 16, 2000
meeting in Savannah, GA.
This is a draft document and thus, is dynamic in nature. It does not reflect a consensus of Committee T1Telecommunications and it may be changed or modified. Neither ATIS nor Committee T1 makes any representation
or warranty, express or implied, with respect to the sufficiency, accuracy or utility of the information or opinion
contained or reflected in the material utilized. ATIS and Committee T1 further expressly advise that any use of or
reliance upon the material in question is at your risk and neither ATIS nor Committee T1 shall be liable for any
damage or injury, of whatever nature, incurred by any person arising out of any utilization of the material. It is
possible that this material will at some future date be included in a copyrighted work by ATIS.
* CONTACT: Craig F. Valenti; email: [email protected]; Tel: 973-829-4203; Fax: 973-829-5962
COMMITTEE T1 – TELECOMMUNICATIONS
Working Group T1E1.4
November 28, 2000
This is a draft document and thus, is dynamic in nature.
T1E1.4/2000-002R6
T1E1.4/2000-002R6
ANSI T1.417
American National Standard
For Telecommunications
Spectrum Management
For Loop Transmission Systems
Secretariat
Alliance for Telecommunications Industry Solutions
Approved <Date to be determined>
American National Standards Institute, Inc.
Abstract
This standard provides spectrum management requirements and recommendations for the administration of
services and technologies that use metallic subscriber loop cables. Spectrum management is the administration
of the loop plant in a way that provides spectral compatibility for services and technologies that use pairs in the
same cable. In order to achieve spectral compatibility, the ingress energy that transfers into a loop pair, from
services and transmission system technologies on other pairs in the same cable, must not cause an unacceptable
degradation of performance. In addition, the egress energy from a particular loop pair must not transfer into
other pairs in a manner that causes an unacceptable degradation in the performance of services and technologies
on those pairs. This standard includes signal power limits and technology deployment guidelines for digital
subscriber line spectrum management classes. It also provides a generic analytical method to determine spectral
compatibility.
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
American
National
Standard
ANSI T1.417
Approval of an American National Standard requires verification by ANSI that the
requirements for due process, consensus, and other criteria for approval have been met by
the standards developer.
Consensus is established when, in the judgment of the ANSI Board of Standards Review,
substantial agreement has been reached by directly and materially affected interests.
Substantial agreement means much more than a simple majority, but not necessarily
unanimity. Consensus requires that all views and objections be considered, and that a
concerted effort be made toward their resolution.
The use of American National Standards is completely voluntary; their existence does not
in any respect preclude anyone, whether he has approved the standards or not, from
manufacturing, marketing, purchasing, or using products, processes, or procedures not
conforming to the standards.
The American National Standards Institute does not develop standards and will in no
circumstances give an interpretation of any American National Standard. Moreover, no
person shall have the right or authority to issue an interpretation of an American National
Standard in the name of the American National Standards Institute. Requests for
interpretations should be addressed to the secretariat or sponsor whose name appears on
the title page of this standard.
CAUTION NOTICE: This American National Standard may be revised or withdrawn at
any time. The procedures of the American National Standards Institute require that action
be taken periodically to reaffirm, revise, or withdraw this standard. Purchasers of
American National Standards may receive current information on all standards by calling
or writing the American National Standards Institute.
Published by
American National Standards Institute
11 West 42nd Street, New York, New York 10036
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ANSI T1.417
Table of Contents
Foreword ...................................................................................................................... xv
1.
Scope, purpose, and application....................................................................................2
Scope .............................................................................................................................2
Purpose ..........................................................................................................................3
Application......................................................................................................................3
2.
Normative references .....................................................................................................3
3.
Definitions, abbreviations, acronyms, and symbols .......................................................4
3.1
Definitions.......................................................................................................................4
3.2
Abbreviations, acronyms, and symbols..........................................................................5
4.
General Information........................................................................................................7
4.1
Crosstalk ........................................................................................................................7
4.2
Spectral compatibility .....................................................................................................8
4.3
Spectrum management ..................................................................................................8
4.3.1 Basis loop systems.........................................................................................................8
4.3.1.1 Voicegrade services................................................................................................9
4.3.1.2 Enhanced Business Services (P-Phone) ................................................................9
4.3.1.3 Digital Data Services (DDS)....................................................................................9
4.3.1.4 Basic Rate ISDN (BRI)..........................................................................................10
4.3.1.5 High-Bit-Rate Digital Subscriber Line (HDSL) ......................................................10
4.3.1.6 HDSL2...................................................................................................................10
4.3.1.7 ADSL, RADSL, and Splitterless ADSL..................................................................10
4.3.1.8 2B1Q SDSL...........................................................................................................10
4.3.2 Legacy systems............................................................................................................10
4.3.3 Signal power limitations (method A).............................................................................11
4.3.3.1 Transceiver unit – remote terminal end (TU-R) ....................................................11
4.3.3.2 Transceiver unit – central office end (TU-C) .........................................................11
4.3.4 Technology deployment guidelines..............................................................................11
4.3.4.1 Deployment guidelines..........................................................................................12
4.3.4.2 Binder Group Considerations................................................................................12
4.3.5 Analytical method of determining spectral compatibility (method B) ...........................12
4.3.5.1 Margin computations.............................................................................................12
4.3.5.2 Evaluation loops....................................................................................................13
5.
Signal power limits and other criteria ...........................................................................13
5.1
Short-term stationary systems .....................................................................................13
5.2
Spectral compatibility of systems using intermediate TU devices ...............................13
5.3
Spectrum management classes...................................................................................13
5.3.1 Spectrum management class 1....................................................................................13
5.3.1.1 Spectrum management class 1 PSD and total average power limitation.............13
5.3.1.2 Spectrum management class 1 transverse balance requirement.........................13
5.3.1.3 Longitudinal Output Voltage..................................................................................14
1.1
1.2
1.3
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5.3.1.4 Spectrum management class 1 deployment guidelines .......................................14
5.3.2 Spectrum management class 2....................................................................................14
5.3.2.1 Spectrum management class 2 PSD and total average power limitation.............14
5.3.2.2 Spectrum management class 2 transverse balance requirement.........................14
5.3.2.3 Longitudinal Output Voltage..................................................................................14
5.3.2.4 Spectrum management class 2 deployment guidelines .......................................14
5.3.3 Spectrum management class 3....................................................................................14
5.3.3.1 Spectrum management class 3 PSD and total average power limitation.............15
5.3.3.2 Spectrum management class 3 transverse balance requirement.........................15
5.3.3.3 Longitudinal Output Voltage..................................................................................15
5.3.3.4 Spectrum management class 3 deployment guidelines .......................................15
5.3.4 Spectrum management class 4....................................................................................15
5.3.4.1 Spectrum management class 4 PSD and total average power limitation.............15
5.3.4.2 Spectrum management class 4 transverse balance requirement.........................15
5.3.4.3 Longitudinal Output Voltage..................................................................................16
5.3.4.4 Spectrum management class 4 deployment guidelines .......................................16
5.3.5 Spectrum management class 5....................................................................................16
5.3.5.1 Spectrum management class 5 PSD and total average power limitation.............16
5.3.5.2 Spectrum management class 5 transverse balance requirement.........................16
5.3.5.3 Longitudinal Output Voltage..................................................................................16
5.3.5.4 Spectrum management class 5 deployment guidelines .......................................17
5.3.6 Spectrum management class 6....................................................................................17
5.3.6.1 Spectrum management class 6 PSD and total average power limitation.............17
5.3.6.2 Spectrum management class 6 transverse balance requirement.........................17
5.3.6.3 Longitudinal Output Voltage..................................................................................17
5.3.6.4 Spectrum management class 6 deployment guidelines .......................................17
5.3.7 Spectrum management class 7....................................................................................17
5.3.7.1 Spectrum management class 7 PSD and total average power limitation.............17
5.3.7.2 Spectrum management class 7 transverse balance requirement.........................18
5.3.7.3 Longitudinal Output Voltage..................................................................................18
5.3.7.4 Spectrum management class 7 deployment guidelines .......................................18
5.3.8 Spectrum management class 8....................................................................................18
5.3.8.1 Spectrum management class 8 PSD and total average power limitation.............18
5.3.8.2 Spectrum management class 8 transverse balance requirement.........................18
5.3.8.3 Longitudinal Output Voltage..................................................................................18
5.3.8.4 Spectrum management class 8 deployment guidelines .......................................18
5.3.9 Spectrum management class 9....................................................................................19
5.3.9.1 Spectrum management class 9 PSD and total average power limitation.............19
5.3.9.2 Spectrum management class 9 transverse balance requirement.........................19
5.3.9.3 Longitudinal Output Voltage..................................................................................19
5.3.9.4 Spectrum management class 9 deployment guidelines .......................................19
5.4
Technology Specific Guidelines ...................................................................................19
5.4.1 2B1Q SDSL..................................................................................................................19
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5.4.1.1 2B1Q SDSL PSD and total average power limitation ...........................................20
5.4.1.2 2B1Q SDSL transverse balance requirement.......................................................20
5.4.1.3 2B1Q SDSL Longitudinal Output Voltage .............................................................20
5.4.1.4 2B1Q SDSL deployment guidelines......................................................................20
5.4.2 G.991.2, Annex A, Symmetric PSDs (G.shdsl) ............................................................20
5.4.2.1 G.shdsl PSD and total average power limitation...................................................21
5.4.2.2 Transverse balance requirement ..........................................................................21
5.4.2.3 Longitudinal Output Voltage..................................................................................21
5.4.2.4 G.shdsl deployment guidelines .............................................................................21
5.4.3 TC-PAM 776/784 kbps Asymmetric PSD (HDSL4 technology) ...................................21
5.4.3.1 PSD and total average power limitation................................................................21
5.4.3.2 Transverse balance requirement ..........................................................................22
5.4.3.3 Longitudinal Output Voltage..................................................................................22
5.4.3.4 Deployment guidelines..........................................................................................22
6.
Conformance testing methodology ..............................................................................22
6.1
General conformance criteria ......................................................................................22
6.2
PSD conformance criteria unique to spectrum management classes .........................23
6.2.1 Specific conformance criteria for spectrum management class 1 ...............................23
6.2.2 Specific conformance criteria for spectrum management class 2 ..............................23
6.2.3 Specific conformance criteria for spectrum management class 3 ...............................24
6.2.4 Specific conformance criteria for spectrum management class 4 ..............................24
6.2.5 Specific conformance criteria for spectrum management class 5 ...............................24
6.2.6 Specific conformance criteria for spectrum management class 6 ...............................24
6.2.7 Specific conformance criteria for spectrum management class 7 ..............................24
6.2.8 Specific conformance criteria for spectrum management class 8 ..............................24
6.2.9 Specific conformance criteria for spectrum management class 9 ..............................24
6.3
Technology Specific PSD conformance criteria...........................................................24
6.3.1 Specific conformance criteria for 2B1Q SDSL .............................................................24
6.3.2 Specific conformance criteria for G.shdsl (Annex A, Symmetric PSDs) ......................25
6.3.3 Specific conformance criteria for HDSL4 technology...................................................26
6.4
PSD and total average power measurement procedure..............................................26
6.4.1 Test circuit for PSD and total average power measurement .......................................26
6.4.2 Calibration of the test circuit and termination impedance ............................................26
6.4.3 Operation of the DUT ...................................................................................................27
6.4.4 Total average power measurement procedure ............................................................27
6.4.5 Power spectral density (PSD) measurement procedure..............................................27
6.4.5.1 PSD resolution bandwidth.....................................................................................27
6.4.5.2 PSD Measurement time duration ..........................................................................27
6.5
Short-term stationary conformance criteria ..................................................................27
6.5.1 Determination of whether to apply short-term stationary conformance criteria ...........27
6.5.2 Continuous mode for conformance testing ..................................................................28
6.5.3 Frequency domain requirements .................................................................................28
6.5.3.1 Continuous mode testing ......................................................................................28
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6.5.3.2 Short-term stationary mode testing .......................................................................28
6.5.4 Time domain requirements ..........................................................................................28
6.6
Transverse balance testing methodology ....................................................................28
6.7
Longitudinal output voltage testing methodology.........................................................29
Annex A: Evaluation of interference from new technologies (Method B).......................................46
A.1
Goals and framework for evaluation ............................................................................46
A.2
Analytical Method: Detailed crosstalk margin evaluations ...........................................46
A.2.1 Loop Characterization ..................................................................................................47
A.2.1.1
Cable primary constants RLCG.........................................................................47
A.2.1.2
Twisted-pair cable insertion gain transfer function............................................47
A.2.2 Cable crosstalk models ................................................................................................47
A.2.2.1
Near end crosstalk, NEXT .................................................................................48
A.2.2.1.1 Simplified NEXT model.....................................................................................48
A.2.2.1.2 Two-piece NEXT model....................................................................................48
A.2.2.2
Far end crosstalk, FEXT....................................................................................48
A.2.2.3
FSAN method for combining crosstalk contributions from unlike types of disturbers
..........................................................................................................................48
A.2.2.3.1 Example application of two NEXT terms ..........................................................49
A.2.2.3.2 Example application of three FEXT terms ........................................................49
A.2.2.4
Total noise power spectral density ....................................................................49
A.2.3 General Methodology ...................................................................................................49
A.2.4 DFE-based PAM signals (e.g., 2B1Q BRI and HDSL).................................................50
A.2.5 DFE-based QAM/CAP signals .....................................................................................50
A.2.6 DMT margin computations ...........................................................................................51
A.2.7 Spectral Compatibility Computation Results................................................................51
A.2.8 Common parameters for computing spectral compatibility ..........................................51
A.3
Compatibility with voicegrade services and technologies ............................................52
A.3.1 Description of voicegrade services and technologies ..................................................52
A.3.1.1
Speech signals ..................................................................................................52
A.3.1.2
Single and dual tone signals..............................................................................52
A.3.1.3
Low frequency (< 100 Hz) signals .....................................................................52
A.3.1.4
Digital data.........................................................................................................52
A.3.1.5
Analog data .......................................................................................................52
A.3.2 Voicegrade evaluation..................................................................................................53
A.3.2.1
Evaluation loop ..................................................................................................53
A.3.2.2
Reference crosstalk environment......................................................................53
A.3.2.3
Crosstalk noise and peak power levels computation ........................................53
A.3.2.3.1 Voiceband NEXT PSD......................................................................................53
A.3.2.3.2 Total voiceband NEXT noise limit.....................................................................53
A.3.3 Spectral compatibility of voicegrade systems with basis systems ...............................54
A.4
Compatibility with Enhanced Business Services .........................................................54
A.4.1 Description of Enhanced Business Services ...............................................................54
A.4.1.1
Speech signals ..................................................................................................54
A.4.1.2
Signaling functions ............................................................................................54
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A.4.2 Enhanced Business Service Evaluation.......................................................................54
A.4.2.1
Evaluation loop ..................................................................................................54
A.4.2.2
Reference crosstalk environment......................................................................54
A.4.2.3
Crosstalk noise and peak power levels computation ........................................55
A.4.2.3.1 Signaling Band NEXT PSD...............................................................................55
A.4.3 Spectral compatibility of Enhanced Business Services with basis systems ................55
A.5
Compatibility with T1.410 .............................................................................................55
A.5.1 Computation of DDS Performance – Margin Computation for AMI Transceivers .......55
A.5.2 Evaluation loops ...........................................................................................................56
A.5.3 Reference crosstalk environment.................................................................................57
A.5.4 Margin computation......................................................................................................57
A.6
Compatibility with BRI ..................................................................................................57
A.6.1 Evaluation loops ...........................................................................................................57
A.6.2 Reference Crosstalk environment................................................................................57
A.6.3 Margin Computation .....................................................................................................57
A.7
Compatibility with HDSL...............................................................................................58
A.7.1 Evaluation loops ...........................................................................................................58
A.7.2 Reference crosstalk environment.................................................................................58
A.7.3 Margin computation......................................................................................................58
A.8
Compatibility with ADSL and RADSL technologies .....................................................58
A.8.1 Compatibility with ADSL ...............................................................................................58
A.8.1.1
Parameters for computing spectral compatibility with ADSL.............................58
A.8.1.2
Evaluation loops and performance levels..........................................................59
A.8.1.3
Reference crosstalk environments ....................................................................59
A.8.1.4
Margin computation ...........................................................................................59
A.8.2 Compatibility with RADSL ............................................................................................61
A.9
Compatibility with HDSL2.............................................................................................61
A.9.1 Evaluation loops ...........................................................................................................61
A.9.2 Reference crosstalk environment.................................................................................61
A.9.3 Margin computation......................................................................................................61
A.10
Compatibility with 2B1Q SDSL.....................................................................................62
A.10.1 Evaluation loops and performance levels ....................................................................62
A.10.2 Reference crosstalk environment.................................................................................62
A.10.3 Margin computation......................................................................................................62
A.10.4 2B1Q SDSL Technology Specification.........................................................................62
A.10.4.1 Power Spectral Density .....................................................................................62
A.10.4.2 Performance ......................................................................................................63
A.10.4.2.1 Test loops .....................................................................................................63
A.10.4.2.2 Test Setup.....................................................................................................63
A.10.4.2.3 Crosstalk noise .............................................................................................63
A.10.4.2.4 Reach target .................................................................................................63
A.10.4.3 Return loss ........................................................................................................63
A.10.4.4 Longitudinal Balance .........................................................................................63
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A.11
Customer end-point separation....................................................................................63
Annex B: Loop Information .............................................................................................................77
B.1
General.........................................................................................................................77
B.1.1 The loop environment...................................................................................................77
B.1.1.1 Background noise .................................................................................................78
B.1.1.2 Impulse noise ........................................................................................................78
B.1.1.3 Radio frequency interference (RFI).......................................................................78
B.1.1.4 Structural cable faults............................................................................................78
B.1.1.5 The loop environment ...........................................................................................78
B.1.1.6 Telephone cable and subscriber loop structures ..................................................78
B.1.2 Loop plant design rules: resistance design..................................................................79
B.1.3 Loop plant design rules: carrier serving area (CSA) ....................................................80
B.1.4 Distribution area (DA)...................................................................................................81
B.1.5 Loop statistics...............................................................................................................81
B.2
AWG and metric cable: diameters and DC resistance and capacitance .....................81
B.3
Cable primary constants (RLCG) characterization ......................................................82
B.3.1 Transmission-Line Characterization.............................................................................82
B.3.1.1 “ABCD” modeling ..................................................................................................82
B.3.1.2 Transmission-line RLCG characterization ............................................................83
B.3.1.3 Power for transmission lines .................................................................................86
B.3.1.4 Reflection coefficients ...........................................................................................87
B.3.1.5 Characterization of a bridged tap section – a three-port.......................................87
B.3.1.6 Computation of transfer function ...........................................................................88
B.3.1.7 Relationship of transfer function and “insertion loss” ............................................88
B.3.1.7.1 Measurement Procedure ..................................................................................89
B.3.1.7.2 Curve-fitting.......................................................................................................90
B.3.2 26-AWG Multi-pair Cable .............................................................................................90
B.3.3 24-AWG multi-pair cable ..............................................................................................90
B.3.4 22-AWG Twisted Pair PIC Cable .................................................................................90
B.3.5 FP .................................................................................................................................90
B.3.6 Category-5 Twisted Pair...............................................................................................90
B.3.7 Two-Pair Twisted Drop.................................................................................................90
B.3.8 Two-Pair Quaded Drop ................................................................................................91
B.3.9 Flat-Pair Drop ...............................................................................................................91
B.3.10 Additional Models .........................................................................................................91
B.4
Cable crosstalk models ................................................................................................91
B.4.1 Near end crosstalk, NEXT............................................................................................91
B.4.2 Far end crosstalk, FEXT ..............................................................................................93
B.4.3 FSAN method: combining crosstalk from unlike disturber types .................................93
B.4.3.1 Base models for NEXT and FEXT ........................................................................93
B.4.3.2 Combining crosstalk from mixed disturber types ..................................................94
B.4.3.3 Application to two NEXT terms .............................................................................94
B.4.3.4 Application to FEXT terms ....................................................................................95
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B.4.3.5 Crosstalk is non-decreasing..................................................................................96
B.4.3.6 All disturbers are treated equally...........................................................................96
B.4.3.7 Adding NEXT and FEXT .......................................................................................96
Annex C: Probability of error estimation.......................................................................................117
C.1
Effect of input bit sequence........................................................................................118
C.2
Period of injected “Gaussian” noise ...........................................................................118
C.3
dB margin and importance sampling..........................................................................119
Annex D: Additional spectrum management topics currently under study by the formulating committee
of this standard...........................................................................................................120
Annex E: Time varying, user data-dependent crosstalk from T1 and DDS services ...................121
Annex F: Non-continuous CO signaling events............................................................................124
F.1
Ringing .......................................................................................................................124
F.2
Supervision (hook flash).............................................................................................124
F.3
Dial Pulse ...................................................................................................................124
Annex G: ADSL Calculated Capacities ........................................................................................128
Annex H: Technology effects of and on T1 carrier systems.........................................................129
H.1
T1 Carrier ...................................................................................................................129
H.1.1 Compatibility with AMI T1...........................................................................................129
H.1.1.1
Evaluation Loops .............................................................................................129
Annex I: C-code............................................................................................................................130
Annex J: Derivation of Equivalent Working Length Definition ......................................................131
Annex K: Interference Level Evaluation for the Case of Unequal Loop Length for Interfering and
Interfered Systems .....................................................................................................133
K.1
General Methodology .................................................................................................133
Annex L: Interference Level Evaluation for Applications that locate a TU at an Intermediate point
between the CO and CI..............................................................................................135
L.1
Applications that use Intermediate Repeaters/Amplifiers ..........................................135
L.2
Applications that use Intermediate TU-C Devices .....................................................138
Annex M: Rationale behind General Conformance Criteria in Clause 6.1...................................141
M.1
Introduction.................................................................................................................141
M.2
Rationale for using the Average Normalized PSD as a “soft” limit ............................141
M.2.1 Derivation of Equivalent Noise for DFE and DMT receivers ......................................141
M.2.2 Criterion for equivalency of SM Class PSD template and member PSD..................143
M.2.3 Developing Compliance Criteria to minimize the error...............................................144
Annex N: Informative references..................................................................................................149
TABLES
Table 1 - SM class 1 PSD template definition ....................................................................................30
Table 2 - Minimum transverse balance requirements ........................................................................30
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Table 3 - SM class 2 PSD template definition ....................................................................................31
Table 4- SM class 3 PSD template definition .....................................................................................31
Table 5 - PSD mask definition for downstream transmission from SM class 4 TU-C .......................32
Table 6 - PSD mask definition for upstream transmission from SM class 4 TU-R.............................32
Table 7 - PSD template definition for downstream transmission from SM class 5 TU-C...................32
Table 8 - SM class 7 PSD template definition ....................................................................................32
Table 9 - SM class 8 PSD template definition. ...................................................................................33
Table 10 – PSD template definition for downstream transmission from SM class 9 TU-C................33
Table 11 - PSD template definition for upstream transmission from SM class 9 or SM class 5 TU-R
............................................................................................................................................33
Table 12 - Deployment guidelines for 2B1Q SDSL............................................................................34
Table 13 - G.shdsl Symmetric PSD parameters ................................................................................34
Table 14 - Deployment guidelines for G.shdsl ...................................................................................35
Table 15 - PSD template definition for downstream transmission from HDSL4 TU-C.......................35
Table 16 - PSD template definition for upstream transmission from HDSL4 TU-R ...........................36
Table 17 - Termination impedance.....................................................................................................36
Table 18 - Resolution bandwidth for measuring a DUT PSD for conformance with SM classes 1, 2, 3, 4,
7 and 8 ................................................................................................................................36
Table 19 - Resolution bandwidth for measuring a DUT PSD for conformance with SM classes 5 and 9
............................................................................................................................................37
Table 20 - Resolution bandwidth for measuring a DUT PSD for conformance with SM class 6 .......37
Table 21 - Summary of transverse balance testing criteria................................................................37
Table 22 - Maximum longitudinal output voltage limit ........................................................................37
Table A. 1 - Curve fit parameters for 26-AWG twisted-pair cable ......................................................64
Table A. 2 - Constants for Two-Piece NEXT Model...........................................................................64
Table A.3 - Code for DFE PAM SNR computation.............................................................................65
Table A. 4 - Code for DFE QAM/CAP computation ...........................................................................66
Table A.5 - Matlab code to set-up ADSL margin computation ...........................................................67
Table A. 6 - Matlab Code to compute a DMT margin.........................................................................69
Table A. 7 - Spectral compatibility with the basis systems.................................................................70
Table A. 8 - HDSL2_delta (in dB) for various test crosstalk combinations ........................................71
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Table A. 9 - 2B1Q SDSL data rate and associated spectrum management classes.........................71
Table B.1 – American wire gauge (AWG) and metric wire.................................................................97
Table B. 2 - Cable model parameters for 26-AWG twisted pair cable ...............................................98
Table B. 3 - Primary constants for 26-AWG twisted pair cable ..........................................................98
Table B. 4 - Cable parameters for 26-AWG PIC air core ...................................................................99
Table B. 5 - Cable parameters for 26-AWG filled PIC........................................................................100
Table B. 6 - Cable model parameters for 24-AWG twisted pair cable ...............................................101
Table B. 7 - Primary constants for 24-AWG twisted pair cable ..........................................................101
Table B. 8 - Cable parameters for 24-AWG PIC air core ...................................................................102
Table B. 9 - Cable parameters for 22-AWG PIC air core ...................................................................103
Table B. 10 - Cable model parameters for FP (1.14 mm flat cable) ..................................................104
Table B. 11 - Primary constants for FP (1.14 mm flat cable) .............................................................105
Table B. 12 - Cable model parameters for category 5 twisted pair ....................................................105
Table B. 13 - Primary constants for category 5 twisted pair...............................................................105
Table B. 14 - Cable parameters, two-pair twisted drop......................................................................106
Table B. 15 - Cable parameters, two-pair quad drop .........................................................................108
Table B. 16 - Cable parameters, flat-pair Drop ..................................................................................110
Table G. 1: ADSL Calculated Capacity ..............................................................................................128
Table H. 1: Coefficients for T1 repeater input filtering gain equation.................................................129
Table I. 1: C-code for DMT margin computation ................................................................................130
FIGURES
Figure 1 - SM class 1 PSD template ..................................................................................................38
Figure 2 - SM class 2 PSD Template .................................................................................................38
This is a draft document and thus, is still dynamic in nature.
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Figure 3 - SM class 3 PSD template ..................................................................................................39
Figure 4 - PSD mask for downstream transmission from SM class 4 TU-C ......................................39
Figure 5 - PSD mask for upstream transmission from SM class 4 TU-R...........................................40
Figure 6 - PSD template for downstream transmission from SM class 5 TU-C .................................40
Figure 7 - PSD template for upstream transmission from SM class 5 or SM class 9 TU-R...............41
Figure 8 - SM class 7 PSD template ..................................................................................................41
Figure 9 - SM class 8 PSD template ..................................................................................................42
Figure 10- PSD template for downstream transmission from SM class 9 TU-C ................................42
Figure 11 - PSD template for downstream transmission from HDSL4 technology TU-C ..................43
Figure 12 - PSD template for upstream transmission from HDSL4 technology TU-R .......................43
Figure 13 - PSD and total average power measurement setup .........................................................44
Figure 14 - Example PSD and total average power measurement setup..........................................44
Figure 15 - Illustrative test configuration for transverse balance conformance testing ......................45
Figure 16 - Measurement method for longitudinal voltage.................................................................45
Figure A. 1 – Two-piece NEXT model and simplified NEXT model ...................................................72
Figure A. 2 – Crosstalk into a Basis System: NEXT and FEXT .........................................................73
Figure A. 3 – Simulation Model for Reference and New Crosstalk into Downstream Receiver ........73
Figure A. 4 - Process flow for spectral compatibility calculations.......................................................74
Figure A. 5 – 2B1Q SDSL PSD at several data rates ........................................................................75
Figure A. 6 - 2B1Q SDSL PSD at several data rates.........................................................................75
Figure A. 7 - Minimum return loss for 784 kbps 2B1Q SDSL system ................................................76
Figure A. 8 - Longitudinal balance for 784 kbps 2B1Q SDSL system ...............................................76
Figure B.1 – Loop ABCD parameters, impedance and voltages .......................................................112
Figure B. 2 – Two-port network model. ..............................................................................................113
Figure B.3 – Incremental section of twisted-pair transmission line. ..................................................113
Figure B.4 – Simple load circuit for power analysis ...........................................................................113
Figure B.5 – Examples of two-port cascades for twisted-pair transmission line configurations ........114
Figure B. 6 – Near end crosstalk (NEXT)...........................................................................................114
Figure B.7 – NEXT power sum losses for 25 pairs of PIC cable binder group ..................................115
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Figure B.8 – Comparison of Simplified Model NEXT with Measured NEXT ......................................115
Figure B.9 – Far end crosstalk (FEXT)...............................................................................................116
Figure B.10 – Comparison of Model FEXT with Measured FEXT......................................................116
Figure E. 1 - Examples of T1 power spectral density variations ........................................................122
Figure E. 2 - Examples of DDS power spectral density variations.....................................................122
Figure E. 3 - Data dependent power changes in a wide band due to T1 data patterns.....................123
Figure E. 4 - Data dependent power changes in a narrow band due to T1 data patterns .................123
Figure F. 1 - Standard ringing potential with best case start/end.......................................................126
Figure F. 2 - Standard ringing potential worst case start/end ............................................................126
Figure F. 3 - Ringing waveforms (worst case generalization) ............................................................127
Figure F. 4 - Triple ringing interval .....................................................................................................127
Figure F. 5 - Simple battery feed arrangement ..................................................................................127
Figure L. 1 - Crosstalk into Basis System Downstream Receiver: NEXT & FEXT with Repeater Device
............................................................................................................................................136
Figure L. 2 - Simulation Model for Reference and New Crosstalk into Downstream Receiver with
Repeater Device .................................................................................................................136
Figure L. 3 - Crosstalk into Basis System Upstream Receiver: NEXT & FEXT with Repeater Device
............................................................................................................................................137
Figure L. 4 - Simulation Model for Reference and New Crosstalk into Upstream Receiver with Repeater
Device .................................................................................................................................137
Figure L. 5 - Crosstalk into Basis System Downstream Receiver: NEXT & FEXT with Intermediate TU-C
Device .................................................................................................................................139
Figure L. 6 - Simulation Model for Reference and New Crosstalk into Downstream Receiver with
Intermediate TU-C Device ..................................................................................................139
Figure L. 7 - Crosstalk into Basis System Upstream Receiver: NEXT & FEXT with Intermediate TU-C
Device .................................................................................................................................140
Figure L. 8 - Simulation Model for Reference and New Crosstalk into Upstream Receiver with
Intermediate TU-C Device ..................................................................................................140
Figure M. 1 - Non-Overlapping PSD example....................................................................................144
Figure M. 2 - Error for non-overlapping Spectra.................................................................................146
Figure M. 3 - PSD Limit Example .......................................................................................................146
This is a draft document and thus, is still dynamic in nature.
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Figure M. 4 - Error for 4 Example Cases with PSD Limit ...................................................................147
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ANSI T1.417
Foreword (This foreword is not part of American National Standard T1.XXX-2000)
Accredited Standards Committee T1, Telecommunications serves the public through improved understanding
between carriers, customers, and manufacturers. Technical Subcommittee T1E1 of Committee T1 develops
telecommunications standards and technical reports related to various digital subscriber line technologies. This
standard is intended to be a living document, subject to revision and updating as warranted by advances in
network and equipment technology.
This standard provides spectrum management requirements and recommendations for the administration of
services and technologies that use metallic subscriber loop cables. Spectrum management is the administration
of the loop plant in a way that provides spectral compatibility for services and technologies that use pairs in the
same cable. In order to achieve spectral compatibility, the ingress energy that transfers into a loop pair, from
services and transmission system technologies on other pairs in the same cable, must not cause an unacceptable
degradation of performance. In addition, the egress energy from a particular loop pair must not transfer into
other pairs in a manner that causes an unacceptable degradation in the performance of services and technologies
on those pairs. This standard includes signal power limits and technology deployment guidelines for the digital
subscriber line spectrum management classes defined herein. It also provides a generic analytical method to
determine spectral compatibility.
Because of the wide range of network switching systems, network transport systems, subscriber loop plant, and
customer installations in North America, conformance with this standard does not guarantee spectral
compatibility or acceptable performance under all possible operating conditions.
ANSI guidelines specify two categories of requirements: mandatory and recommendation. The mandatory
requirements are designated by the word shall and recommendations by the word should. Where both a
mandatory requirement and a recommendation are specified for the same criterion, the recommendation
represents a goal currently identifiable as having distinct compatibility or performance advantages.
There are 14 annexes in this standard. Annex A is normative and considered to be part of this standard; Annexes
B - N are informative and are not considered part of this standard, that is, they do not include requirements but
provide information that may be helpful to users of this standard.
Suggestions for improvement of this standard are welcome. They should be sent to the Alliance for
Telecommunications Industry Solutions, T1 Secretariat, 1200 G Street NW, Suite 500, Washington, DC 20005.
This standard was processed and approved for submittal to ANSI by Accredited Standards Committee on
Telecommunications, T1. Committee approval of the standard does not necessarily imply that all members voted
for its approval. At the time it approved this standard, the T1 Committee had the following members:
E. R. Hapeman, Chair
W.R. Zeuch, Vice-Chair
S. D. Barclay, Secretary
EXCHANGE CARRIERS
Exchange Carrier Member .................................................................................................. J.F. Baskin
Exchange Carrier Member ...................................................................................................... J.L. Eitel
INTEREXCHANGE CARRIERS
Interexchange Carrier Member................................................................................................... J. Lord
Interexchange Carrier Member............................................................................................. M. Neibert
MANUFACTURERS
Manufacturer Member................................................................................................................L. Troy
Manufacturer Member........................................................................................................ M. Woinsky
F. Lucas (Alt.)
GENERAL INTEREST
General Interest Member ....................................................................................................... G. Lynch
General Interest Member ....................................................................................................... R. J. Hall
This is a draft document and thus, is still dynamic in nature.
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ANSI T1.417
Technical Subcommittee T1E1 on Interfaces, Power and Protection of Networks, which is responsible for the
development of this standard, had the following members:
Ed Eckert, Chair
John Roquet, Vice-Chair
Rick Townsend, Secretary
Working Group T1E1.4 on DSL Access, which had the technical responsibility during the development of this
standard, had the following members:
Thomas J. J. Starr,
Syed A. Abbas
Trone Bishop
Nigel Cole
Chairman
Robyn Aber
Ray Blackham
Terry Cole
Massimo Sorbara, ViceOscar Agazzi
Steve Blasingame
Marty Colombatto
Chairman
Cajetan M. Akujuobi
R. T. Bobilin
Kevin Cone
Ron McConnell, Secretary
Ron Allen
Gary Bolton
Greg Copeland
Editors:
Subra Ambati
Jan Bostrom
Graham Copley
Craig Valenti,
Tariq Amjed
Mark F. Bowen
Lawrence Corbett
John E. Roquet,
Candare
M.
Anderson
Bruce
Bowie
Mauro Costa
Richard A. McDonald,
Ephraim Arnon
Peter Brackett
Ray Countermann
Behrooz Rezvani
xvi
James Aslanis
Richard Brandt
Bill Crane
Keith Atwell
Dave Brier
Phil Crawby
Hiromitsu Awai
Les Brown
David Cummings
Jein Baek
Randy Brown
Kim Currie
Scott J. Baer
Curtis Brownmiller
Aaron Dagen
Rupert Baines
William Buck
Tom Daly
H. Charles Baker
William Buckley
Tamar Danon
LeRoy Baker
John Bush
Michel Darveau
John T. Balinski
Richard Cam
Jim Dell
Chuck Balogh
John Camagna
Michael Demjanenko
Art Barabell
Patrick Cameron
Shuang Deng
Uri Baror
Jim Carlo
Andre’ P. des Rosiers
John Barselloti
Art Carlson
Philip DesJardins
Roy Batruni
Paulus Carpelan
Franz Dielacher
Don Bellenger
C. A. Carpenter
Curtis Dodd
Daniel Bengtssen
Ken Cavanaugh
Jean-Louis Dolmeta
Rafi Ben-Michael
Guy Cerulli
Paul Donaldson
Ben Bennett
Paul Chang
Guojie Dong
Bill Bergman
Yen T. Chan
Barry Dropping
Dev Bhattacharya
Giovanni Cherubini
Bernard Dugerdil
Nigel Billington
Jacky Chow
Craig Edwards
Bora Biray
Peter Chow
George Eisler
Larry Bishop
John Cioffi
Tsur Eitan
Richard Bishop
Alan Cohen
Earl Emerson
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Dan Etz-Hadar
Tom Haycock
Marc Kimpe
Joseph Markee
Dave Evans
Shahin Hedayat
Avi Kliger
Ron Marquardt
Vedat Eyuboglu
Chris Heegard
Ron Knipper
Doug W. Marshall
Charles Fadel
Peter Niels Heller
Robert Kniskern
Al Martin
Rocky Flaminio
Brian Henrichs
Ken Ko
Kazuya Matsumoto
Kay Fleskes
Malcolm Herring
Yosef Kofman
Bo Matthys
Steve Follett
Hanan Herzberg
Jouni Koljonen
Thomas Maudoux
Al Forcucci
Curt Hicks
Hajime Koto
Jack Maynard
Klaus Fosmark
Amir Hindie
Tetsu Koyama
Gary McAninch
Kevin Foster
Minnie Ho
Kamila S. Kraba
Kent McCammon
Vladimir Friedman
David Hoerl
James Kroll
John McCarter
Hans-Joerg Frizlen
David Holien
Philip J. Kyees
Shawn McCaslin
Robin Gangopadhya
Mahbub Hoque
Robert LaGrand
Ronald C. McConnell
Clete Gardenhour
James C. Horng
T. K. Lala
Keith McDonald
Juan Garza
Gary R. Hoyne
Chi-Ying Lan
Richard A. McDonald
Amit Gattani
Gang Huang
John Langevin
Peter Melsa
Lajos Gazsi
Les Humphrey
Martin LaRose
Denis J. G. Mestdagh
Tom Geary
Marlis Humphrey
Steven C. Larsen
Harry Mildonian
Nabil Gebrael
Cannon Hwu
Mike Lassandrello
Robert R. Miller
Al Gharakhanian
Ishai Ilani
George J. Lawrence
Dave Milliron
Emil Ghelberg
Greg Ioffe
Dong Chul Lee
Khashayar Mirfakhraei
Mike Gilbert
Mikael Isaksson
Howard Levin
Steve Milkan
Jim Girardeau
Tomokazu Ito
Gabriel Li
Cory Modlin
Hugh Goldberg
Krista S. Jacobsen
Haixiang Liang
Michael Moldoveanu
Yuri Goldstein
Ken Jacobson
Ze’ev Lichtenstein
Steven Monti
David Goodman
Charlie Jenkins
Simon Lin
David R. Moon
Richard Goodson
Ralph Jensen
Jari Lindholm
Lane Moss
Steven Gordon
Scott Jezwinski
Stan Ling
Kevin Mullaney
Linda Gosselin
Jim Jollota
James Liou
Joe Muller
Peter T. Griffiths
Albin Johansson
Dave Little
Babak Nabili
Glen Grochowski
David C. Jones
Fuling Frank Liu
Donovan Nak
John Gruber
Edward Jones
Qing Li Liu
Randy Nash
Sanjay Gupta
Ragnar Jonsson
Valentino Liva
Frank Navavi
L. B. Gwinn
Anjal Joshi
G. W. Lloyd
Gil Naveh
Cliff Hall
John Joyce
Bob Locklear
Gunter F. Neumeier
Rabah Hamdi
Vern Junkmann
Guozhu Long
Mai-Huang Nguyen
Rodney Hanneman
Wen-Juh Kang
Pini Lozowick
Ramin Nobakht
Chris Hansen
Satoru Kawanago
Perry Lu
Andy Norrell
Gopal Harikumar
Ken Kerpez
Ahmed Madani
Rao Nuthalapati
Roy Harvey
Kamran Khadavi
Rabih Makarem
Stephen Oh
Roy Harvey
Babak Khalaj
Marcus Maranhao
Franz Ohen
Josef Hausner
Sayfe Kiaei
Dan Marchok
Hans Öhman
This is a draft document and thus, is still dynamic in nature.
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ANSI T1.417
Yusaku Okamura
Behrooz Rezvani
Don Skinfill
John Ulanskas
Kazu Okazaki
Ron Riegert
Joe Smith
Juan Ramon Uribe
Vladimir Oksman
Terry Riley
P. Norman Smith
Peter Vaclavik
Al Omran
Boaz Rippin
R. K. Smith
Craig Valenti
Mike O’Neill
Jorge Rivera
Stephen Smith
Nick van Bavel
Aidan O’Rourke
Richard Roberts
Edwin J. Soltysiak
Danny Van Bruyssel
Tom O’Shea
Silvana Rodrigues
Ranjan Sonalkar
Harry van der Meer
Eric Paneth
John E. Roquet
Massimo Sorbara
Frank Van der Putten
Panos Papamichalis
John Rosenlof
Andrew Sorowka
Dick van Gelder
Yatendra K. Pathak
Eric J. Rossin
Walt Soto
Jeff Van Horne
Shimon Peleg
Mike Rude
J. Scott Spradley
M. Vautier
Michael Pellegrini
Mark Russell
Paul Spruyt
Robert L. Veal
Matt Pendleton
Christopher J. Rust
Patrick Stanley
Dale Veeneman
Larry Perron
Kimmo K. Saarela
Tom Starr
Rami Verbin
Todd Pett
Ken Sakanashi
Mark Steenstra
Pieter Versavel
Willie Picken
Debbi Sallee
William Stewart
Raman Viswanathan
Ashley Pickering
Henry Samueli
James Stiscia
Jeff Waldhuter
Thierry Pollet
Hal Sanders
Jeff Strait
Josef Waldinger
Michael Polley
Wayne Sanderson
Caleb Strittmatter
Qi Wang
Bob Poniatowski
Jamal Sarma
Richard Stuart
Brian Waring
Boaz Porat
Sabit Say
Ray Subbankar
David Waring
Ron Porat
Denny Schart
Henri Suyderhoud
Dewight Warren
Carl Posthuma
Kevin Schneider
James Szeliga
Curtis Waters
Philip Potter
Gary Schultz
Hiroshi Takatori
Alan Weissberger
Amit Preuss
Bob Scott
Daryl C. Tannis
J. J. Werner
Aleksandar Purkovic
Linda Seale
Larry Taylor
Rick Wesel
Gordon Purtell
Reuven Segev
Matthew Taylor
Greg Whelan
Dan Queen
Radu Selea
Steve Taylor
Albert White
Jim Quilici
Ahmed Shalash
Gary Tennyson
Song Wong
Jack Quinnell
Mark Shannon
Jack Terry
Bernard E. Worne
Ariel Radsky
Donald P. Shaver
Rainer Thoenes
Cliff Yackle
Selem Radu
Tom Shen
Vernon Tice
Jack Yang
Sreen Raghavan
Greg Sherrill
Ed Tirakian
Han Yeh
Ali Rahjou
Tzvi Shukhman
Cherry Tom
Soobin Yim
Jeffrey M. Rakos
Eli Shusterman
Chi-Lin Tom
Kyung-Hyun Yoo
Avi Rapaport
Rex Siefert
Antti Tommiska
Gavin Young
Janice Rathmann
Kevin Sievert
J. Alberto Torres
Irvin Youngberg
Dennis Rauschmaier
Richard Silva
Richard L. Townsend
Xiaolong Yu
Richard Rawson
Doug Silveira
Bob Tracy
Shaike Zalitzky
Gord Reesor
Peter Silverman
Dwen-Ren Tsai
Xuming Zhang
Dave Reilly
Mark Simkins
Marcos Tzannes
George Zimmerman
John Reister
Kamran Sistanizadeh
Masami Ueda
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T1E1.4/2000-002R6
ANSI T1.417
American National Standard
for Telecommunications
Spectrum Management
for Loop Transmission Systems
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
1. Scope, purpose, and application
1.1 Scope
This standard provides spectrum management requirements and recommendations for the administration of services
and technologies that use metallic subscriber loop cables. Spectrum management is the administration of the loop
plant in a way that provides spectral compatibility for services and technologies that use pairs in the same cable. In
order to achieve spectral compatibility, the ingress energy that transfers into a loop pair, from services and
transmission system technologies on other pairs in the same cable, must not cause an unacceptable degradation of
performance. In addition, the egress energy from a particular loop pair must not transfer into other pairs in a manner
that causes an unacceptable degradation in the performance of services and technologies on those pairs.
This standard includes the following types of requirements and recommendations for defined digital subscriber line
spectrum management classes and specified loop technologies:
−
power spectral density (PSD)
−
total average power
−
transverse balance
−
longitudinal output voltage
− deployment guidelines
The standard also specifies a generic analytical method (Annex A) to determine the spectral compatibility of loop
technologies that do not qualify for one of the spectrum management classes defined in this standard.
Requirements in this standard are specified for digital subscriber line systems operating on insulated solid copper
conductor twisted-pair cables used in the North American subscriber loop environment.
A system that fits in a spectrum management class complies with the Spectrum Management and Spectral
Compatibility requirements of this standard. A system that complies with Annex A complies with the Spectrum
Compatibility requirements of this standard. Compliance with a spectrum management class provides knowledge of
the characteristics of the loop system to aid deployment practices that reduce the adverse impact to certain widely
deployed loop transmission systems defined in this standard as “basis systems”.
DSL transmission systems that meet all of the specifications associated with one of the DSL spectrum management
classes are assumed to be spectrally compatible in the same binder group with all of the basis systems defined in this
standard. Meeting the specifications associated with one of the spectrum management classes in this standard does
not assure spectral compatibility with non-basis loop transmission systems.
The requirements in this issue of this standard assume that the DSL system is deployed between a Central Office
(CO) and a customer installation (CI). Applications that locate a TU at an intermediate point between the CO and
CI can, in some cases, cause crosstalk that is greater than those that use only a TU-C at the CO and a TU-R at the
CI. Applications that use intermediate TU devices between the CO and CI are addressed in clause 5.2. Clause 5.2
in this standard encompasses both applications that locate the TU-C at intermediate points between the CO and CI
and applications that use intermediate repeaters between the CO and CI.
Electromagnetic Compatibility (EMC) is outside of the scope of this standard. In addition, the spectrum
management of privately owned twisted-pair cables or customer premises twisted-pair cabling are beyond the scope
of this standard although the information in this standard may be useful in such applications.
The guidelines in this standard are based strictly on spectrum management requirements. It is understood that a
technology may have performance capabilities that either exceed or fall short of the loop lengths specified for
spectrum management.
The signals that the network and CI apply to the loop are basically of two types: normal telecommunications
transmission system voltages and currents, and voltages and currents due to maintenance activities. The normal
network and CI signals are addressed in this standard. Voltages and currents due to network maintenance activities
and abnormal voltages and currents that are the result of the environment (e.g., induced voltages and currents or
lightning) are not covered in this standard.
2
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ANSI T1.417
1.2 Purpose
The purpose of this standard is to facilitate a reasonable spectral environment for the co-existence of multiple
technologies in the loop plant with an acceptable level of crosstalk between them. When only one carrier deploys
technologies in the loop plant, it alone has the responsibility for spectral compatibility and may select any
combination of compatible loop technologies. In an unbundled loop environment however, multiple carriers utilize
pairs in the same loop cables. In such instances, if services and technologies are deployed without regard to spectral
compatibility, they may interfere with each other. This standard assumes that multiple carriers share loop cables and
that all carriers share the responsibility for spectral compatibility.
This standard provides information that will help to ensure that twisted-pair transmission systems can co-exist
without impaired operation due to crosstalk interference. The standard is intended for use by carriers to manage the
loop plant and by manufacturers in the design of loop transmission systems.
This standard was also developed to assist carriers, manufacturers, and users of products connected to local loops, to
understand the characteristics of twisted-pair loop cables. In addition, this standard can be used to determine if new
services and loop transmission system technologies are spectrally compatible with the basis systems that are defined
in this standard.
This standard is intended to be consistent with Part 68, Subpart D, of the FCC Rules and Regulations that contains
requirements for the registration of customer installation terminal equipment to protect the network from harm.
Some of the digital subscriber line spectrum management classes defined in this standard are not covered by Part 68.
If Part 68 rules are subsequently established for technologies that fall into those categories, the requirements in this
standard can be referenced. Tariffs, contracts, or regulatory acts in various jurisdictions may contain requirements
different from those in this standard.
The provisions of this standard are also intended to be consistent with applicable requirements concerning safety and
environmental conditions.
1.3 Application
This standard is applicable to twisted-pair cables that are used by multiple carriers in the local loop environment.
All of the loops described in this standard may not be universally available. For example, a loop that supports Basic
Rate ISDN (BRI) can only be provided if the facilities serving the CI are qualified to support such technology.
Because of the wide range of network switching systems, network transport systems, subscriber loop plant, and CIs
in North America, conformance with this standard does not guarantee acceptable performance under all possible
operating conditions. In some cases, measures beyond the scope of this standard may be necessary.
2. Normative references
The following standards contain provisions that, through reference in this text, constitute provisions of this
American National Standard. At the time of publication, the editions indicated were valid. All standards are subject
to revision, and parties to agreements based on this standard are encouraged to investigate the possibility of applying
the most recent editions of the standards indicated below.
ITU-T Recommendation G.991.1, High Speed Digital Subscriber Line (HDSL) Transmission System on Metallic
Local Lines.
ITU-T Recommendation G.992.1, Asymmetrical Digital Subscriber Line (ADSL) Transceivers.
ITU-T Recommendation G.992.2, Splitterless Asymmetrical Digital Subscriber Line (ADSL) Transceivers.
ITU-T Recommendation O.9, Measuring Arrangements to Assess the Degree of Unbalance about Earth.
T1.413-1998, American National Standard for Telecommunications – Network and Customer Installation Interfaces
– Asymmetrical Digital Subscriber Line (ADSL) Metallic Interface.
T1.418-2000, High Bit Rate Digital Subscriber Line - 2nd Generation (HDSL2).
T1.419-2000, Splitterless Asymmetric Digital Subscriber Line (ADSL) Transceivers.
T1.601-1999, American National Standard for Telecommunications – Integrated Services Digital Network (ISDN) –
Basic Access Interface for Use on Metallic Loops for Application on the Network Side of the NT (Layer 1
Specification).
T1.410-1992, Carrier-to-Customer Metallic Interface - Digital Data at 64kbps and Subrates.
This is a draft document and thus, is still dynamic in nature.
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ANSI T1.417
EIA/TIA TSB-31-B, February 1998; Part 68 Rationale and Measurement Guidelines; Telecommunications Industry
Association, 1998.
Committee T1 Technical Report No. 59, Single Carrier Rate Adaptive Digital Subscriber Line (RADSL).
3. Definitions, abbreviations, acronyms, and symbols
3.1 Definitions
3.1.1
3.1.2
3.1.3
3.1.4
3.1.5
3.1.6
3.1.7
3.1.8
3.1.9
3.1.10
3.1.11
3.1.12
3.1.13
3.1.14
3.1.15
3.1.16
3.1.17
3.1.18
3.1.19
3.1.20
3.1.21
3.1.22
3.1.23
4
American wire gauge: A unit used to measure the diameter of round wire.
balance: See longitudinal balance and transverse balance.
basis system: A term used in this standard to describe a loop transmission system with which DSL systems
and other new loop transmission systems are required to demonstrate spectral compatibility.
binder group: In this standard, the smallest cable unit consisting of a group of twisted pairs that are
wrapped with colored binders for identification and separation from other units.
bit error ratio: A performance measure consisting of the ratio of bits in error to the total number of bits
transmitted.
carrier: An organization that provides telecommunications services to customer installations.
central office: In this standard, the telephone building that is the origin of the outside loop plant.
conductor: A continuous solid copper or aluminum wire that has a circular cross-section.
crosstalk: Electromagnetic energy that couples into a metallic cable pair from signals on other pairs in the
same cable.
customer installation: All cabling and equipment on the customer side of the network interface.
customer premises equipment: Telecommunications equipment located at the customer installation on the
customer side of the network interface.
demarcation point: See network interface.
disturbed pair: A cable pair that has a service or technology that is experiencing crosstalk interference
from one or more other pairs in the same cable.
disturbing pair: A pair with a signal that is contributing to crosstalk interference into a service or
technology on another pair in the same cable.
downstream: The direction of transmission from the carrier Central Office to the Customer Installation.
drop wire: A type of loop cable, consisting or one or more pairs, that is used between the loop cable
terminal and the network interface device.
equivalent working length (EWL):
EWL = L 26 +
3( L 24)
4
, where L26 is the total length of
26-gauge cable in the loop excluding any bridged tap and L24 is the total length of 19, 22 or 24-gauge
cable in the loop excluding any bridged tap. All lengths are in kilofeet (kft). This formula is suitable for
use with systems operating below 1.1 MHz. See Annex J.
far-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at the other (far)
end of the cable as the transmitter of a disturbing pair.
insulated conductor: A conductor that has been surrounded with insulation that is often color-coded.
insulation: The dielectric material that surrounds a conductor and prevents it from contacting other
conductive material.
longitudinal balance: Describes the degree of symmetry with respect to ground of a two-conductor
transmission line. Longitudinal balance may be expressed as 20 times the log10 of the magnitude of the
ratio of an applied longitudinal voltage (referenced to ground) to the resultant metallic voltage.
longitudinal output voltage: The longitudinal signal that a transmitting device outputs onto a balanced
line.
loop: A communication path between the distributing frame in a carrier Central Office and the network
interface at a customer location.
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3.1.24
3.1.25
3.1.26
3.1.27
3.1.28
3.1.29
3.1.30
3.1.31
3.1.32
3.1.33
3.1.34
3.1.35
3.1.36
3.1.37
3.1.38
3.1.39
3.1.40
3.1.41
3.1.42
3.1.43
ANSI T1.417
near-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at the same
(near) end of the cable as the transmitter of a disturbing pair.
network: All equipment and facilities, including loop plant, located on the carrier side of the network
interface.
network interface: The physical demarcation point between carrier network loop facilities and the CI.
non-basis system: A term used in this standard to describe a loop transmission system with which DSL
systems and other new loop transmission systems are not required to demonstrate spectral compatibility.
pair: Two insulated conductors.
power spectral density (PSD): The power level and frequency content of a transmitted signal.
PSD Mask: The PSD Mask defines the absolute upper bound for spectrum management compliance.
PSD Template: The PSD Template defines a nominal signal characteristic for modeling margin of, and
crosstalk due to technologies using this PSD.
short-term stationary: A term used in this standard to describe a loop transmission system in which an
“ON” condition (in which the transmitter generates a signal) alternates with an “OFF” condition (in which
the transmitter is silent or generates only a pilot tone).
spectral compatibility: The capability of two loop transmission system technologies to coexist in the same
cable and operate satisfactorily in the presence of crosstalk noise from each other.
spectrum management: In this standard, the term refers to processes that are intended to minimize the
potential for interference and maximize the utility of the frequency spectrum of metallic loop cables.
spectrum management class: In this standard, the term refers to the classes defined in clause 5.3,
classifying the technologies in terms of their PSD. Abbreviated SM class.
transverse balance: A comparison of the voltage of a transmitted metallic or transverse signal to the
voltage of any resulting longitudinal signal. See clause 6.6.
type I PSDS: A legacy loop transmission system based on 56 kbps digital data service that uses AMI
operating at 56 kbps on two loop pairs to provide a 4-wire full-duplex digital channel. Network signaling is
accomplished using bipolar patterns that include bipolar violations. For more information, see [1].
type II PSDS: A legacy loop transmission system that functions in two modes: analog and digital. Analog
signaling is used to perform network supervisory and address signaling. The system is switched to the
digital mode after a connection is established. Type II PSDS uses Time Compression Multiplexing and
AMI operating at 144 kbps to provide a full-duplex 56 kbps service on a 2-wire loop. For more
information, see [1].
type III PSDS: A legacy loop transmission system that uses Time Compression Multiplexing and AMI
operating at 160 kbps to provide two full-duplex digital channels on a 2-wire loop. One digital channel is
an 8 kbps signaling channel for supervisory and address signaling and the other is a 64 kbps data channel.
For more information, see [1].
twisted pair: A balanced transmission line consisting of two insulated conductors that have been twisted
together during the manufacturing process to reduce coupling to and from external circuits. See balanced.
upstream: The direction of transmission from the Customer Installation to the carrier Central Office.
voicegrade: A term used to qualify a channel, facility, or service that is suitable for the transmission of
speech, data, or facsimile signals; generally with a frequency range of about 300 to 3000 Hz.
working length: The sum of all cable segment lengths from the central office to the network interface at a
customer location, excluding non-working bridged tap.
3.2 Abbreviations, acronyms, and symbols
The following acronyms are used throughout this document:
2B1Q
ADSL
AMI
ANSI
Two Binary, One Quaternary
Asymmetric Digital Subscriber Line
Alternate Mark Inversion
American National Standards Institute
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AWG
BER
BERT
BRI
CAP
CI
CO
CPE
CSA
dB
dBm
dBrn
dBrnC
DC
DDS
DFE
DLC
DMT
DSL
DUT
EC
EWL
FCC
FEXT
FSAN
HDSL
HDSL2
HDSL4
Hz
IDSL
ISDN
ITU-T
kbps
kft
kHz
LBR
L 26
mH
ms
mW
NEXT
NI
PAM
POTS
PSD
PSDS
QAM
RADSL
RLCG
RMS
RRD
6
ANSI T1.417
American Wire Gauge; see definition.
bit error ratio; see definition.
bit error ratio tester
Basic Rate ISDN
Carrierless Amplitude and Phase Modulation
customer installation; see definition.
central office; see definition.
customer premises equipment; see definition.
carrier serving area
decibel
decibel referenced to 1 milliwatt
decibel referenced to noise
decibel referenced to noise with C-message weighting
direct current; used to indicate 0 Hz frequency.
Digital Data Service
decision feedback equalizer
digital loop carrier
Discrete Multitone
digital subscriber line
device under test
echo cancelled
equivalent working length; see definition.
Federal Communications Commission
far-end crosstalk; see definition.
Full Service Access Networks
High-bit-rate Digital Subscriber Line
nd
High-bit-rate Digital Subscriber Line 2 generation
nd
4-wire High-bit-rate Digital Subscriber Line 2 generation
hertz
Integrated Digital Subscriber Line
Integrated Services Digital Network
International Telecommunication Union – Telecommunication
Standardization Sector
kilobits per second
kilofeet
kilohertz
line bit-rate
total working length of 26-AWG cable on a loop
millihenry
millisecond
milliwatt
near-end crosstalk; see definition.
network interface; see definition.
Pulse Amplitude Modulation
Plain Old Telephone Service
power spectral density; see definition.
Public Switched Digital Service. See definitions of type 1, type II,
and type III.
Quadrature Amplitude Modulation
Rate Adaptive Digital Subscriber Line
resistance, inductance, capacitance, and conductance
root mean square
revised resistance design
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RT
SDSL
SHDSL
SM
SNR
T1
TS
TU-C
TU-R
UDC
VDSL
ANSI T1.417
remote terminal
Symmetric Digital Subscriber Line
Single-pair High-speed Digital Subscriber Line
spectrum management, e.g., SM class 1. See definition.
Signal-to-Noise Ratio
type of 4-wire metallic 1.544 Mb/s transmission system
technology specific
Transceiver Unit – Central office end. Sometimes combined with
another letter; e.g., ATU-C for a central office ADSL
transceiver
Transceiver Unit – Remote terminal end. Sometimes combined with
another letter; e.g., ATU-R for a remote ADSL transceiver
Universal Digital Channel
Very-high-bit-rate Digital Subscriber Line
4. General Information
Most of the subscriber loop plant in North America consists of metallic cables that were designed primarily for
voicegrade services. Several other types of services and technologies use these loop cables however including, but
not limited to, digital data services, T1-carrier systems, and Digital Subscriber Line (DSL) transmission systems.
Metallic loop cables generally contain several solid copper conductors that are circular in cross-section. Each
conductor is surrounded by insulation that is usually color-coded. During manufacturing, pairs of insulated
conductors are twisted together. Several twisted pairs are then assembled together into units called binder groups
that are bound with colored tape for identification.
The signals that are transmitted on a loop cable pair create an electromagnetic field that surrounds nearby pairs and
induces voltages into those pairs. The twisting of the insulated conductors into pairs minimizes this coupling as
does the bundling of pairs into binder groups. Despite these measures, capacitive and inductive coupling still exists
between pairs of a multi-pair loop cable.
This clause provides general information about crosstalk interference in metallic loop cables, the spectral
compatibility of loop transmission systems, and various aspects of spectrum management. Clause 5 provides signal
power limitations and deployment guidelines for the DSL spectrum management classes. Conformance testing
methodology is provided in clause 6.
4.1 Crosstalk
The electromagnetic energy that couples into a metallic cable pair from services and transmission system
technologies on other pairs in the same cable is unwanted energy and is called crosstalk noise or simply “crosstalk”.
Crosstalk may, or may not, be disturbing. When crosstalk causes an unacceptable degradation in the performance of
victim services or technologies in the same cable, it is called crosstalk interference. Minimizing crosstalk
interference requires the careful manufacturing, installation, maintenance, and administration of loop cables.
Crosstalk is sensitive to frequency, signal strength, and exposure. High frequency energy couples into other pairs
more than low frequency energy because as the signal frequency increases, the crosstalk coupling loss between the
pairs of a cable decreases. Hence, for two signals of equal strength, the higher the frequency, the greater the
crosstalk noise.
A strong signal will transfer more power into other pairs than will a weaker signal. The amount of crosstalk noise is
directly proportional to the power of the disturbing signal. The stronger the disturbing signal, the greater the
crosstalk noise. Thus, one of the most effective means of controlling crosstalk noise is to limit the signal energy that
is applied to cable pairs. Signal power limitations for several DSL classes are provided in clause 5.
Exposure is a measure of the proximity of metallic pairs at various points along a cable and the length over which
pairs are in close proximity. The greater the exposure, the greater the total crosstalk noise. Since it is impossible to
predict the exact amount of exposure between any two pairs in a cable, statistical exposure models are used for the
crosstalk margin evaluations described in Annex A. In this standard, it is assumed that all loops in a binder are of
the same length; it is known that this assumption does not always hold in the loop plant, and where loops of different
lengths share a binder group, degradation may be more or less. For example, different loop lengths within the same
binder group can result from feeder-distribution cross-connection.
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Crosstalk noise that occurs when a receiver on a disturbed pair is located at the same end of the cable as the
transmitter of a disturbing pair is called Near-End-Crosstalk (NEXT). Crosstalk noise that occurs when a receiver
on a disturbed pair is located at the other end of the cable as the transmitter of the disturbing pair is called Far-EndCrosstalk (FEXT). NEXT coupling is generally greater than FEXT coupling when transmission takes place in both
directions in a binder and there is a frequency overlap between the upstream and downstream signals.
4.2 Spectral compatibility
In general, spectral compatibility is the capability of two loop transmission system technologies to coexist in the
same cable and operate satisfactorily in the presence of crosstalk noise from each other.
A loop transmission system technology is considered to be spectrally compatible with other loop transmission
systems when:
a) It meets the signal power limits, the deployment guidelines and other criteria for one or more of the
spectrum management classes or technologies defined in clause 5 of this standard.
Or
b) It meets the criteria of the analytical method defined in Annex A of this standard.
This standard does not explicitly define “unacceptable or significant service degradation”.
4.3 Spectrum management
In this standard, the term spectrum management refers to processes that are intended to minimize the potential for
crosstalk interference and maximize the utility of the frequency spectrum in multi-pair metallic loop cables.
The spectrum management requirements and recommendations in this standard include signal power limitations,
deployment guidelines based on equivalent working length (EWL), and a generic analytical method that can be used
to define new DSL spectrum management classifications or determine the spectral compatibility of different
technologies. The requirements and recommendations in this standard are intended to provide spectral compatibility
with certain defined basis loop transmission systems and thereby maximize the use of the bandwidth provided by
metallic loop cables.
A critical part of the management process is the equivalent working length knowledge for a given loop. Without
equivalent working length knowledge, it is not possible to determine compliance with this standard.
1
4.3.1 Basis loop systems
Basis systems are loop transmission systems with which the DSL spectrum management classes defined in this
standard and other new loop transmission systems2, are required to demonstrate spectral compatibility. The basis
systems are systems that are currently deployed.
It is not necessary, nor sufficient, for a system to be on the list of basis systems for the system to be compliant with
this standard. The list of basis systems is a living list; new systems may be added to the list, and eventually systems
may be retired from the list when the need for that system has passed its usefulness. To avoid an excessive
impediment to potential new technologies and to simplify the Spectrum Management standard, it is highly desirable
to include in the list of basis systems only those systems that have the greatest total impact on the population of
subscriber line users. To be included in the list of basis systems, the following factors shall apply:
1) It is highly preferred that the system be standardized by the ITU or an ANSI accredited standards organization
or that a draft standard is expected to be approved by the time the forthcoming issue of the Spectrum
Management standard is expected to be published. If an effort has been made to standardize the system and
there is a clear reason why the system can not be standardized, then a physical-layer specification shall be
publicly available.
2) The specification for a non-standard system shall be stable, widely accepted by most of the industry, and shall
specify all aspects necessary to determine spectral compatibility (e.g., transmitted signal PSD, modulation
method, coding, bit-rate, start-up process, and margin to be achieved for certain reference loops and reference
noise).
1
2
8
Basis systems are not defined or intended for the purpose of resolving interference disputes.
This includes high bandwidth CPE-based systems.
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3) Preferably, a new basis system should not require changes to the existing spectrum management classes
(including PSD definition, total average power, transverse balance, longitudinal output voltage and deployment
guidelines) to maintain spectral compatibility with the new basis system.
4) Preferably, a new basis system should not be adequately addressed by the existing systems on the basis system
list.
5) New basis systems should demonstrate possible scenarios where the new system could be disturbed while other
basis systems are not.
In order to assure spectral compatibility with the anticipated mix of current and future technologies on loop binder
groups, this standard has defined a set of loop transmission basis systems3 with regard to which spectral
compatibility shall be demonstrated:
- Voicegrade services4.
- Enhanced Business Services (P-Phone) based5 on NIS S102-2.
- Digital Data Service (DDS) based on T1.410.
- Basic Rate ISDN based on T1.601. Note that this includes 2-channel digital systems (UDC-2) based on BRI
technology.
- High-Bit-Rate Digital Subscriber Line (HDSL) based on G.991.1, Annex A.
- Asymmetrical Digital Subscriber Line (ADSL) based on T1.413-1998 with non-overlapped
upstream/downstream mode. (G.992.1, Annex A)
- RADSL based on Committee T1 Technical Report No. 59.
- Splitterless ADSL based on T1.419-2000.
- HDSL2 (DS1 payload on single pair) based on the T1.418.
- 2B1Q SDSL6 @ 400 kbps, 1040 kbps and 1568 kbps.
This set is defined to take into account: 1) voluntary DSL standards based on industry consensus and open
specifications and 2) several other loop transmission systems. Spectral compatibility with the basis systems listed
above shall be demonstrated by meeting all of the signal power limitations and other criteria for one of the DSL
spectrum management classes defined in clause 5 (Method A) or by meeting the criteria defined in Annex A
(Method B).
4.3.1.1 Voicegrade services
Voicegrade services include speech, network signaling, data, and tone signals that use the frequency spectrum from
0 to 4 kHz. (See Annex A.)
4.3.1.2 Enhanced Business Services (P-Phone)
Enhanced Business Services use the frequency spectrum from 0 to 10 kHz and are used to transport speech signals
in the same way as done by traditional voicegrade services. A digital signaling channel centered on 8 kHz allows
the performance of all functions associated with the setting up and tearing down of voice calls without the use of
high voltage signaling.
4.3.1.3 Digital Data Services (DDS)
Digital Data Services, based upon T1.410, operate at 64 kbps and subrates of 2.4, 4.8, 9.6, 19.2, 38.4 and 56 kbps.
Secondary channel services are also available for all subrates. While all DDS subrates and subrates with secondary
3
Very-high-speed Digital Subscriber Line (VDSL) and G.shdsl technology are currently in the standards development
process. VDSL and G.shdsl are expected to be added to the basis system list. In anticipation of this, some
accommodation of VDSL and G.shdsl has been made in the signal power limitations for the spectrum management
classes defined in clause 5.
4
Voicegrade services include speech, data, and call processing signals that use the frequency spectrum from 0 to 4
kHz. For more information, see Annex A.
5
“Meridian Business Services Network Access,” Interface Specification NIS S102-2, Issues 02.02, October 1998.
This specification may be ordered from Nortel Networks Inc.
6
It is expected that 2B1Q SDSL systems will be replaced when ITU G.shdsl systems are placed on the basis system
list.
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channels are basis systems, the DDS analytical evaluation procedure in Annex A focuses on 9.6 kbps and 64 kbps
DDS in order to reduce the number of DDS evaluations that a new system must undergo. Since Type 1 Public
Switched Digital Service (PSDS) uses the same physical layer as 56 kbps DDS, any new technology that
demonstrates compatibility with 56 kbps DDS will also be compatible with Type 1 PSDS.
4.3.1.4 Basic Rate ISDN (BRI)
In the context of this standard, BRI represents a family of basis loop transmission systems that uses the transceiver
technology described in T1.601. The family includes traditional BRI that uses the ISDN data link layer protocols
described in T1.602 as well as other systems that have adapted the T1.601 layer 1 transceiver technology for use as:
−
Packet network access system (IDSL).
− Point-to-point transport system sometimes referred to as a Universal Digital Channel (UDC).
BRI, IDSL, and UDC are defined in this standard as systems that use the 2B1Q line code, operate at 80 kbaud for
transmission at 160 kbps, and may be transported via DLC by using BRI termination extension (BRITE) devices.
The entire BRI family is a basis system. The analytical method for demonstrating compatibility with BRI in Annex
A does not differentiate between the members of the BRI family and adequately addresses all members of the
family.
4.3.1.5 High-Bit-Rate Digital Subscriber Line (HDSL)
HDSL systems are designed to transport 784 kbps over Carrier Serving Area (CSA) distances on a single non-loaded
twisted pair. The most common application transports a 1.544 Mbps payload on two non-loaded twisted pairs but
some applications may use a single pair. Some HDSL applications extend the reach by the use of intermediate
repeaters. Basis HDSL systems are echo cancelled hybrid systems that use the 2B1Q line code and operate at 392
kbaud. The analytical method for demonstrating compatibility with HDSL in Annex A does not differentiate
between one pair and two pair applications.
4.3.1.6 HDSL2
HDSL2 is a second generation HDSL loop transmission system that is standardized. The system is designed to
transport a 1.544 Mb/s payload on a single non-loaded twisted pair at CSA distances.
4.3.1.7 ADSL, RADSL, and Splitterless ADSL
The basis asymmetrical DSL systems operate using different frequency bands (non-overlapped) for upstream and
downstream operation. The analytical method for demonstrating compatibility with these systems in Annex A is
described in terms of the relevant line code (i.e., DMT or CAP/QAM).
4.3.1.8 2B1Q SDSL
2B1Q SDSL uses 4-PAM modulation. The bit rate, baud rate, and power spectral density at both TU-C and TU-R
transceivers are the same. 2B1Q SDSL systems may operate at data rates from 64 kbps to 2320 kbps, with
granularity of data rate ≥ 8 kbps. Spectral compatibility with 2B1Q SDSL at 160 kbps and 784 kbps is demonstrated
through spectral compatibility with BRI and HDSL, respectively.
4.3.2 Legacy systems
Newly deployed loop services and technologies may encounter a loop environment that includes one or more legacy
systems. A legacy system is a loop service or technology that was defined many years ago and is nearing the end of
its life cycle.
The following services and technologies are legacy systems. However, this list may not be all-inclusive:
- 15 kHz Program Audio services
- Type II PSDS
- Type III PSDS
- Local Area Data Channels
- Data-Over-Voice services and technologies
- Analog Carrier technologies
- AMI T1
The legacy systems listed above were not addressed during the development of this standard.
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4.3.3 Signal power limitations (method A)
Since stronger signals couple more crosstalk power into other pairs than weaker signals, the most widely used and
most successful method of controlling crosstalk interference and achieving spectral compatibility is through the use
of signal power limitations. Signal power limitations specify the amplitude, frequency distribution, and total power
of electrical signals at the point where the signal enters the subscriber loop cable.
For all DSL spectrum management classes addressed in this standard, clause 5 defines signal power limits. The
requirements apply to signals transmitted by DSL transceiver units whether located in a Central Office (TU-C) or a
remote terminal location (TU-R). The remote terminal location is usually on or near the customer premises.
The set of spectrum management classes is a living list; new classes may be added to the list and eventually classes
could be retired from the list when there is widespread agreement that a class is no longer desirable or useful. To
simplify the spectrum management process and this Standard, it is desirable that the number of spectrum
management classes be no larger than necessary.
A new spectrum management class may be added if the following five conditions are satisfied:
1) The new class is fully specified.
2) The new class is spectrally compatible with all basis systems, per Annex A.
3) Preferably, a new class should offer substantial benefits beyond the existing classes. For example:
a) Have a deployment guideline substantially different from existing similar classes.
b) Enable members of the class to achieve the same bit rate and loop reach as members of the existing
similar classes while reducing the SNR margin impact of crosstalk to a basis system, and causing no
more crosstalk impact to all other basis systems.
c) Meets a) or b) and replaces or modifies an existing class.
The characterization of a transmitted signal by power level and frequency content is called the power spectral
density (PSD) of the signal. The primary signal power requirements in this standard are specified through the use of
PSD masks and templates. The PSD mask shows the maximum power boundary or limit, in dBm per Hz, for the
transmitted signal. The use of the PSD masks and templates is described more fully in 6.1, 6.2, and 6.4.
4.3.3.1 Transceiver unit – remote terminal end (TU-R)
Part 68 of the FCC Rules and Regulations contain mandatory signal power limits for several types of customer
premises equipment (CPE) including voice, voiceband data, DDS subrates, PSDS, BRI, local area data channel
(LADC), and DS1. Clause 5 of this standard defines signal power limits for several DSL spectrum management
classifications that are not currently covered by Part 68.
The TU-R equipment used with DSL systems is usually CPE. However, in some cases it may be network
equipment. The TU-R signal power limits in clause 5 shall be applicable regardless of whether or not the TU-R is
network equipment or CPE. Any TU-R that transmits a signal into a metallic loop cable shall meet the relevant
upstream signal power limitations and other criteria associated with one of the DSL spectrum management
classifications defined in clause 5.3 or clause 5.4.
4.3.3.2 Transceiver unit – central office end (TU-C)
Historically, carriers have controlled the transmitted signal power of network elements through the development and
use of voluntary industry standards related to particular technologies. Clause 5 of this standard defines signal power
limits for several DSL spectrum management classifications. The DSL classifications defined in clause 5 are based
on the industry’s current view of requirements for spectrum management.
The TU-C is network equipment. Any TU-C that transmits a signal into a metallic loop cable shall meet the relevant
downstream signal power limitations and other criteria associated with one of the DSL spectrum management
classifications defined in clause 5.3 or clause 5.4.
4.3.4 Technology deployment guidelines
Some loop transmission system technologies can be deployed in a manner that substantially increases the likelihood
of crosstalk interference. To prevent interference in such instances, it is necessary to adhere to the guidelines in
clause 5.
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4.3.4.1 Deployment guidelines
Deployment guidelines constrain the way loop transmission systems are operated so that the assumptions on which
spectral compatibility was determined will remain valid. Deployment guidelines may include such things as loop
reach guidelines or prohibitions against spectrum management class 5 systems using reverse upstream/downstream
operation. If applicable, deployment guidelines shall be specified in clause 5.
4.3.4.2 Binder Group Considerations
Binder group integrity is not always maintained in the loop plant. This standard assumes collocated TU-R and that
all loops within a binder group are the same length and that loops of significant difference in length will be in
separate binder groups. However, technologies that demonstrate spectral compatibility by using the analytical
method in Annex A (Method B) shall not rely upon administrative binder group separation between services or
technologies in order to achieve full compatibility with any basis transmission system. It should be noted that this
standard does not preclude the use of binder group separation.
4.3.5 Analytical method of determining spectral compatibility (method B)
It is recognized that future technologies may transmit signals that do not conform to the signal power specifications
for one of the spectrum management classes defined in clause 5.3, but which might still be spectrally compatible
with the basis systems listed in clause 4.3.1. In order to nurture innovation in the development of new technologies
which further maximize the utility of the copper loop plant, an analytical method for evaluating new technologies is
provided in Annex A.
This method (referred to as Method B) involves the computation of signal to noise margins for basis systems, and
provides an industry-approved method of determining the spectral compatibility of any loop transmission system
with the basis loop transmission systems defined in this standard. For each of these basis systems, Annex A
provides the specific NEXT margin formulas, evaluation loops, and defined crosstalk environments required by
Method B.
The analytical method in Annex A should be used to develop new signal power limits and deployment guidelines for
new DSL spectrum management classes. It is expected that this analytical method will also be used to provide
guidance during new system development. However, as noted in 4.3, such use could lead to the introduction of
several new technologies that would be compatible with basis systems but not necessarily compatible with each
other. Therefore, system developers are encouraged to bring new DSL technologies that do not fit into existing
spectrum management classes into the formulating group for this standard, so that the creation of a new class and
any associated deployment guidelines can be considered. Other processes, such as the disclosure of verifiable
methods to assess spectral compatibility with the new technology, may also help avoid the uncoordinated
introduction of new technologies that could result in crosstalk interference.
However, it is also recognized that the spectrum management classes represent an overly conservative estimation of
many actual loop transmission systems (e.g., PAM systems fitting within the limits of the masks defined in clause
5.3). In such cases, the techniques of Method B (Annex A) may be used to derive the technology-specific guidelines
found in clause 5.4. As these calculations represent a more accurate indication of the spectral compatibility of these
technologies, their use and derived deployment guidelines may be used instead of the equivalent spectrum
management class.
The telephone loop plant consists of 12, 13, 25, 50, and 100 pair binder group cables. This standard employs a 50
pair binder group model for the analysis of spectral compatibility.
4.3.5.1 Margin computations
Margin computations determine the crosstalk margin in decibels (dB). Each basis system should have the margin
specified for that system in Annex A. The margin shall be calculated with BER ≤ 10 –7 unless otherwise specified.
Margin is a function of many variables including:
a) Crosstalk coupling loss,
b) Loss characteristics of loop cables,
c) Characteristics of the disturbed signal,
d) Receiver technology of the disturbed system, and
e) Characteristics of the disturber signal.
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Annex A provides the margin formula and the information associated with item a) thorough d). The user will have
to supply the information for item e).
4.3.5.2 Evaluation loops
For each basis system, Annex A provides a set of loops that shall be used for analytical evaluations.
5. Signal power limits and other criteria
Crosstalk noise is controlled primarily through the use of signal power limits that consist of PSD limitations and
total average power limitations. Additional criteria, such as transverse balance requirements and deployment
guidelines, are also important. This clause provides all of these specifications for the DSL spectrum management
classes. The conformance testing methodology in clause 6 shall be used to determine compliance with the
requirements in this clause.
DSL transmission systems that meet the PSD limitations, total average power limitations, longitudinal output
voltage and transverse balance requirements for one of the DSL spectrum management classes defined in this clause
shall be considered spectrally compatible with all basis systems if they are deployed according to the applicable
deployment guidelines that are specified in this clause or determined by Annex A. The deployment guidelines limit
the distance that a system can operate so that crosstalk from that system will not impair the basis systems.
A multi-rate DSL system shall be considered spectrally compatible if it is deployed according to the applicable
deployment guidelines associated with the class or data rate for which it is configured.
5.1 Short-term stationary systems
Some types of DSL transmitters operate in transmission modes in which an “ON” condition (in which the
transmitter generates a signal) alternates with an “OFF” condition (in which the transmitter is silent or generates
only a pilot tone). Examples of such transmitters include burst transmission systems and systems that use quiescent
modes to reduce power consumption during idle data periods. Such transmitters are referred to as “short-term
stationary,” since during the ON condition the transmitted signal has the same effect as a stationary (or cyclostationary) signal when observed over an appropriately short time interval. Due to the relative frequency of
ON/OFF and OFF/ON transitions in short-term stationary transmitters, additional conformance criteria are applied
to these transmitters.
Clause 6.5 defines a test to determine whether short-term stationary conformance criteria shall be applied to a
system and defines the short-term stationary conformance criteria.
5.2 Spectral compatibility of systems using intermediate TU devices
It is recognized that systems with intermediate TU devices between the CO and CI can be deployed in a manner that
substantially increases the likelihood of crosstalk interference to the basis systems. Annex L contains models that
may be used to estimate the levels of interference. Annex G contains some calculated results of a model in which
the NEXT / FEXT effects of the intermediate device are not considered. It is expected that Issue 2 of this standard
will address this topic.
5.3 Spectrum management classes
5.3.1 Spectrum management class 1
Spectrum management class 1 is intended for DSL transmission systems that operate in the frequency spectrum up
to about 115 kHz, including most, but not all, T1.601 compliant systems.
5.3.1.1 Spectrum management class 1 PSD and total average power limitation
Spectrum management class 1 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using the PSD template described in Table 1 and Figure 1. The total average power into 135 Ω and below 115 kHz
that is transmitted by the spectrum management class 1 TU-C and TU-R equipment shall be 14.0 dBm or less.
5.3.1.2 Spectrum management class 1 transverse balance requirement
The transverse balance of spectrum management class 1 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 1 TU-C and TU-R equipment shall equal or exceed the values in
This is a draft document and thus, is still dynamic in nature.
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Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 1
PSD template defined in Table 1.
5.3.1.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 1 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 1 PSD template defined in Table 1. There is no requirement for frequencies below the
operating band.
5.3.1.4 Spectrum management class 1 deployment guidelines
Loop transmission systems that meet the signal power and transverse balance requirements associated with spectrum
management class 1 will be considered spectrally compatible on any non-loaded loop facility. The provisions of
clause 5.2 do not apply to spectrum management class 1.
5.3.2 Spectrum management class 2
Spectrum management class 2 is intended for DSL transmission systems that operate in the frequency spectrum
from 0 to about 238 kHz.
5.3.2.1 Spectrum management class 2 PSD and total average power limitation
Spectrum management class 2 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using the PSD template described in Table 3 and Figure 2. At frequencies at or below 500 kHz, linear interpolation
of the frequency and PSD table entries is used to define the template.
The total average power below 238 kHz that is transmitted by spectrum management class 2 TU-C and TU-R
equipment shall be 14.0 dBm or less.
5.3.2.2 Spectrum management class 2 transverse balance requirement
The transverse balance of spectrum management class 2 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 2 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 2
PSD template defined in Table 3.
5.3.2.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 2 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 2 PSD template defined in Table 3. There is no requirement for frequencies below the
operating band.
5.3.2.4 Spectrum management class 2 deployment guidelines
Non-repeatered spectrum management class 2 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 2 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 11.5 kilofeet or less. Repeatered systems are
addressed per the provisions of clause 5.2.
5.3.3 Spectrum management class 3
Spectrum management class 3 is intended for DSL transmission systems that operate in the frequency spectrum up
to about 370 kHz.
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5.3.3.1 Spectrum management class 3 PSD and total average power limitation
Spectrum management class 3 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using the PSD template described in Table 4 and Figure 3. At frequencies at or below 1.05 MHz, linear
interpolation of the frequency and PSD table entries is used to define the template.
The total average power below 370 kHz that is transmitted by spectrum management class 3 TU-C and TU-R
equipment shall be 14.0 dBm or less.
5.3.3.2 Spectrum management class 3 transverse balance requirement
The transverse balance of spectrum management class 3 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 3 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 3
PSD template defined in Table 4.
5.3.3.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 3 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 3 PSD template defined in Table 4. There is no requirement for frequencies below the
operating band.
5.3.3.4 Spectrum management class 3 deployment guidelines
Non-repeatered spectrum management class 3 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 3 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 9 kilofeet or less. Repeatered systems are addressed
per the provisions of clause 5.2.
5.3.4 Spectrum management class 4
Spectrum management class 4 is intended to include standard compliant HDSL2 equipment (T1.418) and other DSL
transmission systems that have TU-C equipment that operates in the frequency spectrum up to about 440 kHz and
TU-R equipment that operates in the frequency spectrum up to about 300 kHz.
5.3.4.1 Spectrum management class 4 PSD and total average power limitation
Spectrum management class 4 TU-C equipment shall meet the PSD conformance criteria in clause 6.2.4 using the
downstream PSD mask described in Table 5 and Figure 4. At frequencies at or below 3 MHz, linear interpolation of
the frequency and PSD table entries is used to define the template.
Spectrum management class 4 TU-R equipment shall meet the PSD conformance criteria in clause 6.2.4 using the
upstream PSD mask described in Table 6 and Figure 5. At frequencies at or below 1.4 MHz, linear interpolation of
the frequency and PSD table entries is used to define the template.
The total average downstream power (into 135 Ω) below 450 kHz that is transmitted by the spectrum management
class 4 TU-C shall not exceed 17.3 dBm. The total average upstream power (into 135 Ω) below 350 kHz that is
transmitted by the spectrum management class 4 TU-R shall not exceed 17.0 dBm.
5.3.4.2 Spectrum management class 4 transverse balance requirement
The transverse balance of spectrum management class 4 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 4 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 4
PSD masks defined in Table 5 and Table 6.
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5.3.4.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 4 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 4 PSD masks defined in Table 5 and Table 6. There is no requirement for frequencies
below the operating band.
5.3.4.4 Spectrum management class 4 deployment guidelines
Non-repeatered spectrum management class 4 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 4 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 10.5 kilofeet 7 or less, but are not spectrally
compatible in the following mode:
−
SM 4 TU-C transceivers located at the customer end of the loop transmitting in the downstream frequency band
up to 440 kHz. This does not preclude configurations involving carrier TU-C located adjacent to the CO.
Repeatered systems are addressed per the provisions of clause 5.2.
5.3.5 Spectrum management class 5
Spectrum management class 5 is intended for DSL transmission systems that have TU-C equipment that operates in
the frequency spectrum from about 25 kHz to about 1104 kHz and TU-R equipment that operates in the frequency
spectrum from about 25 kHz to about 138 kHz.
5.3.5.1 Spectrum management class 5 PSD and total average power limitation
Spectrum management class 5 TU-C equipment shall meet the PSD conformance criteria in clause 6.1 using the
downstream PSD template defined in Table 7 and Figure 6.
Spectrum management class 5 TU-R equipment shall meet the PSD conformance criteria in clause 6.1 using the
upstream PSD template defined in Table 11 and Figure 7.
The total average downstream power between 25 kHz and 1104 kHz that is transmitted by the spectrum
management class 5 TU-C shall not exceed 20.9 dBm.
The total average upstream power below 138 kHz that is transmitted by the spectrum management class 5 TU-R
shall not exceed 13 dBm.
5.3.5.2 Spectrum management class 5 transverse balance requirement
The transverse balance of spectrum management class 5 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 100 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 5 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 5
PSD templates defined in Table 7 and Table 11.
5.3.5.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 5 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 5 PSD templates defined in Table 7 and Table 11. There is no requirement for
frequencies below the operating band.
7
See T1.418 for HDSL2 performance requirements.
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5.3.5.4 Spectrum management class 5 deployment guidelines
Non-repeatered spectrum management class 5 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 5 will be considered spectrally compatible on any
non-loaded loop facility, but are not spectrally compatible in the following modes:
−
Power boost mode described in the first version of the ADSL standard (T1.413-1995).
−
Transceivers located at the customer end of the loop transmitting in the downstream frequency band (138 - 1104
kHz). This does not preclude configurations involving carrier TU-C located adjacent to the CO.
Repeatered systems are addressed per the provisions of clause 5.2.
5.3.6 Spectrum management class 6
Spectrum management class 6 is intended for DSL transmission systems that operate in the frequency spectrum up
to about 10 - 20 MHz.
5.3.6.1 Spectrum management class 6 PSD and total average power limitation
Spectrum management class 6 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using a PSD template (or templates) that shall address both CO and remote deployments. The spectrum
management class 6 PSD template should be based on emerging VDSL standards, which were not completed in time
for this issue of this standard. Spectrum management class 6 should be frequency-division duplex (FDD), with
distinct PSD templates for upstream and downstream transmission.
5.3.6.2 Spectrum management class 6 transverse balance requirement
The transverse balance of spectrum management class 6 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 100 Ω measurement configuration specified in clause 6.6.
The transverse balance of spectrum management class 6 TU-C and TU-R equipment shall equal or exceed the values
in Table 2 over the entire range of frequencies between the upper and lower -20 dB points (relative to peak PSD) of
the signal passband. Above 3 MHz, the transverse balance requirements shall address both CO and remote
deployments.
5.3.6.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 6 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband. There is no requirement
for frequencies below the operating band.
5.3.6.4 Spectrum management class 6 deployment guidelines
Spectrum management class 6 DSL transmission systems shall use non-loaded loop facilities. Unlike other DSL
systems, spectrum management class 6 systems were created to offer high bit rates over short ranges when deployed
from remote fiber-fed terminals, pedestals, or cases. Deployment guidelines for spectrum management class 6
systems shall address both CO and remote deployments.
5.3.7 Spectrum management class 7
Spectrum management class 7 is intended for DSL transmission systems that operate in the frequency spectrum
from 0 to about 776 kHz.
5.3.7.1 Spectrum management class 7 PSD and total average power limitation
Spectrum management class 7 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using the PSD template described in Table 8 and Figure 8. At frequencies at or below 2 MHz, linear interpolation of
the frequency and PSD table entries is used to define the template.
The total average power below 776 kHz that is transmitted by spectrum management class 7 TU-C and TU-R
equipment shall be 14.0 dBm or less.
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5.3.7.2 Spectrum management class 7 transverse balance requirement
The transverse balance of spectrum management class 7 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 7 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 7
PSD template defined in Table 8.
5.3.7.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 7 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 7 PSD template defined in Table 8. There is no requirement for frequencies below the
operating band.
5.3.7.4 Spectrum management class 7 deployment guidelines
Non-repeatered spectrum management class 7 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 7 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 6.5 kilofeet or less. Repeatered systems are addressed
per the provisions of clause 5.2.
5.3.8 Spectrum management class 8
Spectrum management class 8 is intended for DSL transmission systems that operate in the frequency spectrum
from 0 to about 584 kHz.
5.3.8.1 Spectrum management class 8 PSD and total average power limitation
Spectrum management class 8 TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.1
using the PSD template described in Table 9 and Figure 9. At frequencies at or below 2 MHz, linear interpolation of
the frequency and PSD table entries is used to define the template.
The total average power below 584 kHz that is transmitted by the spectrum management class 8 TU-C and TU-R
equipment shall be 14.0 dBm or less.
5.3.8.2 Spectrum management class 8 transverse balance requirement.
The transverse balance of spectrum management class 8 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 8 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 8
PSD template defined in Table 9.
5.3.8.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 8 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 8 PSD template defined in Table 9. There is no requirement for frequencies below the
operating band.
5.3.8.4 Spectrum management class 8 deployment guidelines
Non-repeatered spectrum management class 8 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 8 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 7.5 kilofeet or less. Repeatered systems are addressed
per the provisions of clause 5.2.
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5.3.9 Spectrum management class 9
Spectrum management class 9 is intended for DSL transmission systems that have TU-C equipment that operates in
the frequency spectrum from about 25 kHz to about 1104 kHz and TU-R equipment that operates in the frequency
spectrum from about 25 kHz to about 138 kHz.
5.3.9.1 Spectrum management class 9 PSD and total average power limitation
Spectrum management class 9 TU-C equipment shall meet the PSD conformance criteria in clause 6.1 using the
downstream PSD template defined in Table 10 and Figure 10.
Spectrum management class 9 TU-R equipment shall meet the PSD conformance criteria in clause 6.1 using the
upstream PSD template defined in Table 11 and Figure 7.
The total average downstream power between 25 kHz and 1104 kHz that is transmitted by the spectrum
management class 9 TU-C shall not exceed 20.9 dBm.
The total average upstream power below 138 kHz that is transmitted by the spectrum management class 9 TU-R
shall not exceed 13 dBm.
5.3.9.2 Spectrum management class 9 transverse balance requirement
The transverse balance of spectrum management class 9 TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and 100 Ω measurement configuration specified in clause 6.6. The
transverse balance of spectrum management class 9 TU-C and TU-R equipment shall equal or exceed the values in
Table 2 over the entire range of frequencies between the upper and lower -20 dB points of the signal passband. The
upper and lower -20 dB points (relative to peak PSD) can be determined from the spectrum management class 9
PSD templates defined in Table 10 and Table 11.
5.3.9.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 9 TU-C and TU-R equipment shall be measured over
the applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
spectrum management class 9 PSD templates defined in Table 10 and Table 11. There is no requirement for
frequencies below the operating band.
5.3.9.4 Spectrum management class 9 deployment guidelines
Non-repeatered spectrum management class 9 loop transmission systems that meet the signal power and transverse
balance requirements associated with spectrum management class 9 will be considered spectrally compatible on any
non-loaded loop facility with an equivalent working length of 13.5 kilofeet or less, but are not spectrally compatible
in the following modes:
−
Power boost mode described in the first version of the ADSL standard (T1.413-1995).
−
Transceivers located at the customer end of the loop transmitting in the downstream frequency band (138 - 1104
kHz). This does not preclude configurations involving carrier TU-C located adjacent to the CO.
Repeatered systems are addressed per the provisions of clause 5.2.
5.4 Technology Specific Guidelines
Systems as defined in this clause are spectrally compatible with all basis systems when deployed in accordance with
the associated deployment guidelines. This spectral compatibility has been determined by the calculations defined
in Annex A of this standard.
5.4.1 2B1Q SDSL
In this standard, the term 2B1Q SDSL applies to systems that have characteristics as defined in this clause. 2B1Q
SDSL transmits the same data rate in the upstream and downstream directions and the same transmitted PSD in the
upstream and downstream directions. 2B1Q SDSL transmits baseband 4-level signals during normal operation.
2B1Q SDSL systems operating at different bit rates have different transmitted PSD. Spectral compatibility of nonrepeatered 2B1Q SDSL with all basis systems is determined in this standard by using the 2B1Q SDSL transmitted
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PSD model defined by the equation for SDSLu ( f ) as a function of the symbol rate, fsym , which is equal to onehalf of the line bit rate u :
2
 π f

sin

f sym  
2.7 × 2.7
1


 ×
SDSLu ( f ) =
8
135 × f sym  π f



f sym 



f
 1+ 

 240 f 

sym 
 392

Watts / Hz .
Non-repeatered 2B1Q SDSL systems transmitting line bit rates up to 2320 kbps are spectrally compatible with all
basis systems if they meet the conditions in this clause.
5.4.1.1 2B1Q SDSL PSD and total average power limitation
2B1Q SDSL TU-C and TU-R equipment shall meet the PSD conformance criteria in clause 6.3.1. 2B1Q SDSL
transmitting no more than 1568 kbps shall conform, using the general conformance criteria in clause 6.1, with the
spectrum management class template that is determined in Table A. 9 by the transmitted 2B1Q SDSL line bit rate.
The total average power transmitted by 2B1Q SDSL TU-C and TU-R equipment in frequencies below a frequency
equal to the bit rate of 2B1Q SDSL shall be 14.0 dBm or less when measured with a 135 Ω termination impedance.
The measurement specification for 2B1Q SDSL systems transmitting no more than 1568 kbps shall comply with the
specification for the spectrum management class determined by the transmitted 2B1Q SDSL line bit rate using Table
A. 9. The measurement specification for 2B1Q SDSL systems transmitting more than 1568 kbps shall comply with
the specification used for spectrum management class 7.
5.4.1.2 2B1Q SDSL transverse balance requirement
The transverse balance of 2B1Q SDSL TU-C and TU-R equipment shall be measured over the applicable frequency
range using the procedures and 135 Ω measurement configuration specified in clause 6.6. The transverse balance
of 2B1Q SDSL TU-C and TU-R equipment shall equal or exceed the values in Table 2 over the entire range of
frequencies between the upper and lower -20 dB points of the signal passband. The upper and lower -20 dB points
(relative to peak PSD) can be determined from the 2B1Q SDSL PSD template defined in clause 5.4.1.
5.4.1.3 2B1Q SDSL Longitudinal Output Voltage
The longitudinal output voltage of 2B1Q SDSL TU-C and TU-R equipment shall be measured over the applicable
frequency range using the procedures and measurement configuration specified in clause 6.7. The longitudinal
output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table 22 over the
indicated range of frequencies. For this requirement, the operating band is the range of frequencies between the
upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the SDSL PSD
template defined in clause 5.4.1.
5.4.1.4 2B1Q SDSL deployment guidelines
Spectrally compatible, non-repeatered 2B1Q SDSL systems are deployed on non-loaded loops with working length
no greater than the deployment guidelines defined in Table 12 while transmitting both upstream and downstream
gross line bit rates that are at or below the line bit rate defined in the same row of Table 12.
Repeatered 2B1Q SDSL systems are addressed per the provisions of clause 5.2.
5.4.2 G.991.2, Annex A, Symmetric PSDs (G.shdsl)
ITU recommendation G.991.2, hereafter referred to as G.shdsl, defines symmetric PSDs as a function of multiple
symmetric data rates.8 The modulation scheme used is 16 level Trellis Coded Pulse Amplitude Modulation.
8
The term G.shdsl is used here to refer to equipment that meets the requirements of this clause. Since some
requirements of the spectrum management standard are not included in the G.991.2 specification (i.e. transverse
balance), it is possible that a G.991.2 compliant system will not meet the requirements of this clause.
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Spectral compatibility of G.shdsl systems using the symmetric PSDs of G.991.2 Annex A with all basis systems is
determined in this standard based on the following PSD template:
2

  πf  


sin

K
f
  sym 
1
1
f2

SHDSL
×
×
×
×
,

2
12
SHDSLT ( f ) =  135
f sym
f 2 + f c2
 πf 
 f 



1 + 

 f sym 
f 3dB 





−4
−1.5
0.5683 × 10 × f
, f int ≤ f ≤ 1.1MHz



f < f int 
.




f int is the frequency where the two functions governing SHDSLT ( f ) intersect in the range 10 kHz to f sym .
K SHDSL , f sym , f 3dB and the line bit rate LBR are defined in Table 13. f c is the transformer cutoff frequency,
and is set to 5 kHz for the purposes of spectral compatibility calculations.
5.4.2.1 G.shdsl PSD and total average power limitation
G.shdsl TU-C or TU-R equipment shall meet the PSD conformance criteria in clause 6.3.2.
The total average power below f sym into 135 Ω that is transmitted by the G.shdsl TU-C or TU-R shall not exceed
14 dBm.
5.4.2.2 Transverse balance requirement
The transverse balance of G.shdsl TU-C and TU-R equipment shall be measured over the applicable frequency
range using the procedures and 135 Ω measurement configuration specified in clause 6.6, with Z L = 500 Ω below
12 kHz and 90 Ω above 12 kHz, Z M = 135 Ω and V M = 0.367 volts. The transverse balance of G.shdsl
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and lower -20
dB points of the signal passband. The upper and lower -20 dB points (relative to peak PSD) can be determined from
the PSD template defined in clause 5.4.2.
5.4.2.3 Longitudinal Output Voltage
The longitudinal output voltage G.shdsl TU-C and TU-R equipment shall be measured over the applicable frequency
range using the procedures and measurement configuration specified in clause 6.7. The longitudinal output voltage
in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table 22 over the indicated
range of frequencies. For this requirement, the operating band is the range of frequencies between the upper and
lower -30 dB points (relative to peak PSD) of the signal passband as determined from the PSD template defined in
clause 5.4.2. There is no requirement for frequencies below the operating band.
5.4.2.4 G.shdsl deployment guidelines
The deployment guidelines of G.shdsl are defined in Table 14.
Repeatered G.shdsl systems are addressed per the provision of clause 5.2.
5.4.3 TC-PAM 776/784 kbps Asymmetric PSD (HDSL4 technology)
HDSL4 technology is intended to include standard compliant G.shdsl 768/776 kbps service with asymmetric PSDs
and other DSL transmission systems that have TU-C equipment operating in the frequency spectrum up to about 400
kHz and TU-R equipment operating in the frequency spectrum up to about 130 kHz.
5.4.3.1 PSD and total average power limitation
HDSL4 technology TU-C equipment shall meet the PSD conformance criteria in clause 6.3.3 using the downstream
PSD template described in Table 15 and Figure 11. At frequencies at or below 3 MHz, linear interpolation of the
frequency and PSD table entries is used to define the template.
HDSL4 technology TU-R equipment shall meet the PSD conformance criteria in clause 6.3.3 using the upstream
PSD template described in Table 16 and Figure 12.
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The total average downstream power (into 135 Ω) below 600 kHz that is transmitted by the HDSL4 technology TUC shall not exceed 14.6 dBm. The total average upstream power (into 135 Ω) below 307 kHz that is transmitted by
HDSL4 technology TU-R shall not exceed 14.6 dBm.
5.4.3.2 Transverse balance requirement
The transverse balance of HDSL4 technology TU-C and TU-R equipment shall be measured over the applicable
frequency range using the procedures and 135 Ω measurement configuration specified in clause 6.6, with ZL = 500
Ω below 12 kHz and 90 Ω above 12 kHz, ZM = 135 Ω and VM = 0.367 volts. The transverse balance of HDSL4
technology equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower -20 dB points of the signal passband. The upper and lower -20 dB points (relative to peak PSD) can be
determined from the HDSL4 technology PSD templates defined in Table 15 and Table 16.
5.4.3.3 Longitudinal Output Voltage
The longitudinal output voltage of HDSL4 technology TU-C and TU-R equipment shall be measured over the
applicable frequency range using the procedures and measurement configuration specified in clause 6.7. The
longitudinal output voltage in all 4 kHz frequency bands averaged over 1 second shall not exceed the values in Table
22 over the indicated range of frequencies. For this requirement, the operating band is the range of frequencies
between the upper and lower -30 dB points (relative to peak PSD) of the signal passband as determined from the
HDSL4 technology PSD templates defined in Table 15 and Table 16. There is no requirement for frequencies below
the operating band.
5.4.3.4 Deployment guidelines
Non-repeatered HDSL4 technology loop transmission systems that meet the signal power and transverse balance
requirements associated HDSL4 technology will be considered spectrally compatible on any non-loaded loop
facility, but are not spectrally compatible in the following mode:
−
HDSL4 technology TU-C transceivers located at the customer end of the loop transmitting in the downstream
frequency band. This does not preclude configurations involving carrier TU-C located adjacent to the CO.
Repeatered systems are addressed per the provisions of clause 5.2.
6. Conformance testing methodology
The conformance testing methodology in this clause shall be used to determine compliance with the signal power
limitations, transverse balance and longitudinal output voltage requirements in clause 5.
6.1 General conformance criteria 9
The conformance testing methodology is designed for the purposes of lab evaluation of equipment compliance to the
spectrum management classes defined in clause 5.3. It consists of three parts: a peak PSD constraint, an average
PSD constraint, and a total power constraint. As explained in clause 6.4 and Table 18, Table 19 and Table 20, the
PSD is defined at a number of discrete points with resolution bandwidth as defined for each spectrum management
class and each frequency. Let the PSD template of a spectrum management class be denoted as PT ( n ) in units of
dBm/Hz, where 1 ≤ n ≤ N , let fr ( n ) denote the center frequency in kHz at which PT ( n ) is defined, and let fr ( N )
be equal to the highest frequency to be measured, FH = 30 MHz. The template used for conformance is the
applicable template from clause 5.3 (for frequencies below 10 kHz) or the maximum of the applicable template from
clause 5.3 and –110 dBm/Hz (for frequencies above 10 kHz). Therefore we define
PT (n ) =
where
PT′ (n ) dBm / Hz,
max (PT′ (n ), − 110) dBm / Hz,
fr (n ) ≤ 10kHz
fr (n ) > 10kHz
,
PT′ (n ) is the applicable template from clause 5.3.
At frequencies where
PT′ (n ) is not defined, PT (n ) = -110 dBm/Hz. The points of PT ( n ) are in order of
increasing frequency so that fr ( n ) monotonically increases with n . The resolution bandwidth is a function of the
9
Additional rationale underpinning the development of this conformance criteria is given in Annex M.
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ANSI T1.417
spectrum management class and the frequency, and is denoted as BW ( n ) kHz at frequency fr ( n ) kHz for point n
as defined in clause 6.4 and Table 18, Table 19 and Table 20.
The PSD mask associated with a spectrum management class is denoted as PM ( n ) dBm/Hz, and unless otherwise
stated, the PSD mask is equal to the PSD template plus 3.5 dB. Above 10 kHz a minimum value of -90 dBm/Hz is
applied, so that
PM (n ) =
PT (n ) + 3.5 dBm / Hz,
fr (n ) ≤ 10kHz
max (PT (n ) + 3.5, − 90 ) dBm / Hz, fr (n ) > 10kHz
.
The first step in the testing process is measurement of the transmitted PSD of the equipment under test, which is
done with the procedure described in clause 6.4. The result of the PSD measurement, in units of dBm/Hz and at a
center frequency of fr ( n ) kHz is denoted by Pa (n ) , and is recorded with resolution bandwidth BW ( n ) kHz as
defined for the appropriate spectrum management class in clause 6.4 and Table 18, Table 19 and Table 20.
PSD conformance is achieved by meeting all the following conditions:
a)
For all n such that 1 ≤ n ≤ N , Pa (n ) ≤ PM (n ) , where PM (n ) is the PSD mask at frequency fr ( n ) as defined
above.
b) For all integers m such that 1 ≤ m and M ≤ N (i.e., for all possible WS kHz sliding windows):
 M
10 Pa (n ) 10
 ∑ BW ( n ) P (n ) 10
10 T

10 × log10  n = m M

∑ BW (n)
n=m




 ≤ 1 dB ,


where M is the maximum integer such that fr (M ) < fr (m ) + WS kHz (the inequality is strict, so if
BW (m ) = WS kHz , then M = m ), and WS is the width of the sliding window in kHz.
This equation can be described in words as follows: the PSD power of each measured point in mW is divided by
the PSD template in mW at that point; then summed and averaged over a left-aligned bandwidth that is as close
as possible to the window width WS kHz; the result must be less than or equal to 1 dB in all WS kHz sliding
windows.
The value of WS is 100 kHz where fr ( m ) ≤ f1 − 100 kHz and 1 MHz where fr ( m ) > f1 − 100 kHz.
Frequency f1 is the greater of either 1.1 MHz or the point where PT ( n ) drops below -93.5 dBm/Hz and never
rises above it: i.e.,
f1 is the lowest frequency fr (m) such that PT ( n ) < -93.5 dBm/Hz for all n ≥ m .
c)
The total power of the transmitted PSD shall be no greater than the total power limit for that spectrum
management class, as defined in clause 5.3.
The transverse balance of the associated TU-C and TU-R shall be greater than or equal to the requirement for that
spectrum management class, as defined in clause 5.3.
The longitudinal output voltage of the associated TU-C and TU-R shall not exceed the requirement for that spectrum
management class, as defined in clause 5.3.
6.2 PSD conformance criteria unique to spectrum management classes
6.2.1 Specific conformance criteria for spectrum management class 1
The criteria of clause 6.1 are to be utilized.
6.2.2 Specific conformance criteria for spectrum management class 2
The criteria of clause 6.1 are to be utilized.
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ANSI T1.417
6.2.3 Specific conformance criteria for spectrum management class 3
The criteria of clause 6.1 are to be utilized.
6.2.4 Specific conformance criteria for spectrum management class 4
The general PSD conformance criteria in clause 6.1 do not apply to spectrum management class 4 at frequencies
below 3.1 MHz. A PSD mask is specified for spectrum management class 4 at these frequencies instead of a PSD
template. The PSD templates for the TU-C and TU-R are the associated PSD masks reduced by 1 dB. At
frequencies below 3.1 MHz, a member of spectrum management class 4 shall have a measured PSD that shall not
exceed the PSD mask that is specified for spectrum management class 4 in Table 5 and Table 6 and Figure 4 and
Figure 5 at any frequency. For frequencies between 3.1 and 30 MHz, the general conformance requirements from
clause 6.1 do apply, with PT ( n ) = –110 dBm/Hz and PM ( n ) = -90 dBm/Hz. A member of spectrum management
class 4 shall also meet the total average power limitations, transverse balance requirement, longitudinal output
voltage requirement and deployment guidelines defined in clause 5.3.4 of this standard as well as meeting all other
applicable requirements in this standard.
6.2.5 Specific conformance criteria for spectrum management class 5
Since equipment conforming to T1.413-1998, ITU-T Recommendation G.992.1 or ITU-T Recommendation
G.992.2, cannot transmit its maximum PSD level at all frequencies without exceeding its maximum power
constraint, equipment compliant with these standards shall be tested at its maximum total average power as limited
by clause 5.3.5.1 and with equal gains on all DMT subcarriers used by the equipment. A member of spectrum
management class 5 shall also meet the total average power limitations, transverse balance requirement, longitudinal
output voltage requirement and deployment guidelines defined in clause 5.3.5 of this standard as well as meeting all
other applicable requirements in this standard. Non-standard DMT systems and non-DMT systems must comply
with the general conformance criteria under all operating conditions.
6.2.6 Specific conformance criteria for spectrum management class 6
The criteria of clause 6.1 are to be utilized.
6.2.7 Specific conformance criteria for spectrum management class 7
The criteria of clause 6.1 are to be utilized.
6.2.8 Specific conformance criteria for spectrum management class 8
The criteria of clause 6.1 are to be utilized.
6.2.9 Specific conformance criteria for spectrum management class 9
Since equipment conforming to T1.413-1998, ITU-T Recommendation G.992.1 or ITU-T Recommendation
G.992.2, cannot transmit its maximum PSD level at all frequencies without exceeding its maximum power
constraint, equipment compliant with these standards shall be tested at maximum total average power as limited by
clause 5.3.9.1 and with equal gains on all DMT sub-carriers used by the equipment. A member of spectrum
management class 9 shall also meet the total average power limitations, transverse balance requirement, longitudinal
output voltage requirement and deployment guidelines defined in clause 5.3.9 of this standard as well as meeting all
other applicable requirements in this standard.
6.3 Technology Specific PSD conformance criteria
6.3.1 Specific conformance criteria for 2B1Q SDSL
1. The general conformance criteria in Section 6.1 is applied using the PSD template defined by the equation for
SDSLu ( f ) in clause 5.4.1 for all frequencies at or below 0.8 × f sym , where f sym is the symbol rate which
equals one-half of the bit rate. The highest frequency band over which the sliding-window PSD conformance
criterion in clause 6.1 b) is computed is from 0.8 × f sym – (Sliding-Window Bandwidth) to 0.8 × f sym . The
sliding-window bandwidth is defined as
f sym 2 instead of 100 kHz as listed in clause 6.1 b) and shall be used
for all possible 2B1Q SDSL sliding-window measurements.
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2.
For all frequencies
ANSI T1.417
f above 0.8 × f sym and at or below 6.0 × f sym , the measured PSD shall be no greater
than 6 dB above the maximum of:
3.
−
The PSD template defined by the equation for SDSLu ( f ) in clause 5.4.1, and
−
 (1000 × f )3 2
The noise floor defined as -143 - 10 log 10 
 1.134 × 1013

For all frequencies
f above 6.0 × f sym , the measured PSD shall be no greater than –90 dBm/Hz with
maximum average power in the
4.

 dBm/Hz with f in kHz.


[f ,
f + 1 MHz ] window of -50 dBm.
The measured PSD shall exhibit a null at each integer multiple of the symbol frequency. The nulls shall be at
least 50 dB below the peak of the previous main or side lobe, or below the maximum of:
−
 (1000 × f )3 2
-143 - 10 log 10 
 1.134 × 1013


 dBm/Hz with f in kHz, and


−
-90 dBm/Hz.
5. The 2B1Q SDSL PSD shall be measured with the procedure specified in clause 6.4 while terminated in 135
Ohms, and using the resolution bandwidths specified in Table 18.
These conformance criteria shall not be used to set a precedent for the conformance criteria of other technologies.
6.3.2 Specific conformance criteria for G.shdsl (Annex A, Symmetric PSDs)
At frequencies below 1.1 MHz, the general conformance criteria from clause 6.1 do not apply. Equipment that is
classified as G.shdsl according to clause 5.4.2 shall have a measured PSD that shall not exceed the following PSD
mask ( SHDSLM ( f ) ):
2

  πf  


sin 

  fsym  
K
1
1
SHDSL

×
×
×
2
fsym
SHDSLM ( f ) =  135
 f
 πf 


1 + 

 fsym 
 f3dB




0.5683 × 10 − 4 × f −1.5 , fint ≤ f ≤ 1.1MHz



12


MaskedOffsetdB( f )

10
, f < fint 
× 10





.
MaskOffsetdB ( f ) is defined as
f 3dB − f

1 + 0.4 ×
MaskOffsetdB(f) = 
f 3dB

1
, f < f 3dB
.
, f ≥ f 3dB
fint is the frequency where the two functions governing
SHDSLM ( f ) intersect in the range 0 to f sym .
K SHDSL , f sym , f 3dB , and the payload data rate R are defined in Table 13. At frequencies above 1.1 MHz, the
conformance criteria of clause 6.1 shall apply with
PT ( f ) = -110 dBm/Hz and PM ( f ) = -90 dBm/Hz.
The G.shdsl PSD shall be measured with the procedure specified in Section 6.4 while terminated in 135 Ohms, and
using the resolution bandwidths specified in Table 18.
Equipment that is classified as G.shdsl according to clause 5.4.2 shall also meet the total average power limitations,
transverse balance requirement, longitudinal output voltage requirement and deployment guidelines defined in
clause 5.4.2.
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ANSI T1.417
6.3.3 Specific conformance criteria for HDSL4 technology
The criteria of clause 6.1 are to be utilized.
6.4 PSD and total average power measurement procedure
The test methodology for measuring the PSD and the total average power of a device under test (DUT) are defined
in this clause. For each spectrum management class, there are two different transmitted PSD test cases:
a) Downstream (CO to Remote) transmission: the measured output of a central office transmission unit (TUC).
b) Upstream (Remote to CO) transmission: the measured output of a remote transmission unit (TU-R).
A DUT shall have total average power and PSD measured as described in this clause for both the upstream case and
the downstream case in order to determine compliance with the total average power, PSD conformance test, and
other applicable conditions of a spectrum management class as defined in this standard. Unless otherwise stated, all
specifications apply to both the upstream case and the downstream case. All measurements are performed directly
at the transmitter output of the DUT with no additional attenuation. Alternatively this test may be performed by
connecting the transmission units with a resistive pad.
6.4.1 Test circuit for PSD and total average power measurement
A test setup as pictorially shown in Figure 13 shall be used for measuring total average power and PSD. An
example of a specific embodiment of this test setup is the circuit in Figure 14. VOUT is connected to a highimpedance wideband RMS voltmeter or spectrum analyzer. The PSD may be tested while line powered or locally
powered as required by the intended application of the DUT.
If the DUT is line powered then the test circuit shall contain provisions for DC power feed. If the DUT is not line
powered then the DC power-feed circuitry may be omitted from the test circuit. For line powered applications, if the
DUT is a TU-C the test shall be performed with the line power supply activated and an appropriate DC current sink
(with high AC impedance) attached to the test circuit. If the DUT is a TU-R the test shall be performed with power
(DC voltage) applied at the line interface (TIP/RING) by an external voltage source feeding through an AC blocking
impedance. Note that the DC current source/sink must present high impedance (at signal frequencies) to common
ground. The test circuit contains provisions for transformer isolation for the measurement instrumentation.
Transformer isolation of the instrumentation input prevents measurement errors from unintentional circuit paths
through the common ground of the instrumentation and the DUT power feed circuitry. When the termination
impedance of the test circuit seen by the DUT output meets the calibration requirements defined in 6.4.2 the test
circuit will not introduce more than ± 0.25 dB error with respect to a perfect test load of exactly the specified
resistance.
The DUT shall be measured by equipment that is not synchronous with the transmitted symbols of the DUT, and
there shall be no synchronization between the measurement equipment and the DUT. This is to avoid making an
inaccurate measurement because of the effects of cyclo-stationarity.
6.4.2 Calibration of the test circuit and termination impedance
The nominal termination impedance of the test circuit as seen by the DUT output shall be resistive with a resistance
of Z Term as specified in Table 17 for the appropriate spectrum management class. The minimum return loss with
respect to the termination impedance Z Term over the frequency band of 1 kHz to 5 MHz shall be 35 dB from 10
kHz to 2 MHz with a slope of 20 dB/decade below and above these corner frequencies for measuring a DUT for
conformance with spectrum management classes 1, 2, 3, 4, 5, 7, 8 and 9.
The minimum return loss with respect to the termination impedance Z Term over the frequency band of 1 kHz to 30
MHz shall be 35 dB from 10 kHz to 20 MHz with a slope of 20 dB/decade below and above these corner
frequencies for measuring a DUT for conformance with spectrum management class 6.
Note: 35 dB return loss will allow ±0.20 dB measurement error with respect to the nominal termination impedance
value, Z Term .
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6.4.3 Operation of the DUT
The DUT shall be tested while it transmits maximum power and maximum PSD levels at all frequencies over which
it can transmit data when deployed. The DUT shall not have any power cutback enabled. The DUT shall be tested
under steady state conditions, after all start-up and initialization procedures have been completed and while the DUT
is transmitting data. To ensure that the DUT is in a steady-state condition, while undergoing test the DUT shall not
have measured total average powers in distinct 1.25 millisecond time intervals that differ by more than 8 dB.
Although specific measurements of average power and PSD during start-up and other non-data transmission phases
are not provided, a DUT that transmits inordinately high power or PSD levels during these phases may be
considered to be in non-compliance with this standard. The DUT input shall consist of a pseudo-random uniformly
distributed data sequence, and the DUT output shall be a fully modulated transmitted signal with all overhead,
framing, coding, scrambling, modulation, filtering and all other operations performed on the data stream that the
modem would normally perform while transmitting data.
6.4.4 Total average power measurement procedure
The average power of a DUT shall meet the total average power requirements as specified in clause 5.3 of this
standard over the bandwidth specified in clause 5.3 of this standard for conformance with a spectrum management
class. The total average power may be tested while line powered or locally powered as required by the intended
application of the DUT. The total average power shall be measured and averaged over a time span of at least 10
seconds.
6.4.5
Power spectral density (PSD) measurement procedure
6.4.5.1 PSD resolution bandwidth
The PSD is measured by averaging the observed output power of the DUT on each of a number of contiguous,
regularly spaced, small frequency bands with each frequency band having a defined resolution bandwidth. The PSD
of a DUT that is measured for conformance with spectrum management classes 1, 2, 3, 4, 7 and 8 shall be recorded
with frequency spacing equal to the resolution bandwidths specified in Table 18 at all frequencies from 1 kHz to 30
MHz.
The PSD of a DUT that is measured for conformance with spectrum management classes 5 and 9 shall be recorded
with frequency spacing equal to the resolution bandwidths specified in Table 19 at all frequencies from 1 kHz to 30
MHz.
The PSD of a DUT that is measured for conformance with spectrum management class 6 shall be recorded with
frequency spacing equal to the resolution bandwidths specified in Table 20 at all frequencies from 1 kHz to 30 MHz.
6.4.5.2 PSD Measurement time duration
Each frequency point (corresponding to a measurement in a single resolution bandwidth) of a PSD shall be
measured by averaging the power in the resolution bandwidth of that frequency point for a time period of at least 2.0
seconds. This requirement is equivalent to setting the sweep time for a single sweep of a spectrum analyzer for
duration equal to at least 2.0 seconds per frequency point.
Note: this requirement is based on a statistical derivation showing that to measure the average power in a given
resolution bandwidth within 0.1 dB accuracy with 99% confidence, required observation of about 9,000 transmitted
symbols, and the slowest common signal is an ADSL tone which is at a 4 kHz rate. Measuring an entire PSD for 2.0
seconds in all of each of the resolution bandwidths in Table 18, Table 19 and Table 20 requires minimum
observation times of 44, 20 and 70 minutes, respectively.
6.5 Short-term stationary conformance criteria
6.5.1 Determination of whether to apply short-term stationary conformance criteria
The short-term stationary conformance criteria in clause 6.5.2 through 6.5.4 shall be applied to a DUT if the total
average power transmitted by the DUT in any two non-overlapping 1.25 millisecond time intervals separated by less
than 60 seconds can differ by more than 8 dB. This includes variation due to the presence or absence of input data
for transmission or the presence of specific input data sequences but does not include variations due to external
stimuli such as the application of externally controlled power management, externally initiated retrain, or a change
in crosstalk levels or loop conditions that causes automatic retrain.
This is a draft document and thus, is still dynamic in nature.
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ANSI T1.417
Continuous mode for conformance testing
6.5.2
Equipment to which short-term stationary conformance criteria are applied shall provide a test configuration in
which the transmitter remains in the ON condition continuously. In the ON condition, the DUT shall transmit the
maximum power and the maximum PSD levels at all frequencies, at which it can transmit data when deployed. The
DUT shall not have any power cutback enabled. The DUT shall not have measured total average powers in any
distinct 1.25 millisecond time intervals that differ by more than 8 dB, including variation due to the presence or
absence of input data for transmission or the presence of specific input data sequences.
Frequency domain requirements
6.5.3
6.5.3.1 Continuous mode testing
Equipment to which short-term stationary conformance criteria are applied shall be tested in the continuous ON
condition specified in clause 6.5.2 using the conformance testing methodology defined in clauses 6.1, 6.2, and 6.4.
6.5.3.2 Short-term stationary mode testing
Equipment to which short-term stationary conformance criteria are applied shall be tested with input conditions that
generate the most frequent mode transitions permitted by the equipment. The conformance testing methodology
shall be as defined in clauses 6.1, 6.2, and 6.4 with the following exceptions:
−
The requirement in clause 6.4.3 that the DUT shall not have measured total average powers in distinct
1.25 millisecond time intervals that differ by more than 8 dB shall be waived.
−
Each frequency point (corresponding to a measurement in a single resolution bandwidth) of a PSD shall
be measured by averaging the power in the resolution bandwidth of that frequency point for a time
period of at least 4.0 seconds. This requirement is equivalent to setting the sweep time for a single
sweep of a spectrum analyzer for duration equal to at least 4.0 seconds per frequency point. This
requirement is used in place of the requirement in clause 6.4.5.2.
−
The equipment vendor shall identify the input conditions necessary to generate the mode transitions for
this test.
6.5.4 Time domain requirements
Equipment to which short-term stationary conformance criteria are applied shall transmit in the ON condition for a
minimum cumulative total of 40 milliseconds in any 4 second sliding window. In addition, the short-term stationary
transmitter shall continuously transmit in the ON condition for a minimum of 246 µsec. These requirements are
intended to facilitate detection of crosstalk from short-term stationary equipment by other receivers within a defined
time interval.
There shall not be any intentional synchronization of transmission bursts of short-term stationary systems.
6.6 Transverse balance testing methodology
Transverse balance10 is a comparison of the voltage of a transmitted metallic signal to the voltage of any resulting
longitudinal signal. It is defined in dB as:
Transverse Balance M −L = 20 Log10 [VM (f ) VL (f )]
,
where VM ( f ) = the metallic voltage at frequency f applied across tip and ring conductors of the port under test by
a balanced source with metallic impedance Z M , and VL ( f ) = the resultant longitudinal voltage appearing across a
longitudinal impedance Z L .
The greater the VM to VL ratio, the better the transverse balance of the transceiver unit and the less likelihood that it
will contribute to a crosstalk interference problem.
When calibrating the testing arrangement, the source metallic voltage should equal VM volts for each DSL class
when a metallic termination of Z M is substituted for the equipment under test. The metallic impedance Z M shall
be either 100 or 135 Ω as specified in clause 5.3.
10
Transverse Balance is equivalent to Transverse Conversion Loss as defined in ITU Recommendation O.9.
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The applicable Z L , Z M and VM values for each DSL class are summarized in Table 21.
The minimum transverse balance requirements for the TU-C and TU-R equipment shall be equaled or exceeded
during all operating states and under all reasonable application of earth ground to the equipment for the range of
applicable frequencies at all 2-wire loop ports with all values of loop current that the port under test is capable of
drawing when attached to the appropriate loop simulator circuit.
The transverse balance testing methodology in TIA/EIA TSB31-B (or equivalent) shall be used to determine
conformance with the transverse balance requirements as specified in clause 5.3 for each spectrum management
DSL class. An illustrative test configuration for transverse balance conformance testing is shown in Figure 15.
Table 2 provides the transverse balance requirements for various frequency ranges. The actual frequency range over
which the requirements apply is dependent on the system under test.
Transverse balance testing shall only be performed over the range of frequencies included in the power spectral
density applicable to the equipment under test and actually used in data transmission. For that purpose, all of the
signal passband shall be included, between the upper and lower -20 dB points (relative to peak PSD).
Transverse balance may be measured while the DUT is line powered or locally powered. If the DUT is line
powered then the test circuit shall contain a dc voltage source. In such applications, if the DUT is a TU-C the test
shall be performed with TU-C line power activated and an appropriate dc current sink (with high ac impedance)
attached to the test circuit. If the DUT is a TU-R, the test shall be performed with the appropriate dc voltage source
applied between the tip and ring conductors through an ac blocking impedance. The dc current source or sink must
present high impedance (at signal frequencies) to common ground. In line powered applications, the test circuit
shall contain provisions for isolation of the measurement instrumentation from unintentional circuit paths through
the common ground of the instrumentation and the DUT power feed circuitry.
6.7 Longitudinal output voltage testing methodology
Compliance with the limits as specified in clause 5.3 for each spectrum management DSL class is required with a
longitudinal termination having impedance equal to or greater than a 100 Ω resistor in series with a 0.15 µF
capacitor. An illustrative test configuration for longitudinal output voltage11 limit conformance testing is shown in
Figure 16. The metallic impedance Z M is defined as in Table 21. For direct use of that test configuration, the near
end transmitter should be able to generate a signal in the absence of a signal from the far-end transceiver. The
ground reference for these measurements shall be the building or green-wire ground of the DUT.
11
Longitudinal Output Voltage is equivalent to Output Signal Balance as defined in ITU Recommendation O.9.
This is a draft document and thus, is still dynamic in nature.
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Table 1 - SM class 1 PSD template definition
Frequency Band (kHz)
0 < f ≤ 25
PSD (dBm/Hz)
-32.5
25 < f ≤ 76
 f 
− 32.5 − 10.35 × log10  
 25 
76 < f ≤ 79
 f − 76 
− 37.5 − 0.5 × 

 3 
79 < f ≤ 85
 f − 69 
− 38 − 19.6 × log10 

 10 
85 < f ≤ 100
100 < f ≤ 115
115 < f ≤ 120
f − 85
15
f − 100
− 46 − 7 ×
15
-53
− 42 − 4 ×
120 < f ≤ 225
 f 
- 53 − 55 × log10 

 120 
225 < f ≤ 635
 f 
- 68 − 70 × log 10 

 225 
635 < f
 (1000 × f )3 2
-143 - 10 log 10 
 1.134 × 1013





Table 2 - Minimum transverse balance requirements
Frequency Band
200 Hz -12 kHz
12 kHz - 1544 kHz
1544 kHz - 3000 kHz
30
Minimum Transverse Balance
40 dB
35 dB
30 dB
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table 3 - SM class 2 PSD template definition
Frequency (kHz)
0
25
75
100
150
200
230
245
335
390
440
475
500
500 < f
PSD (dBm/Hz)
-36
-36
-36.5
-39
-45
-54
-64
-71
-72
-76
-83
-90
-98
 (1000 × f )3 2
-143 - 10 log 10 
 1.134 × 1013





Table 4- SM class 3 PSD template definition
Frequency (kHz)
0
50
125
210
310
370
550
670
750
980
1050
1050 < f
PSD (dBm/Hz)
-37
-37
-38
-41
-57
-73
-75
-85
-97
-98
-102.75
 (1000 × f )3 2 

-143 - 10 log10 
 1.134 × 1013 


This is a draft document and thus, is still dynamic in nature.
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Table 5 - PSD mask definition for downstream transmission from SM class 4 TU-C
Frequency
(kHz)
≤1
2
12
190
236
PSD
(dBm/Hz)
-54.2
-42.2
-39.2
-39.2
-46.2
Frequency
(kHz)
280
375
400
440
600
PSD
(dBm/Hz)
-35.7
-35.7
-40.2
-68.2
-76.2
Frequency
(kHz)
1000
2000
>3000
PSD
(dBm/Hz)
-89.2
-99.7
-108
Table 6 - PSD mask definition for upstream transmission from SM class 4 TU-R
Frequency
(kHz)
≤1
2
10
175
PSD
(dBm/Hz)
-54.2
-42.1
-37.8
-37.8
Frequency
(kHz)
220
255
276
300
PSD
(dBm/Hz)
-34.4
-34.4
-41.1
-77.6
Frequency
(kHz)
555
800
1400
≥ 2000
PSD
(dBm/Hz)
-102.6
-105.6
-108
-108
Table 7 - PSD template definition for downstream transmission from SM class 5 TU-C
Frequency Band (kHz)
0<f≤4
4 < f ≤ 25.875
25.875 < f ≤ 81
81 < f ≤ 92.1
92.1 < f ≤ 121.4
121.4 < f ≤ 138
138 < f ≤ 1104
1104 < f ≤ 3093
PSD (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21 × log2(f/4)
-40
-40 - 70 × log2(f/81)
-53
-53 + 70 × log2(f/121.4)
-40
-40 - 36 × log2(f/1104)
min(–36.5 –36 × log2(f/1104), -93.5)
- 110
3093 < f ≤ 4545
4545 < f ≤ 11040
Table 8 - SM class 7 PSD template definition
Frequency (kHz)
0
100
150
200
300
390
420
500
32
PSD (dBm/Hz)
-40
-40
-40.5
-41.5
-42
-42
-43
-51
Frequency (kHz)
775
1000
1100
1300
1500
1900
2000
> 2000
PSD (dBm/Hz)
-77
-77
-80
-86
-102
-104
-107
 (1000 × f )3 2
-143 - 10 log 10 
 1.134 × 10 13

This is a draft document and thus, is still dynamic in nature.




T1E1.4/2000-002R6
ANSI T1.417
Table 9 - SM class 8 PSD template definition.
Frequency
(kHz)
0
60
200
250
PSD
(dBm/Hz)
-39
-39
-40
-40.5
Frequency
(kHz)
400
500
550
750
PSD
(dBm/Hz)
-53
-66
-75
-76
315
-41
950
-84
Frequency
(kHz)
1120
1500
2000
> 2000
PSD
(dBm/Hz)
-95
-95
-107
 (1000 × f )3 2 

-143 - 10 log10
 1.134 × 1013 


Table 10 – PSD template definition for downstream transmission from SM class 9 TU-C
Frequency Band (kHz)
0<f<4
4 < f < 25.875
25.875 < f < 1104
1104 < f < 3093
3093 < f < 4545
4545 < f < 11040
PSD (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21 × log2(f/4)
-40
-40 – 36 × log2(f/1104)
min(–36.5 –36 × log2(f/1104), -93.5)
- 110
Table 11 - PSD template definition for upstream transmission from SM class 9 or SM
class 5 TU-R
Frequency Band (kHz)
0<f<4
4 < f < 25.875
25.875 < f < 138
138 < f < 307
307 < f < 1221
1221 < f < 1630
1630 < f < 11040
PSD (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21.5 × log2(f/4)
-38
-38 – 48 × log2(f/138)
-93.5
min (–90 – 48 × log2(f/1221), -93.5)
-110
This is a draft document and thus, is still dynamic in nature.
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Table 12 - Deployment guidelines for 2B1Q SDSL
SM1 PSD template
SDSLu ( f ) with fsym = 160000
Maximum
2B1Q SDSL
line bit rate (kbps)
300
320
2B1Q SDSL
deployment guideline,
EWL (kft)
all non-loaded loops
15.5
TS103
SDSLu ( f ) with fsym = 168000
336
14.5
TS104
SDSLu ( f ) with fsym = 208000
416
13.5
TS105
SDSLu ( f ) with fsym = 232000
464
12.5
TS106
SDSLu ( f ) with fsym = 264000
528
12
TS107
SDSLu ( f ) with fsym = 296000
592
11.5
TS108
SDSLu ( f ) with fsym = 328000
656
11
TS109
SDSLu ( f ) with fsym = 360000
720
10.5
TS110
SDSLu ( f ) with fsym = 392000
784
10
TS111
SDSLu ( f ) with fsym = 456000
912
9.5
TS112
SDSLu ( f ) with fsym = 520000
1040
9
TS113
SDSLu ( f ) with fsym = 552000
1104
8.5
TS114
SDSLu ( f ) with fsym = 616000
1232
8
TS115
SDSLu ( f ) with fsym = 712000
1424
7.5
TS116
SDSLu ( f ) with fsym = 840000
1680
7
TS117
SDSLu ( f ) with fsym = 936000
1872
6.5
TS118
SDSLu ( f ) with fsym = 1064000
2128
6
TS119
SDSLu ( f ) with fsym = 1128000
2256
5.5
TS120
SDSLu ( f ) with fsym = 1160000
2320
5
Designation
PSD
TS101
TS102
Table 13 - G.shdsl Symmetric PSD parameters
Line Bit Rate
LBR (kbps)
K SHDSL
f sym
f3dB
LBR ≠ 1544 or
7.86
(ksymbol/s)
LBR /3
1552
= 1544 or
1552
1.0 × f sym /2
8.32
LBR /3
0.9 × f sym /2
LBR
34
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Table 14 - Deployment guidelines for G.shdsl
Designation
TS201
TS202
TS203
TS204
TS205
TS206
TS207
TS208
TS209
TS210
TS211
TS212
TS213
TS214
TS215
TS216
TS217
TS218
TS219
G.shdsl Line
Bit Rate (kbps)
LBR ≤ 592
600 ≤ LBR ≤ 616
624 ≤ LBR ≤ 648
656 ≤ LBR ≤ 688
696 ≤ LBR ≤ 800
808 ≤ LBR ≤ 832
840 ≤ LBR ≤ 896
904 ≤ LBR ≤ 952
960 ≤ LBR ≤ 1000
1008 ≤ LBR ≤ 1088
1096 ≤ LBR ≤ 1160
1168 ≤ LBR ≤ 1320
1328 ≤ LBR ≤ 1472
1480 ≤ LBR ≤ 1536
1544 ≤ LBR ≤ 1552
1560 ≤ LBR ≤ 1664
1672 ≤ LBR ≤ 1880
1888 ≤ LBR ≤ 2008
2016 ≤ LBR ≤ 2320
G.shdsl deployment
guideline, EWL(kft)
All non loaded loops
15.0
14.5
14.0
13.5
12.5
12.0
13.0
12.5
12.0
11.5
11.0
10.5
10.0
10.5
10.0
9.5
9.0
8.5
Table 15 - PSD template definition for downstream transmission from HDSL4 TU-C
Frequency
(kHz)
≤0.2
2
5
50
80
90
105
PSD
(dBm/Hz)
-51
-41
-37.5
-37.5
-40.5
-45
-58
Frequency
(kHz)
110
135
145
150
155
200
210
PSD
(dBm/Hz)
-58
-46.5
-40.5
-38.5
-37.5
-40.25
-43
Frequency
(kHz)
250
400
600
1000
2000
3000
≥3100
This is a draft document and thus, is still dynamic in nature.
PSD
(dBm/Hz)
-51.5
-46.5
-70
-89.2
-99.7
-108
-110
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Table 16 - PSD template definition for upstream transmission from HDSL4 TU-R
Frequency, f (Hz)
0 < f ≤ 200
200 < f ≤ 2000
2000 < f ≤ 5000
5000 < f ≤ 50000
50000 < f ≤ 125000
125000 < f ≤ 130000
130000 < f ≤ 307000
307000 < f ≤ 1221000
1221000 < f ≤ 1630000
PSD (dBm/Hz)
-51
-41+10(f−2000)/1800
-37+4(f−5000)/3000
-37
-37 – ((f−50000)/75000)
-38
-38 – 142log10(f/130000)
-93.5
min (-90 – 48 × log2(f/1221000), -93.5)
f > 1630000
-110
Table 17 - Termination impedance
Spectrum management
class
Termination impedance
Z Term (Ω)
Class 1
Class 2
Class 3
Class 4
Class 5
Class 6
Class 7
Class 8
Class 9
135
135
135
135
100
100
135
135
100
Table 18 - Resolution bandwidth for measuring a DUT PSD for conformance with SM
classes 1, 2, 3, 4, 7 and 8
Frequency Band
f ≤ 10 kHz
10 kHz ≤ f ≤ 3.1 MHz
3.1 MHz ≤ f ≤ 30 MHz
36
Resolution Bandwidth
1 kHz
3 kHz
100 kHz
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table 19 - Resolution bandwidth for measuring a DUT PSD for conformance with SM
classes 5 and 9
Frequency Band
f ≤ 10 kHz
10 kHz < f ≤ 25.875 kHz
25.875 kHz < f ≤ 3.1 MHz
3.1 MHz < f ≤ 30 MHz
Resolution Bandwidth
100 Hz
1 kHz
10 kHz
100 kHz
Table 20 - Resolution bandwidth for measuring a DUT PSD for conformance with SM
class 6
Frequency Band
f ≤ 10 kHz
10 kHz < f ≤ 20 MHz
20 MHz < f ≤ 30 MHz
Resolution Bandwidth
1 kHz
10 kHz
100 kHz
Table 21 - Summary of transverse balance testing criteria
ZL
(Ω)
ZM
SMC 1 SMC 2 SMC 3 SMC 4 SMC 5 SMC 6 SMC 7 SMC 8 SMC 9
500/90 500/90 500/90 500/90
90
TBD 500/90 500/90
90
(1)
(1)
(1)
(1)
(1)
(1)
135
135
135
135
100
TBD
135
135
100
(Ω)
VM
0.367
0.367
0.367
0.367
0.316
TBD
0.367
0.367
0.316
(volts)
NOTES: (1) The longitudinal impedance (ZL) shall be 500 Ω for frequencies from 200 Hz to 12 kHz and 90 Ω for
frequencies above 12 kHz.
Table 22 - Maximum longitudinal output voltage limit
Applicable
Frequency
Range
Operating band
From upper –30 dB (relative to peak PSD)
frequency to 4 × the upper -30 dB frequency
Maximum Longitudinal Output
Voltage (rms) in all
4 kHz Frequency Bands
averaged over 1 second
-50 dBV
-80 dBV
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R6
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-30
PSD Template (dBm/Hz)
-40
-50
-60
-70
-80
-90
-100
-110
0
100
200
300
400
500
600
700
800
Frequency (kHz)
Figure 1 - SM class 1 PSD template
-30
-40
PSD (dBm/Hz)
-50
-60
-70
-80
-90
-100
-110
0
100
200
300
400
500
600
Frequency (kHz)
Figure 2 - SM class 2 PSD Template
38
This is a draft document and thus, is still dynamic in nature.
700
800
T1E1.4/2000-002R6
ANSI T1.417
-30
-40
PSD (dBm/Hz)
-50
-60
-70
-80
-90
-100
-110
0
200
400
600
800
1000
1200
Frequency (kHz)
Figure 3 - SM class 3 PSD template
-30
PSD (dBm/Hz)
-40
-50
-60
-70
-80
-90
-100
0
100
200
300
400
500
600
700
800
900 1000
Frequency (kHz)
Figure 4 - PSD mask for downstream transmission from SM class 4 TU-C
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R6
ANSI T1.417
-30
-40
P SD (dB m /Hz )
-50
-60
-70
-80
-90
-100
0
100
20 0
300
40 0
5 00
600
Freq uen cy (kHz)
Figure 5 - PSD mask for upstream transmission from SM class 4 TU-R
-30
-40
-50
PSD (dBm/Hz)
-60
-70
-80
-90
-100
-110
-120
0
1000
2000
3000
4000
5000
F req uen cy (kH z)
Figure 6 - PSD template for downstream transmission from SM class 5 TU-C
40
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
-3 0
-4 0
-5 0
PSD (dBm/Hz)
-6 0
-7 0
-8 0
-9 0
-1 0 0
-1 1 0
-1 2 0
0
200
400
600
800
1000
1200
1400
1600
1800
2000
F re q u e n c y (k H z )
Figure 7 - PSD template for upstream transmission from SM class 5 or SM class 9 TU-R
-40
PSD (dBm/Hz)
-50
-60
-70
-80
-90
-100
-110
0
200
400 600 800 1000 1200 1400 1600 1800 2000 2200
Frequency (kHz)
Figure 8 - SM class 7 PSD template
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R6
ANSI T1.417
-30
PSD (dBm/Hz)
-40
-50
-60
-70
-80
-90
-100
-110
0
500
1000
1500
2000
Frequency (kHz)
Figure 9 - SM class 8 PSD template
-3 0
-4 0
-5 0
PSD (dBm/Hz)
-6 0
-7 0
-8 0
-9 0
-1 0 0
-1 1 0
-1 2 0
0
500
1000
1500
2000
2500
3000
3500
4000
4500
5000
F re q u e n c y (k H z )
Figure 10 - PSD template for downstream transmission from SM class 9 TU-C
42
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
-30
-40
P S D (dB m /Hz )
-50
-60
-70
-80
-90
-100
0
100
200
300
400
500
600
Frequenc y (k Hz )
700
800
900
1000
Figure 11 - PSD template for downstream transmission from HDSL4 technology TU-C
-30
-40
P S D (dB m /Hz )
-50
-60
-70
-80
-90
-100
0
100
200
300
400
500
600
Frequenc y (k Hz )
700
800
900
1000
Figure 12 - PSD template for upstream transmission from HDSL4 technology TU-R
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R6
ANSI T1.417
1 uF (min)
V
Resistive
Termination, R Ohms
(ground isolated
input)
out
Tip
Return loss
as per calibration
DC
current
sink
Device
under test (DUT)
of the test circuit
Ring
1 uF (min)
Figure 13 - PSD and total average power measurement setup
1:1
(+/- 1%)
Vout
(To highimpedance
load)
20 mH (min)
1 uF (min)
R
DC
current
sink
Tip
Return loss
as per calibration
of the the test circuit
Device
under test (DUT)
Ring
1 uF (min)
Figure 14 - Example PSD and total average power measurement setup
44
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
T1 (2)
S3
S2
1:1
S1
20 pF (5)
Tracking
Generator
EM
Equipment
Under Test
VM (6)
R2 (3)
R3 (7)
S2
R1 (1)
EL
S3
Spectrum
Analyzer (4)
EL = longitudinal voltage
EM = metallic voltage
1- Combined resistance of R1 and tracking generator output resistance shall equal DUT impedance (I.e., 100 or 135 Ω).
2- Use center-tapped 1:1 transformer (e.g., Midcom 671-5767 or equivalent.
3- R2 provides the desired longitudinal impedance using 90 Ω or 500 Ω metal film or other non-inductive resistor.
4- High impedance spectrum analyzer or frequency selective voltmeter. It may be unbalanced.
5- Differential trimmer capacitor, 2.4 to 24.5 pF, Johnson 189-0759-005 or equivalent.
6- Any high impedance balanced or floating voltmeter with adequate frequency response. It need not be frequency selective.
7- R3 provides the desired calibration impedance. Should be a 100 or 135 Ω metal film or other non-inductive resistor.
Figure 15 - Illustrative test configuration for transverse balance conformance testing
ZM 2
NT
(see Note)
ZM 2
(see Note)
0.15 µF
100 Ω
Spectrum
analyzer
NOTE – These resistors to be matched better than 0.1% tolerance.
Figure 16 - Measurement method for longitudinal voltage
This is a draft document and thus, is still dynamic in nature.
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Annex A: Evaluation of interference from new technologies (Method B)
(Normative)
A.1 Goals and framework for evaluation
The goal of spectral compatibility analysis described in this clause is two-fold:
a) to provide tests to validate that new service technologies will not interfere with the basis systems, and
b) to allow sufficient flexibility to nurture innovation in new subscriber line transmission technologies that
further maximize the utility of the copper loop plant.
To achieve both goals simultaneously, this clause describes computations that may be performed on new signals to
demonstrate spectral compatibility with the basis systems.
The rates and reaches of basis systems in this Annex are provided only for analytical evaluation of spectral
compatibility with the basis systems.
Complying with a spectrum management class in clause 5.3 of this standard provides a simplified test for spectral
compatibility. However, this test alone would preclude large classes of new transmission schemes which are
spectrally compatible, and would stifle creativity for providing copper access solutions. In order to nurture
spectrally compatible innovation, this clause describes a second, more complicated evaluation (Method B) that may
be used to demonstrate spectral compatibility, technology by technology, with basis systems. Method B establishes
industry practices for demonstrating spectral compatibility of new technologies or new spectrum management
classes with the basis systems. These analyses can be used as an aid to add to the spectrum management classes in
this standard at later dates. The adoption of a new spectrum management class is defined in clause 4.3.3.
The use of information in this Annex should be limited to the analysis of new technologies and proposed spectrum
management classes. This Annex, including the definitions and performance criteria, does not and is not intended to
define “significant degradation”. Nor should any expectation of actual performance be drawn or extrapolated from
the information in this Annex.
Whether Method B or Method A is used to demonstrate spectral compatibility, the new technology shall meet the
transverse balance specifications in Table 2 and the longitudinal voltage limits in Table 22 using the testing
methodologies for those parameters in clauses 6.6 and 6.7.
A.2 Analytical Method: Detailed crosstalk margin evaluations
Detailed margin calculations are required to demonstrate spectral compatibility of new technologies outside of the
established spectrum management classes. These calculations are described in this clause and must be calculated for
each basis system. Because some technologies are spectrally asymmetric, that is, use a different transmitted
spectrum in each direction, evaluations must be performed in both the upstream and downstream directions.
The use of this clause establishes spectral compatibility of a new technology with the basis system set. In this
method, the established reference disturbers are replaced in equal number by the new technology under trial, and the
performance margin of the technology being disturbed by the new technology is compared to the established
reference case. Appropriate reference evaluation loops, specified herein, are used for both the reference and new
technology disturber calculations. As noted in clause A.1, the specified reference performance levels are not
intended to be performance targets for systems in the real world; they are only useful for comparing the impacts on a
basis technology due to crosstalk from a new technology and crosstalk from reference technologies.
This section is organized as follows. The sub-sections of A.2 describe the loop characterization and crosstalk
models, the general methodology and the specific margin calculations for a variety of technologies. Subsequent
sections give the transmission and performance parameters, and reference performance levels associated with each
basis system. As new basis systems are designated, a section can be inserted into future versions of this standard
detailing the established performance benchmarks and method for calculating compatibility with the new
technology.
There are four types of margin computations described in this clause: DFE-based PAM signals (e.g., 2B1Q BRI,
HDSL and HDSL2), DFE-based QAM/CAP signals (e.g. CAP signals), DMT-based signals (e.g., T1.413-1998
ADSL), and linear-equalization based signals (e.g., DDS). Which computation is used depends on the basis system
being tested, not the nature of the proposed technology.
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A.2.1 Loop Characterization
This clause addresses the transmission characteristics of 26-AWG twisted-pair copper phone lines. See B.3 for
general information about loop characterization.
A.2.1.1 Cable primary constants RLCG
The primary constants of resistance (R, Ω/km), inductance (L, H/km), capacitance (C, F/km), and conductance (G,
Mho/km) are used to model most transmission lines. Secondary parameters such as impedance, attenuation and
phase or the two-port modeling parameters may be calculated from the primary constants. Because of error in
practical measurements of the impedance, tabulated RLCG values may not follow smooth curves with frequency so
parameterized models of RLCG are then fit to the measured values. The models of R, L, C and G for copper cable
are:
R( f ) = 4 r04c + ac ⋅ f 2
 f 
l0 + l∞  
 fm 
L( f ) =
b
 f 
1 +  
 fm 
b
C ( f ) = c∞
G( f ) = g 0 ⋅ f
ge
The parameters for 26-AWG twisted-pair cable are found in Table A. 1
A.2.1.2 Twisted-pair cable insertion gain transfer function
Insertion gain transfer function is the ratio of the voltage
to the voltage
VL across the load Z L with the transmission line inserted
Vno across the load Z L with no transmission line. For twisted-pair cable, the insertion gain transfer
function is given by
H( f ) =
VL
ZL + ZS
=
Vno Z ⋅ cosh(γd ) + Z ⋅ sinh (γd ) + Z S ⋅ Z L ⋅ sinh (γd ) + Z ⋅ cosh(γd )
L
S
0
Z0
where the characteristic impedance of the transmission line
Z0 =
R + jωL
G + jωC
the propagation constant
γ =
(R +
,
Z 0 at any given frequency ω = 2πf is
,
per unit length for the twisted-pair at any given frequency
jωL ) ⋅ (G + jωC )
ω = 2πf
is
,
d is the length of the twisted-pair cable, Z L is the load impedance, and Z S is the source impedance.
The dB insertion loss of the twisted-pair cable is
IL( f ) = −10 ⋅ Log10 H ( f )
2
.
A.2.2 Cable crosstalk models
See B.4 for general information about cable crosstalk.
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A.2.2.1 Near end crosstalk, NEXT
Crosstalk noise that occurs when a receiver on a disturbed pair is located at the same end of the cable as the
transmitter of a disturbing pair is called Near-End-Crosstalk (NEXT).
A.2.2.1.1
Simplified NEXT model
The simplified NEXT model has loss values of 57 dB, 61.1 dB, and 67.1 dB for 49 disturbers, 10 disturbers, and 1
disturber, respectively, at a frequency of 80 kHz and a linear (log-log) slope of –15 dB per decade. See Figure A. 1.
The simplified NEXT model is expressed by
NEXT [ f , n ] = S ( f ) ⋅ X N ⋅ n 0.6 ⋅ f 3 / 2
where
X N = 8.536 × 10−15 , n = number of disturbers, f = frequency in Hz, and S ( f ) is the power spectrum
of the interfering system.
A.2.2.1.2
Two-piece NEXT model
The two-piece model is identical to the Unger model for 1, 10, and 49 disturbers, and is an interpolated extension of
that model for 24 disturbers. See Figure A. 1. The two-piece NEXT model is expressed by
NEXT [f , n ] = S(f ) ⋅
U 0 ⋅ f U1 , f < 20,000Hz
U 2 ⋅ f U3 , f ≥ 20,000Hz
where f = frequency in Hz, S(f ) is the power spectra of the interfering system, and
U 0 , U1 , U 2 and U 3 are
defined in Table A. 2 for 1, 10, 24, and 49 disturbers.
A.2.2.2 Far end crosstalk, FEXT
Crosstalk noise that occurs when a receiver on a disturbed pair is located at the other end of the cable as the
transmitter of the disturbing pair is called Far-End-Crosstalk (FEXT). The FEXT model is expressed by
2
FEXT [f , n, l ] = S(f ) ⋅ H (f ) ⋅ X F ⋅ n 0.6 ⋅ l ⋅ f 2
where
H (f )
is the magnitude of the insertion gain transfer function affecting the disturber signal,
X F = 7.74 × 10 −21 , n = number of disturbers, l = the coupling path length in feet, f = frequency in Hz, and
S(f ) is the power spectrum of the interfering system.
The FEXT model assumes the insertion gain transfer function is computed for the total cable path located between
the interfering transmitter and the victim receiver. On the other hand, the coupling loss is computed only over the
coupling path length l . The coupling path length is the length of cable over which the victim receiver and far-end
disturbing transmitter have a common cable path.
A.2.2.3 FSAN method for combining crosstalk contributions from unlike types of disturbers
Instead of directly adding the crosstalk power terms, each term is first arbitrarily raised to the power 1/0.6 before
carrying out the summation. Then, after the summation, the resultant expression is raised to the power 0.6. This can
be expressed as:

Xtalk  f , n =


where Xtalk is either NEXT or FEXT,
unlike disturbers and

∑ n i  =
i =1 
N
N

 ∑ Xtalk (f , n i )1/ 0.6 


 i =1

0.6
,
n is the total number of crosstalk disturbers, N is the number of types of
n j is the number of each type of disturber. Example uses of this equation are given in the
following subsections.
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Example application of two NEXT terms
Take the case of two sources of NEXT at a given receiver. In this case there are
S1 (f ) and
n1 disturber systems of spectrum
n2 disturber systems of spectrum S 2 (f ) .
The combined NEXT is expressed as:
(

3
0.6
NEXT [f , n ] =  S1 (f ) X N f 2 n1

A.2.2.3.2
)
1
0. 6
+
( S (f ) X
2
N
f
3
2
n2
0.6
)
1
0. 6



0.6
Example application of three FEXT terms
Take the case of three sources of FEXT at a given receiver. In this case there are n1 disturber systems of spectrum
S1 (f ) at range l 1 , a further n 2 disturber systems of spectrum S 2 (f ) at range l 2 and yet another n 3 disturber
systems of spectrum S 3 (f ) at range l 3 .
The expected crosstalk is built in exactly the same way as before, taking the base model for each source, raising it to
power 1/0.6, adding these expressions, and raising the sum to power 0.6:
1
1


 S1(f )] H1(f ) 2 X F f 2 l 1 n10.6  0.6 + S2 (f ) H2 (f ) 2 X F f 2 l 2 n2 0.6  0.6






FEXT[f , n, l ] = 

1
2
0.6 0.6 

+ S3 (f ) H3 (f ) X F f 2 l 3 n3 





0.6
A.2.2.4 Total noise power spectral density
The NEXT term and the FEXT term are computed to arrive at separate NEXT and FEXT disturbance power spectra.
These power spectra should then be summed with the background noise to determine the total disturbance seen by
the victim receiver. The total noise seen by the victim receiver is given by:
N (f ) = NEXT [f , n ] + FEXT [f , n, l ] + 10 −14 mW/Hz
A.2.3 General Methodology
The general model used for calculating both the reference performance levels and the performance of the existing
system in the presence of the new technology is shown in Figure A. 2. Note that the crosstalk noise may be a mix of
NEXT and FEXT from reference disturbers and/or new disturbers. When either NEXT or FEXT is made up of
different overlapping noise spectra, each should be constructed independently of the other using the Full Service
Access Networks (FSAN) method described in section A.2.2.3. The NEXT may then be combined with the FEXT
by a simple power sum.
The simulation model for the downstream is shown in Figure A. 3. Note that the simulation model for the upstream
is exactly analogous to Figure A. 3, but with TU-C and TU-R transposed in the diagram.
The calculations to determine spectral compatibility with the basis systems proceed as follows:
1. For both upstream and downstream directions, calculate the target margin for the basis system per Figure A. 2
and Figure A. 3 at target equivalent working length Z, using only reference disturbers. When calculating the
impact of the new technology on the basis system, the crosstalk noise from the new technology replaces the
appropriate number of reference disturbers.
2. When calculating the effect of new technologies on a basis system performance, relative margin calculations
shall be performed for all basis systems, except for Voice, P-Phone and ADSL basis systems that use fixed
margins. The same parameters used to calculate the target margin are used to calculate the margin in the
presence of crosstalk noise from the new technology. Target loop length Z and target bit rate are input to the
calculation and the resulting margin is output. This new margin is compared with the target margin obtained in
step 1. If the new margin is no more than some delta below the target margin for both upstream and
downstream calculations (the value of delta, if any, is defined in clauses A.3 and following), then the new
system is spectrally compatible with the basis system.
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3.
If the upstream margin of the basis is below the target margin by some value greater than delta, then a new test
shall be performed with a decreased value of loop length Z called U′. All new and reference disturbers as well
as the basis technology are maintained on the same U′-kft loop, and the NEXT and FEXT levels they present
are calculated for the U′ loop.
The calculation shall be repeated for the upstream in an iterative fashion, reducing the loop length, until the
largest value of U′ is found that allows the basis system to maintain a margin which is no more than delta below
the target margin obtained in step 1.
4. If the downstream margin of the basis system is below the target margin by some value greater than delta, then
a new test shall be performed with a decreased value of loop length Z called D′. All new and reference
disturbers as well as the basis technology are maintained on the same D′-kft loop, and the NEXT and FEXT
levels they present are calculated for the D′ loop.
The calculation shall be repeated for the downstream in an iterative fashion, reducing the loop length, until the
largest value of D′ is found that allows the basis system to maintain a margin which is no more than delta below
the target margin obtained in step 1.
5. Steps 1 to 4 are repeated for all cases of different crosstalk combinations specified for the basis system, and the
minimum of all values of loop lengths U’ and D′ found in steps 3 and 4 is called Y.
6. Steps 1 to 5 are repeated for all basis systems in clauses A.3 and following, and the minimum of all values of
loop length Y found in step 5 is called Z’. Then Z’ is rounded to the nearest 500-ft increment to equal the loop
deployment guideline for the new technology.
This process is depicted in Figure A. 4. A variant of the general model is applied to the ADSL basis system and is
described in clause A.8. The new technology shall then be considered spectrally compatible only when it is
deployed on loops of length less than or equal to the round-off to the nearest 500-ft increment of Z′.
A.2.4 DFE-based PAM signals (e.g., 2B1Q BRI and HDSL)
Margin for DFE-based PAM technologies is computed using an Optimal DFE calculation for PAM:
fbaud
Margin =
1 ⌠
fbaud 
⌡0
10 ∗ log 10( 1 + f _ SNR( f ))df − SNR_req dB
where f _ SNR( f ) is the folded received signal-to-noise ratio, defined as:
1
f _ SNR( f ) =
∑
n = −2
S( f + fbaud × n ) | H( f + fbaud × n ) | 2
N( f + fbaud × n )
where S( f ) is the desired signal’s (e.g., BRI or HDSL) transmitted power spectral density in mW/Hz, H ( f )
2
is the
magnitude squared of the loop insertion gain transfer function, and N( f ) is the total noise power spectral density
(crosstalk plus background noise) at the receiver in mW/Hz computed as described in clause A.2.2. SNR folding,
calculated out to 4 times the Nyquist rate (twice the baud rate) is sufficient for all current xDSL signals. If future
signals use more bandwidth, they may require expansion of the range of n in the summation.
The C code in Table A.3 computes the optimal DFE SNR for PAM signals, from the given two arrays containing
received signal and received noise power spectral densities. By using the code and subtracting the required SNR
from the result, one can compute PAM DFE margins. PSDs are double-sided and negative frequencies must be
taken into account in this analysis.
A.2.5 DFE-based QAM/CAP signals
Margin for DFE-based CAP/QAM technologies is computed using an Optimal DFE calculation for QAM:
fbaud
1 ⌠
Margin =
fbaud 
⌡0
10 ∗ log 10(1 + f _ SNR( f ))df − SNR _ req dB
where f _ SNR( f ) is the folded received signal-to-noise ratio, defined as:
3
f _ SNR( f ) =
∑
n =0
50
S( f + fbaud × n ) | H( f + fbaud × n ) | 2
N ( f + fbaud × n )
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where S( f ) is the desired signal’s transmitted power spectral density in mW/Hz, H ( f )
2
is the magnitude squared
of the loop insertion gain transfer function, and N( f ) is the total noise power spectral density (crosstalk plus
background noise) at the receiver in mW/Hz computed as described in clause A.2.2.
As in the PAM case, SNR folding is calculated out to 4 times the Nyquist rate, yet for QAM this is 4 times the baud
rate. As for PAM, future signals that use more bandwidth may require expansion of the range of n in the
summation.
Unlike PAM signals, it is important that the region of folding be sufficient to include any offset for the carrier
frequency of the QAM/CAP signal. This may be included either by changing the limits of integration or by
changing the limits on n in the SNR folding summation to adequately span the frequencies used by the signal.
The C code in Table A. 4 computes the optimal DFE SNR for QAM/CAP signals, given two arrays containing
received signal and received noise power spectral densities. By using the code in Table A. 4 and subtracting the
required SNR from the result, one can compute QAM/CAP DFE margins.
A.2.6 DMT margin computations
DMT systems allocate bits to individual carriers based on the Shannon capacity of the tones. Margin for these
systems is determined by the Shannon capacity (minus appropriate SNR gap, and plus coding gain), and then
degrading the SNR at all frequencies until the capacity is equal to the desired data rate. Capacity per Hz at an
individual frequency is given by:
2

S ( f ) × H ( f ) 

C ( f ) = log 2 1 +

N ( f ) × 10 Γ /10 

where S( f ) is the desired signal’s transmitted power spectral density in mW/Hz,
H( f )
2
is the magnitude
squared of the loop insertion gain transfer function, and N( f ) is the total noise power spectral density (crosstalk
plus background noise) at the receiver in mW/Hz computed as described in clause A.2.2, and Γ is the effective SNR
gap in dB. For coded systems, SNR gap is defined as (9.75 - (effective coding gain)) dB. For the purposes of
margin calculations, the effective SNR gap is increased by the desired margin, and is defined as Γ = 9.75 - (effective
coding gain) + Margin (dB).
Total capacity for the DMT system is then computed by integrating C ( f ) across the frequency band used by the
DMT system. Some DMT systems have a minimum number of bits per tone (such as T1.413-1995, T1.413-1998,
and ITU-T G.992.2, all of which support a minimum of 2 bits/tone ( MINBITS = 2 )). In calculations for these
systems, C ( f ) must be further limited not to exceed the prescribed maximum.
When computing DMT capacity, the resulting integration is conditional at each frequency:
C=
∫DMT bandwidth C ′( f )df ,
where C ′( f ) = min(C ( f ), MAXBITS ) if C ( f ) > MINBITS , and C ′( f ) = 0 if C ( f ) < MINBITS , and DMT
bandwidth is the frequency range used by the data carrying tones of the desired DMT signal. It is worth noting that
implemented DMT systems go through a process of bit loading and adjustment of powers to each of the tones.
However, studies have shown that margins achieved by such algorithms closely match those achieved by the less
implementation dependent capacity calculation shown here.
The Matlab-code in Table A.5 and Table A. 6 computes DMT margins.
A.2.7 Spectral Compatibility Computation Results
Computational results to determine the spectral compatibility of the spectrum management classes and some
technologies are shown in Table A. 7.
Technologies not included in these results can use the same method of Annex A to determine spectral compatibility.
A.2.8 Common parameters for computing spectral compatibility
Unless otherwise specified, the sum of NEXT plus FEXT plus -140 dBm/Hz background noise is used in all
calculations. The FEXT model described in clause A.2.2.2 is used. Crosstalk sources that may be represented by
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either a PSD template or a PSD mask are simulated with the PSD template generating the crosstalk. Remote
transmission units and remote disturbers are collocated.
The parameters for the loop model are described in clause A.2.1. Loop loss shall be calculated with source and load
impedance of 135 Ω resistive for single-carrier modulation systems and DFE-based PAM signals. Loop loss shall
be calculated with source and load impedance of 100 Ω resistive for ADSL and RADSL.
A.3 Compatibility with voicegrade services and technologies
A.3.1 Description of voicegrade services and technologies
Voicegrade services and technologies use the frequency spectrum from 0 to 4 kHz and often employ various types
of dc and ac signaling. There are several types of voicegrade signals and the impact of crosstalk interference varies
depending upon the type of disturbed signal. For example, voice systems are concerned about the subjective effects
of background noise during silent intervals when no speech is present, whereas analog voiceband data systems are
concerned about the signal-to-noise ratio during data transmission.
Voicegrade services and technologies transmit signals that can be placed into one of five general categories:
– speech signals
– single and dual tone signals
– low frequency signals
– digital data
– analog data.
A.3.1.1 Speech signals
Speech signals include live voice as well as recorded announcements. Most of the speech energy is in the frequency
range from 300 to 3000 Hz. The most sensitive speech receiver is the human ear. It has been found that background
noise during silent intervals when no speech is present is the most disturbing noise to the average listener.
Background noise is measured with a C-message weighting filter that simulates the effects of the average human ear
with a 500-type telephone set. A background noise level of 20 dBrnC or less is considered to be acceptable.
A.3.1.2 Single and dual tone signals
Single and dual frequency tones are used as network control and addressing signals, call progress signals, and
alerting signals. Network control and addressing signals include dual-tone Multi-frequency (DTMF) signaling,
multi-frequency (MF) signaling, single frequency (SF) signaling, and coin deposit signals. Call progress signals
include dial tone, busy tone, reorder tone, audible ring, special information tones, and receiver off-hook tone. Call
waiting tone is an example of a single frequency alerting tone that is used with a supplemental feature on analog
access lines.
Single and dual-tone signals range in frequency from 440 Hz to 2600 Hz and require signal-to-noise ratios on the
order of 16 to 28 dB for reliable detection.
A.3.1.3 Low frequency (< 100 Hz) signals
Ringing, maintenance signals, and sub-voice data systems are examples of signals that use low (< 100 Hz)
frequencies. The actual frequency range of the various signals is from about 17 to 83 Hz. These signals have a
relatively high tolerance for noise compared to other voicegrade signals.
A.3.1.4 Digital data
Digital data sub-rates use voiceband frequencies. The lowest digital data rates are entirely within the voiceband.
Digital data at 2.4 kbps has nulls at 0 and 2.4 kHz with maximum power at 1.2 kHz. Digital data at 3.2 kbps has
nulls at 0 and 3.2 kHz with maximum power at 1.6 kHz. Digital data rates at 4.8 kbps and above use bandwidths
that are wider than the 4 kHz voiceband. For example, the 4.8 kbps digital data signal has nulls at 0 and 4.8 kHz
with energy concentrated at 2.4 kHz.
The maximum loop loss at a frequency equal to one-half of the data rate for any digital data service is 31 dB for 135
Ω terminations. The minimum signal-to-noise ratio that provides acceptable performance is 20 dB.
A.3.1.5 Analog data
Several types of analog data are used in the loop environment. The most common types are:
– Low-speed frequency shift keying (FSK) associated with supplemental network features such as Calling
Number Delivery, Calling Name Delivery, and Visual Message Waiting Indicator.
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– Customer data using one of the ITU-T standards such as V.34 or V.90.
The network-originated FSK data messages associated with network supplemental features on analog access lines
generally require a signal-to-noise ratio of at least 25 dB. V.34 modems require a signal to noise ratio of 39 dB.
V.90 modems are the most sensitive voiceband data modems requiring a 50 dB signal-to-noise ratio to operate at the
maximum speed. The high signal-to-noise ratio makes the V.90 modem the most sensitive of all of the voicegrade
technologies.
A.3.2 Voicegrade evaluation
Because of the subjective effects of speech crosstalk, particularly intelligible crosstalk, special consideration must be
given to crosstalk between loops that carry speech signals. In addition, voiceband signals that have narrow spectral
characteristics also require complicated evaluations to determine the subjective effects of single frequency crosstalk
interference on a human listener. This standard does not provide guidance for evaluating the subjective effects of
speech crosstalk or single frequency interference. This standard assumes that the transmission system under
evaluation is a DSL system that has spectral energy that is dispersed across a portion of the voiceband and that the
crosstalk noise from such a system will have a Gaussian noise distribution.
The voicegrade spectral compatibility evaluation assumes that the V.90 modem is the victim technology. If the DSL
system under evaluation passes this evaluation, then it is unlikely that crosstalk interference problems will result
with the other, more robust, types of voicegrade systems.
It is convenient to evaluate V.90 performance in terms of the total crosstalk noise power that occurs in the frequency
band from 0 to 4 kHz.
A.3.2.1 Evaluation loop
No evaluation loop is needed.
A.3.2.2 Reference crosstalk environment
Spectral compatibility evaluations that use the V.90 modem as the victim technology shall assume forty-nine
disturbers in a 50-pair binder group.
A crosstalk model as described in clause A.2.2 is used for evaluation. A 49-disturber model that has 66 dB of loss at
20 kHz and a linear (log-log) slope of -4 dB per decade can be expressed as:
NEXT49 = 10 log10 ( f 2 / 5 ÷ 2.11× 108 )
where f is in Hz from 200 to 20,000.
A.3.2.3 Crosstalk noise and peak power levels computation
Evaluations shall be performed in both the upstream and downstream directions. The DSL system under evaluation
shall be considered spectrally compatible with the V.90 modem, and voicegrade services and technologies in
general, if the NEXT caused by 49-disturbers in the same binder group meets the voiceband NEXT PSD and total
voiceband NEXT noise objective. The DSL system under evaluation shall be considered spectrally compatible with
voicegrade services and technologies in general, if the NEXT caused by 49-disturbers in the same binder group
meets the voiceband NEXT PSD requirement and total voiceband NEXT noise requirement.
A.3.2.3.1
Voiceband NEXT PSD
The NEXT PSD at any frequency from 200 to 4,000 Hz caused by 49-disturbers on a victim pair in the same binder
group shall not exceed -97.5 dBm/Hz. To determine compliance, the 200 to 4,000 Hz PSD of the system under
evaluation is passed through the 49-disturber crosstalk model. The resultant NEXT power level for each frequency
is compared to the requirement
PSDD + 10 log 10 ( f 2 / 5 ÷ 2.11× 10 8 ) ≤ −97.5 dBm / Hz .
The voiceband NEXT PSD requirement is met by any DSL system that has a transmitted PSD which is less than -29
dBm/Hz across the frequency band from 200 to 4000 Hz.
If the voiceband NEXT PSD requirement is not met, the system under evaluation has failed to demonstrate spectral
compatibility with the V.90 modem, and voicegrade systems in general.
A.3.2.3.2
Total voiceband NEXT noise limit
The total NEXT noise on a victim pair caused by 49-disturbers in the same binder group should not exceed -75 dBm
(15 dBrn). To determine compliance, the 200 to 4,000 Hz PSD of the system under evaluation is passed through the
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49-disturber crosstalk model at each frequency and the NEXT noise at each frequency is then summed on a power
basis. The resulting total voiceband NEXT noise is then compared to the objective:
∫ PSDD ( f ) × [f
4000
10 log10
2/ 5
]
÷ 2.11× 108 df ≤ −75dBm ;
200
where the PSD is expressed in linear units (e.g., mW/Hz).
This objective is met by any DSL system that has a transmitted PSD that is less than -41 dBm/Hz across the
frequency band from 200 to 4000 Hz.
If the total NEXT power level objective of ” -75 dBm is not met, the system under evaluation has failed to
demonstrate spectral compatibility with the V.90 modem. In order to demonstrate spectral compatibility with
voicegrade systems in general, the total NEXT noise in the frequency band from 1 to 4000 Hz on a victim pair
caused by 49-disturbers in the same binder group shall not exceed -66 dBm (24 dBrn). This requirement is met by
any DSL system that has a transmitted PSD less than -32 dBm/Hz across the frequency band from 200 to 4000 Hz.
A.3.3 Spectral compatibility of voicegrade systems with basis systems
The FCC has adopted rules and regulations in Part 68 for CPE in order to protect the network from harm. One of
the harms recognized by the FCC is crosstalk interference. The FCC has adopted signal power limitations and
transverse balance limitations to prevent crosstalk interference from being caused by voicegrade CPE.
CPE that meets the voice or voiceband data signal power limitations in Part 68 will have spectral compatibility in
the voiceband with all of the basis loop transmission systems listed in clause 4.3.1.
Likewise, network equipment that meets the encoded analog content specifications in Part 68 will have spectral
compatibility in the voiceband with all of the basis loop transmission systems listed in clause 4.3.1.
A.4 Compatibility with Enhanced Business Services
A.4.1 Description of Enhanced Business Services
Enhanced Business Services use the frequency spectrum from 0 to 10 kHz and are used to transport speech signals
in the same way as done by traditional voicegrade services. A signaling channel is also present that performs all
functions associated with the setting up and tearing down of voice calls without the use of high voltage signaling.
A.4.1.1 Speech signals
The speech signals are carried in the 200 Hz to 4 kHz band in the same way as done by voicegrade services
described in clause A.3. Compatibility with speech signals must be assessed in the same manner as described in
clause A.3.
A.4.1.2 Signaling functions
Signaling functions required to set up or tear down a call, and also to transmit information required to implement
service features such as Caller Id Display are transported over a digital signaling channel. Data transmission over
that channel is performed by modulating an 8-kHz carrier.
A.4.2 Enhanced Business Service Evaluation
Because the impact of the DSL system under evaluation on the speech signal of the victim Enhanced Business
Service line has been evaluated following the criteria for voicegrade services evaluation, only the impact on the
signaling channel needs to be assessed.
A.4.2.1 Evaluation loop
No evaluation loop is needed.
A.4.2.2 Reference crosstalk environment
Spectral compatibility evaluations that use the Enhanced Business Services as the victim technology shall assume
forty-nine disturbers of the DSL system under evaluation in a 50-pair binder group.
A crosstalk model as described in clause A.2.2 is used for evaluations. A 49-disturber NEXT model that has 66 dB
of loss at 20 kHz and a linear (log-log) slope of -4 dB per decade can be expressed as:
NEXT49 = 10 log10 ( f 2 / 5 ÷ 2.11× 108 )
where f is in Hz from 200 to 20,000.
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A.4.2.3 Crosstalk noise and peak power levels computation
Evaluations shall be performed in both the upstream and downstream directions if the DSL system under evaluation
uses different PSD masks for each direction. Otherwise, only one direction suffices. The DSL system under
evaluation shall be considered spectrally compatible with the Enhanced Business Services, if the NEXT caused by
49-disturbers in the same binder group meets the signaling band NEXT PSD and if the voicegrade requirements of
clause A.3 are met.
A.4.2.3.1
Signaling Band NEXT PSD
The NEXT PSD at any frequency from 6,000 to 10,000 Hz caused by 49-disturbers on a victim pair in the same
binder group shall not exceed -96.0 dBm/Hz. To determine compliance, the 6,000 to 10,000 Hz PSD of the system
under evaluation is passed through the 49-disturber crosstalk model. The resultant NEXT power level for each
frequency is compared to the requirement.
PSDD + 10 log 10 ( f 2 / 5 ÷ 2.11 × 108 ) ≤ −96.0 dBm / Hz .
The signaling band NEXT PSD requirement is met by any DSL system that has a transmitted PSD less than -29
dBm/Hz across the frequency band from 6,000 to 10,000 Hz.
If the signaling band NEXT PSD requirement is not met, the system under evaluation has failed to demonstrate
spectral compatibility with Enhanced Business Services.
A.4.3 Spectral compatibility of Enhanced Business Services with basis systems
The FCC has adopted rules and regulations in Part 68 for CPE in order to protect the network from harm. One of
the harms recognized by the FCC is crosstalk interference. The FCC has adopted signal power limitations and
transverse balance limitations to prevent crosstalk interference from being caused by voicegrade CPE.
CPE that meets the voice or voiceband data signal power limitations in Part 68 will have spectral compatibility in
the voiceband with all of the basis loop transmission systems listed in clause 4.3.
Likewise, network equipment that meets the encoded analog content specifications in Part 68 will have spectral
compatibility in the voiceband with all of the basis loop transmission systems listed in clause 4.3.
A.5 Compatibility with T1.410
T1.410-1992 (alternatively known as the Digital Data System or DDS) operates at rates from 2.4 kbps to 64 kbps,
symmetrically, using simplex transmission over two non-loaded wire pairs. It is the primary means for low rate
connections for Frame Relay service, and is still quite popular, with over 200,000 new installations each year. While
56 or 64 kbps service is primarily used for Frame Relay, there still is a significant deployment of sub-rate (2.4, 4.8
or 9.6 kbps) service for automated teller machines and lottery networks.
T1.410 uses 50% duty-cycle AMI transmission, similar to that of T1. The main lobe of the transmitted spectrum lies
in the frequencies between DC and the signaling rate, with the peak at ½ the bit rate. As specified in the standard,
the transmitter filter is 1st order, with a 3 dB point at 1.3 times the signaling rate. (At rates below 19.2 kbps, some
additional filtering is present.) Maximum transmitted power is 6 dBm into 135 Ω, except at the 9.6 kbps rate,
where the transmitted power is limited to 0 dBm (both numbers computed for equal-probable 0s and 1s, since
T1.410 does not employ data-randomizing scramblers). For single channel service up to 56 kbps, the signaling rate
is the same as the service rate. For a service rate of 64 kbps, the signaling rate is 72 kbps. Optionally, at rates of 56
kbps and below, a secondary channel is present, which increases the signaling rate by approximately 30%.
T1.410 specifies that transceivers operate on loops where the insertion loss at 1/2 the signaling frequency can be no
more than 34 dB. Additional loop deployment practices limit the length of bridged taps that can be present on the
line. At rates below 19.2 kbps, single and total bridged tap lengths are limited to 6-kft. At rates of 19.2 kbps and
above, the total bridged tap length is limited to 2.5-kft with no single bridged tap exceeding 2.0-kft.
A.5.1 Computation of DDS Performance – Margin Computation for AMI Transceivers
DDS uses AMI transmission with a 50% duty cycle. Historically, the receivers have used a rather simple structure,
which incorporates a linear equalizer with only a single zero, and a 3rd order low-pass filter (See [2]).
The optimal (from a minimum mean squared error perspective) linear receiver for a 50% duty cycle pulse can be
obtained through the procedure described in [3]. For DDS, the resulting equalized channel resembles a 60% raised
cosine channel, which rolls off much faster than the third order low-pass filter suggested in [2].
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Since bipolar violations are used as control codes, the DDS receiver is not able to fully exploit the correlation in the
AMI signal for maximum performance. To derive the optimum receiver margin, we assume a 2 level signal, and
then increase the required SNR to compensate for the power difference between the AMI and 2 level signals. (In
fact, the result is nearly the same as we get if the correlation is taken into account.) Starting from the work in [2],
the minimum mean squared error is given by
MSE =
where
fbaud / 2
1
M( f )
df ,
fbaud −fbaud / 2 M( f )L( f ) + 1
∫
L( f ) is defined as:
∞ G( f − fbaud × n ) × H ( f − fbaud × n ) 2
L( f ) = ∑
N( f − fbaud × n )
n = −∞
2
and M( f ) is the spectrum of the message sequence ( sin x shaped for AMI). G( f ) represents the transmitted
pulse shape; for AMI this includes the 50% duty cycle and any other filtering (1st order for DDS) and is defined in
T1.410. The transmitted signal spectrum of DDS (without correlation) is:
2
  π f 

 sin
2 f 0  
1


S( f ) = K
  π f   1 + ( f (1.3 f 0 ))2
 
 
  2 f 0  
(
)
Watts/Hz ,
f 0 is the signaling rate and K is a constant. For the 9.6 kbps, 9.6 kHz signaling rate, K = 1/(3634868).
For the 64 kbps, 72 kHz signaling rate K = 1/(6848000).
where
H( f ) is the loop insertion gain transfer function, and N( f ) is the noise power spectrum. Assuming M ( f ) = 1 ,
this can be reduced to the familiar margin equation for linear equalization:
 1
M argindB = −10∗ log 10
 fbaud
fbaud / 2

1
∫− fbaud / 2 f _ SNR( f ) + 1 df  − SNR _ reqdB
where f _ SNR , the folded SNR, is given by
∞
f _ SNR =
∑
n = −∞
=
∞
∑
M 0 G ( f − fbaud × n ) × H ( f − fbaud × n )
N ( f − fbaud × n )
S ( f − fbaud × n) × H ( f − fbaud × n)
N ( f − fbaud × n)
n = −∞
2
2
.
To account for the transmitted power increase caused by the AMI correlation, we increase the required SNR by 3 dB
(the power difference for a ternary signal compared to a binary signal with the same level separation). Then for a
10-7 error rate, the SNR _ reqdB for a pseudo-optimum AMI receiver is approximately 17.3 dB.
Since actual receivers have additional impairments (mis-equalization, timing jitter, etc.), the actual required SNR is
often higher than the 17.3 dB listed here. In addition, since the actual receive filters don’t roll off as fast at the
optimal receiver, additional noise power may reach the decision device, reducing the actual SNR from that
theoretically calculated. These conditions noted, we present this calculation as the basis for the relative performance
measures to be used in this clause.
A.5.2 Evaluation loops
The maximum metallic loop loss for T1.410 is 34 dB at ½ the signaling frequency (Nyquist frequency.) Loop loss
shall be calculated assuming 135 Ω terminations. Because DDS transceivers use linear equalization, both upstream
and downstream scenarios use the worst case loops listed below:
−
56
For the 72 kHz signaling rate, the Nyquist frequency is 36 kHz. 13-kft of 26-AWG is representative of a worst
case loop, and is used for 64 kbps evaluation.
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−
ANSI T1.417
For the 9.6 kHz signaling rate, the Nyquist frequency is 4.8 kHz. 27-kft of 26-AWG is representative of a 34
dB loop, and is used for 9.6 kbps evaluation.
A.5.3 Reference crosstalk environment
T1.410 is deployed today in the same loop plant with T1.601 BRI. BRI is the worst existing disturber for DDS. To
assess the effect of crosstalk from new technologies on DDS, a relative comparison will be made with BRI crosstalk.
If a new technology produces the same or higher margins than that obtained with BRI crosstalk, then it is deemed
compatible with DDS. The SM class 1 template will represent BRI crosstalk.
Since DDS only transmits on 1 of 2 pairs in use, spectral compatibility studies that use DDS as the disturber
technology should assume 24 disturbing DDS systems in a 50-pair binder group.
The reference crosstalk environment against which new technologies shall be compared is 49 spectrum management
class 1 disturbers. The two-piece NEXT model described in clause A.2.2.1.2 is to be used for crosstalk into DDS
due to the low frequency nature of the signal.
A.5.4 Margin computation
DDS margin is computed as described in clause A.5.1 for AMI signals. For the new technology to be considered
spectrally compatible with DDS, the following scenarios must produce margins no lower than that computed using
the same loop and noise-coupling models with 49-spectrum management class 1 disturbers:
a) 49 new technology NEXT/FEXT
b) 24 SM class 1 NEXT/FEXT + 24 new technology NEXT/FEXT.
Noise spectra should include -140 dBm/Hz background noise added to the NEXT/FEXT PSD. DDS evaluations at
9.6 kbps and 64 kbps are sufficient to ensure spectral compatibility with all DDS rates.
Required SNR (SNR_req) for DDS is 17.3 dB. The transmitted signal spectrum used in the calculation is that of a
50% duty cycle bipolar signal, balanced about DC (50% positive pulses, 50% negative pulses) and passed through a
1 pole filter with 3 dB point at 1.3 times the signaling rate. Transmitted power is 6 dBm for 64 kbps 72 kHz
signaling rate, and 0 dBm for 9.6 kbps. For modeling purposes, frequency resolution, f delta ≤ 100 Hz , should be
used for 64 kbps 72 kHz signaling rate DDS margin calculations, and f delta ≤ 20 Hz for 9.6 kbps DDS margin
calculations due to the narrow bandwidth of the signal.
A.6 Compatibility with BRI
Using the transmitted spectrum for BRI described in Annex B of T1.413-1998, spectral compatibility with BRI is
verified by performing an Optimal DFE margin calculation for DFE-based PAM signals to determine BRI margin in
the presence of the proposed signal. The remainder of this clause defines the test parameters.
A.6.1 Evaluation loops
Evaluation should be performed on the shorter of either (a) the longest single length of 26-AWG copper that the
proposed technology will operate, or (b) 17.5-kft of 26-AWG.
A.6.2 Reference Crosstalk environment
The reference crosstalk environment against which new technologies shall be compared is:
− 49-spectrum management class 1 template (self-NEXT) disturbers.
The two-piece NEXT model described in clause A.2.2.1.2 is to be used for crosstalk into BRI due to the low
frequency nature of the BRI signal.
A.6.3 Margin Computation
BRI margin is computed as described for DFE-based PAM signals. The computed margin for BRI against the
proposed technology as a disturber should be compared against a calculation using the same loop and noise coupling
models for:
a) 49 new technology NEXT/FEXT
b) 24 SM class 1 Template NEXT/FEXT + 24 new technology NEXT/FEXT.
Noise spectra should include -140 dBm/Hz background noise added to the NEXT/FEXT PSD. Spectral
compatibility requires that computations for the maximum allowable numbers of the proposed technology (based on
self-crosstalk limitations) disturbing BRI produce no lower margins than 49 SM class 1 template NEXT.
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Required SNR (SNR_req) for BRI is 21.4 dB. The baud rate fbaud for the 2B1Q BRI signal is 80 kHz. For
modeling purposes, frequency resolution, f delta ≤ 100 Hz , should be used for BRI margin calculations.
A.7 Compatibility with HDSL
Using the transmitted spectrum for HDSL described in Annex B of T1.413-1998, spectral compatibility with HDSL
is verified by performing an Optimal DFE margin calculation for DFE-based PAM signals to determine HDSL
margin in the presence of the proposed signal. The remainder of this clause defines the test parameters.
A.7.1 Evaluation loops
Evaluation should be performed on the shorter of either (a) the longest single length of 26-AWG copper that the
proposed technology will operate, or (b) 9-kft of 26-AWG.
A.7.2 Reference crosstalk environment
The reference crosstalk environment against which new technologies shall be compared is:
− 49 SM class 3 template disturbers.
The simplified NEXT model described in clause A.2.2.1.1 shall be used for crosstalk evaluation.
A.7.3 Margin computation
HDSL margin is computed as described for DFE-based PAM signals. The computed margin for HDSL against the
proposed technology as a disturber should be compared against a calculation using the same loop and noise-coupling
models for
a) 49 new technology NEXT/FEXT
b) 24 SM class 3 template NEXT/FEXT + 24 new technology NEXT/FEXT.
Noise spectra should include -140 dBm/Hz background noise added to the NEXT/FEXT PSD. Spectral
compatibility requires that computations for the maximum allowable numbers of the proposed technology (based on
self-crosstalk limitations) disturbing HDSL produce no lower HDSL margins than 49 SM class 3 template NEXT.
The baud rate fbaud for the HDSL signal is 392 kHz. Required SNR (SNR_req) for HDSL is 21.4 dB. For
modeling purposes, frequency resolution, f delta ≤ 500 Hz , should be used for the HDSL margin calculations.
A.8 Compatibility with ADSL and RADSL technologies
ADSL compatibility is inherently more complicated than for fixed-rate technologies. Compatibility with ADSL
must consider performance levels at different loop reaches, as appropriate to the deployment reach of the technology
being evaluated as a disturber to ADSL. This clause addresses T1.413-1998, CAP/QAM RADSL and ITU
Recommendations G.992.1.12
A.8.1 Compatibility with ADSL
This clause presents methodology that computes spectral compatibility with T1.413-1998 (non-overlapped
upstream/downstream spectra with the reduced NEXT transmitted spectra of T1.413-1998 annex F), and G.992.1
(also with non-overlapped upstream/downstream spectra).
A.8.1.1 Parameters for computing spectral compatibility with ADSL
Spectral compatibility with ADSL is evaluated with the DMT margin calculation in clause A.2.6. Evaluations are
performed for ADSL as described in T1.413-1998, with non-overlapped upstream/downstream spectra with the
reduced NEXT transmitted spectra of annex F.
The sum of NEXT plus FEXT plus -140 dBm/Hz background noise is in all calculations. ADSL remote
transmission units and remote disturbers are co-located. The simplified NEXT model described in clause A.2.2, and
the FEXT model described in clause A.2.2.2, are used. 26-AWG loops with no bridged taps are used in all cases.
The parameters for the loop model are described in clause A.2.1.
Source and load impedance of 100 Ω resistive is used for all loop terminations. The ADSL passband transmitted
PSD is -40 dBm/Hz downstream, and -38 dBm/Hz upstream. There is 3 dB of net coding gain.
12
ADSL compatibility criteria is based on G.992.1 and not T1.419. Spectral compatibility with T1.419 basis systems
will be addressed in issue 2 of the Spectrum Management standard.
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For modeling purposes, frequency resolution f delta = 4312.5 4 Hz . Tones 6 through 31 inclusive are used for
upstream calculations. Tones 33 through 255 inclusive are used for downstream calculations. Tones which are
normally reserved for DMT pilot frequencies are included as data-bearing. The combination of frequency spacing
and tone selection results in frequency ranges from (6-3/8)*4312.5 to (31+3/8)*4312.5 Hz for upstream calculations,
and (33-3/8)*4312.5 to (255+3/8)*4312.5 Hz for downstream calculations.
The theoretical bit capacity is calculated at each frequency as a real number. The capacity density in bits per Hz is
clipped to a maximum value of 14 at each frequency, and truncated to 0 if the bit per Hz capacity at that frequency is
less than 2. Then the bit capacities of all frequencies are summed, and the total sum is multiplied by 1000 to
generate the calculated rate. (1000 represents the 4 kHz data symbol rate divided by 4 frequencies per tone.) The
calculated rate is not reduced to account for the overhead channel (AOC and EOC), nor is it rounded. The bit rates
here and in clause A.8.1.2 represent "Aggregate Data Rates" as defined in ITU-T Recommendation G.992.1.
A.8.1.2 Evaluation loops and performance levels
Three performance levels of ADSL are determined:
A) 4850 kbps downstream, 645 kbps upstream at a reach of 9-kft of 26-AWG
B) 3095 kbps downstream, 415 kbps upstream at a reach of 11.5-kft of 26-AWG
C) 425 kbps downstream, 105 kbps upstream at a reach of 15.5-kft of 26-AWG.
Performance Level A Evaluation loops: For both the upstream and downstream directions, evaluation shall be done
first on performance level A. If performance level A passes, then evaluation is done on performance level B. If
performance level A fails, the data rate algorithm in clause A.8.1.4 shall determine the target data rate for ADSL
below the 9-kft loop length of performance level A. The 26-AWG loop length is reduced until both the upstream
and downstream target data rates are met with no less than 6 dB of margin. The loop length is then rounded to the
nearest 500-ft increment to determine the deployment guideline.
Performance Level B Evaluation loops: For both the upstream and downstream directions, evaluation shall be done
on performance level B only if performance level A passes. If performance level B passes, then evaluation is done
on performance level C. If performance level B fails, the data rate algorithm in clause A.8.1.4 shall determine the
target data rate for ADSL below the 11.5-kft loop length of performance level B. The 26-AWG loop length is
increased from 9-kft while both the upstream and downstream target data rates are met with no less than 6 dB of
margin. The loop length is then rounded to the nearest 500-ft increment to determine the deployment guideline.
Performance Level C Evaluation loops: For both the upstream and downstream directions, evaluation shall be done
on performance level C only if performance level A and B pass. If performance level C passes, then the technology
is considered spectrally compatible on all loops. If performance level C fails, the data rate algorithm in clause
A.8.1.4 shall determine the target data rate for ADSL below the 15.5-kft loop length of performance level C. The
26-AWG loop length is increased from 11.5-kft while both the upstream and downstream target data rates are met
with no less than 6 dB of margin. The loop length is then rounded to the nearest 500-ft increment to determine the
deployment guideline.
A spectrum management class or technology uses the method described above in order to determine the loop length
at which it is considered spectrally compatible with ADSL. Note that in order to accurately determine the
deployment guideline of a particular technology, the increase or decrease of loop length in the method described
above must be done in steps no larger than half the rounding interval value of 500 feet.
A.8.1.3 Reference crosstalk environments
Upstream and Downstream:
Performance Level A: 24 SM class 3 template NEXT/FEXT disturbers.
Performance Level B: 24 SM class 2 template NEXT/FEXT disturbers.
Performance Level C: 24 SM class 1 template NEXT/FEXT disturbers.
A.8.1.4 Margin computation
A test for spectral compatibility with ADSL passes only if the computed margins of both upstream and downstream
ADSL for both crosstalk cases a) and b) (given below) with the proposed new technology as a crosstalker, equal 6
dB or greater. This is equivalent to checking that all the ADSL bit rates with 6 dB margin with the proposed new
technology as a crosstalker are equal to or greater than the ADSL bit rates specified for the performance level. All
loop lengths are EWL in-kft.
This is a draft document and thus, is still dynamic in nature.
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Downstream
Performance Level A:
Loop Length (kft, EWL)
L≤9
Data Rate (kbps)
4850
a) 24 new technology NEXT/FEXT
b) 12 SM class 3 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Performance Level B:
Loop Length (kft, EWL)
9 < L ≤ 10
10 < L ≤ 11.5
Data Rate (kbps)
4595 + 255 × (10 − L )
3095 + 1000 × (11.5 − L )
a) 24 new technology NEXT/FEXT
b) 12 SM class 2 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Performance Level C:
Loop Length (kft, EWL)
11.5 < L ≤ 11.8
11.8 < L ≤ 13.0
13.0 < L ≤ 14.0
14.0 < L ≤ 15.5
Data Rate (kbps)
3095
2045 + 875 × (13.0 - L)
1265 + 780 × (14.0 - L)
425 + 560 × (15.5 - L)
a) 24 new technology NEXT/FEXT
b) 12 SM class 1 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Upstream
Performance Level A:
Loop Length (kft, EWL)
L≤6
6< L≤9
Data Rate (kbps)
950
644 + 102 × (9 - L)
a) 24 new technology NEXT/FEXT
b) 12 SM class 3 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Performance Level B:
Loop Length (kft, EWL)
9 < L ≤ 11.5
Data Rate (kbps)
415 + 92 × (11.5 - L)
a) 24 new technology NEXT/FEXT
b) 12 SM class 2 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Performance Level C:
Loop Length (kft, EWL)
11.5 < L ≤ 13.0
13.0 < L ≤ 15.5
Data Rate (kbps)
202 + 142 × (13.0 - L)
104.5 + 39 × (15.5 - L)
a) 24 new technology NEXT/FEXT
b) 12 SM class 1 template NEXT/FEXT + 12 new technology NEXT/FEXT.
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A.8.2 Compatibility with RADSL
Single carrier RADSL per TR-59 uses the same FDD PSD as ADSL per T1.413-1998. The spectral compatibility
conditions for ADSL are defined in clause A.8.1. Since both RADSL and ADSL use the same PSD, the spectral
compatibility conditions for the two systems are equivalent. Spectral compatibility conditions for RADSL are
required as per clause A.8.1.
A.9 Compatibility with HDSL2
Using the transmitted spectrum for HDSL2 (PSD Mask 1) described in T1.418, spectral compatibility with HDSL2
is verified by performing an Optimal DFE Margin calculation for DFE-based PAM signals to determine HDSL2
margin in the presence of the proposed signal. The remainder of this clause defines the test parameters.
A.9.1 Evaluation loops
Evaluation should be performed on the shorter of either (a) the longest single length of 26-AWG copper for which
the proposed technology operates, or (b) 9-kft 26-AWG.
A.9.2 Reference crosstalk environment
The reference crosstalk environment against which new technologies shall be compared is:
Downstream:
Upstream:
24 T1 template disturbers (defined in Annex B of T1.413-1998) and 24 SM class 4 template
disturbers.
24 SM class 3 template disturbers and 24 SM class 5 template disturbers.
The simplified NEXT model described in clause A.2.2.1.1 shall be used for crosstalk coupling.
A.9.3 Margin computation
HDSL2 margin is computed as described for DFE-based PAM signals. The HDSL2 margin with the proposed
technology as a disturber should be compared against calculations under the reference crosstalk scenarios, with the
same loop and noise coupling models used in each case. The following crosstalk combinations, using the loop
topologies described in clause A.9.1, should all be used to compute margins with the proposed technology as a
disturber:
Downstream:
(a) 49 new technology NEXT/FEXT,
(b) 24 G.shdsl at LBR=1848 as per clause 5.4.2 (G.shdsl_1848) + 24 new technology NEXT/FEXT,
(c) 24 new technology NEXT/FEXT + 24 SM class 4 template NEXT/FEXT,
(d) 12 G.shdsl_1848 + 12 SM class 4 template NEXT/FEXT + 24 new technology NEXT/FEXT.
Upstream:
(a) 49 new technology NEXT/FEXT,
(b) 24 SM class 3 template NEXT/FEXT + 24 new technology NEXT/FEXT,
(c) 24 new technology NEXT/FEXT + 24 SM class 5 template NEXT/FEXT,
(d) 12 SM class 3 template NEXT/FEXT + 12 SM class 5 template NEXT/FEXT + 24 new technology
NEXT/FEXT.
Spectral compatibility requires that the computed HDSL2 margin, using each of the eight test crosstalk
combinations specified above, is not more than HDSL2_delta dB lower than the HDSL2 margin for the
corresponding reference case. The values of HDSL2_delta for the various test crosstalk combinations shall be as
specified in Table A. 8. The comparisons shall be done under the following conditions:
−
Required SNR ( SNR _ req ) for HDSL2 is 27.7 dB – 5.1 dB for coding gain (i.e. 22.6 dB).
−
For modeling purposes, frequency resolution, f delta ≤ 500 Hz , should be used for the HDSL2 margin
calculations.
Noise floor is -140 dBm/Hz.
FSAN combination method for mixed disturbers as described in clause A.2.2.3.
−
−
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The simplified T1E1 NEXT model is used for the noise-coupling model.
A.10 Compatibility with 2B1Q SDSL
Using the transmitted spectrum described in clause A.10.4.1, spectral compatibility with 2B1Q SDSL is verified by
performing an optimal DFE calculation for DFE-based PAM signals to determine SDSL margin in the presence of
the proposed signal.
A.10.1 Evaluation loops and performance levels
Three performance classes of 2B1Q SDSL are determined:
1. 2B1Q SDSL at 400 kbps at reaches up to 13.5-kft.
2. 2B1Q SDSL at 1040 kbps at reaches up to 8.5-kft.
3. 2B1Q SDSL at 1568 kbps at reaches up to 7.0-kft.
Evaluation needs to consider all of the three performance classes regardless of downstream or upstream direction.
A.10.2 Reference crosstalk environment
The reference crosstalk environments against which new technologies shall be compared are:
a. 49 SM class 2 template disturbers for 2B1Q SDSL at 400 kbps.
b. 49 SM class 8 template disturbers for 2B1Q SDSL at 1040 kbps.
c. 49 SM class 7 template disturbers for 2B1Q SDSL at 1568 kbps.
The simplified NEXT model described in clause A.2.2.1.1 shall be used for crosstalk coupling.
A.10.3 Margin computation
2B1Q SDSL margin is computed as described for DFE-based PAM signals. The computed margin for 2B1Q SDSL
against the proposed technology as a disturber should be compared against a calculation using the same loop and
noise coupling models for
a) 49 new technology NEXT, and
b) 24 reference disturbers + 24 new technology NEXT.
Noise spectra should include -140 dBm/Hz background noise added to the NEXT/FEXT PSD. Spectral
compatibility requires that computations for the maximum allowable numbers of the proposed technology (based on
self-crosstalk limitations) disturbing 2B1Q SDSL produce no lower 2B1Q SDSL margins than 49 reference NEXT
by 0.5 dB at 400 kbps, 0.7 dB at 1040 kbps, and 0.6 dB at 1568 kbps.
Required SNR for SDSL is 21.4 dB. f delta ≤ 500 Hz should be used for 2B1Q margin calculation.
A.10.4 2B1Q SDSL Technology Specification
This subsection defines SDSL technology; it does not define spectral compatibility tests.
2B1Q SDSL uses 4-PAM modulation. Bit rate, baud rate, and power spectral density at both TU-C and TU-R
transceivers are the same. 2B1Q SDSL system may vary its data rate from 64 kbps to 2320 kbps. The granularity of
data rate is not specified, but is expected to be in the range of 8 kbps to 64 kbps.
A.10.4.1 Power Spectral Density
The power spectral density of 2B1Q SDSL systems for the TU-C or TU-R can be approximated by filtering a square
pulse at the symbol rate followed by a 4 th order Butterworth filter with 3dB cutoff frequency equal to 240/392 of
the symbol rate. It is described by:
  πf

sin

f sym  
2 .7 × 2 .7


×
SDSLu ( f ) =
135 × f sym  πf

f sym 


2
8
 
 
 
 
f
 
1 + 
  240 f  
  392 sym  


.
where f sym is the symbol rate. The PSD of 2B1Q SDSL at several data rates are plotted in Figure A. 5 and Figure
A. 6.
62
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
The actual PSD may differ from this template specification. However, this specification is adequate to determine
spectral compatibility with 2B1Q systems.
A.10.4.2 Performance
Spectral compatibility with 2B1Q SDSL at 160 kbps is demonstrated by showing spectral compatibility with BRI
and spectral compatibility with 2B1Q SDSL at 784 kbps is demonstrated by showing spectral compatibility with
HDSL. Performance in this clause does not apply to 2B1Q SDSL at these two data rates.
At a given data rate, performance of 2B1Q SDSL is specified as a target reach on a test loop in the presence of
crosstalk noise. At the target reach, SDSL transceivers shall achieve 10-7 bit error ratio (BER) with 3 dB of noise
margin. The required SNR at 10-7 bit error ratio for 4-PAM signal with 0 dB of noise margin is 21.4 dB. This
clause describes the test loop, test setup, crosstalk noise, and reach target.
A.10.4.2.1 Test loops
Test loop lengths are in kilofeet of Equivalent Working Length (e.g., 26-AWG, with no bridged taps). Test loops
are defined in clause A.10.1. The parameters for the loop model are described in clause A.2.1.
A.10.4.2.2 Test Setup
Test setup is the same as for the HDSL2 noise impairment test given in T1.418.
A.10.4.2.3 Crosstalk noise
The simplified NEXT model as described in clause A.2.2.1.1 shall be used.
The crosstalk noise for 2B1Q SDSL at both TU-C and TU-R is specified as 49 Self-NEXT ( n = 49 ). The PSD
used for producing NEXT noise is specified in clause A.10.4.1.
A.10.4.2.4 Reach target
2B1Q SDSL shall have the target reach specified in clause A.10.1 in the presence of crosstalk source specified in
clause A.10.4.2.3.
A.10.4.3 Return loss
The minimum return loss with respect to 135Ω over a frequency band of 1 kHz to 1 MHz shall be 12 dB from 40
kHz to f sym / 2 , with a slope of 20 dB/decade below 40 kHz and above f sym / 2 . An example of minimum return
loss for 784 kbps system is shown is Figure A. 7.
A.10.4.4 Longitudinal Balance
2B1Q SDSL system shall meet the following longitudinal requirement:
−
40 dB between 20 kHz and f sym / 2 , with a slope of –20 dB/decade below 20 kHz and above f sym / 2 .
As an example, the requirement for a 784 kbps system is shown in Figure A. 8.
A.11 Customer end-point separation
It is often the case that TU-R are in separate locations, and in this case the NEXT from a remote upstream
transmitter is attenuated by a length of cable before it couples into a remote downstream receiver. To account for
this customer end-point separation, it is reasonable to assume that the NEXT from remote transmitters is attenuated
by a 150 foot section of 24-AWG cable before this NEXT couples into the remote receiver. Furthermore, this model
assumes the customer end-point separation is not cumulative, and is equal to a fixed 150 foot length between the
disturbed receiver and every crosstalker. Numerous and detailed calculations during the course of this standards
development have indicated that utilizing this customer end-point separation model does not change compatibility
results much from the case of a simplified zero end-point separation (collocated customer end-points). Therefore,
the simplified zero end-point separation model shall be used.
This is a draft document and thus, is still dynamic in nature.
63
T1E1.4/2000-002R6
ANSI T1.417
Table A. 1 - Curve fit parameters for 26-AWG twisted-pair cable
Resistance
r0c
ac
(value)
Inductance
286.176 Ω/km
0.14769620
l0
l∞
b
(value)
Capacitance
675.369 µH/km
488.952 µH/km
0.929
(value)
Conductance
49 nF/km
g0
ge
(value)
43 nMho/km
.70
fm
806.339 kHz
c∞
Table A. 2 - Constants for Two-Piece NEXT Model
Number of
Disturbers
1
10
24
49
64
U0
-11
6.598 x 10
-10
7.071 x 10
-9
1.925 x 10
-9
4.782 x 10
U1
U2
0.6
0.5
0.45
0.4
8.881 x 10
-14
9.518 x 10
-13
1.580 x 10
-13
2.391 x 10
U3
-15
1.5
1.4
1.4
1.4
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table A.3 - Code for DFE PAM SNR computation
/* OPTIMAL DFE PAM SNR computation */
float pamsnr (
float
*signal, /* array of received signal psd samples (resolution = FDELTA Hz)*/
float
*noise, /* array of received noise psd samples (resolution = FDELTA Hz) */
int
baud, /* PAM baud rate expressed in units of FDELTA (frequency resolution) */
int
end, /* Maximum number of frequency samples */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in linear units */
{
int
i = 0, cnt = 0;
float snr,temp = 0.0;
while((i<end) && (i<baud)) {
if (in_dB == 1) {
if ((2*(baud-i)) < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
pow(10.0,0.1*(signal[2*baud-i]-noise[2*baud-i]))+
1.0);
} else if (i+baud < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
1.0);
} else if (baud-i < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
1.0);
} else {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);
}
} else { /* i_dB != 1 */
if ((2*(baud-i)) < end) {
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i] +signal[i+baud]/noise[i+baud] +
signal[2*baud-i]/noise[2*baud-i] + 1.0);
} else if (i+baud < end){
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i] + signal[i+baud]/noise[i+baud] + 1.0);
} else if (baud-i < end){
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i]+1.0);
} else {
temp += log(signal[i]/noise[i] + 1.0);
}
}
cnt++;
i++;
}
temp /= (float) cnt;
snr=10.0*temp*log10(exp(1.));
return(snr);
/* dB */
}
This is a draft document and thus, is still dynamic in nature.
65
T1E1.4/2000-002R6
ANSI T1.417
Table A. 4 - Code for DFE QAM/CAP computation
/* OPTIMAL DFE QAM/CAP SNR computation */
float qamsnr (
float
*signal, /* array of received signal psd samples (resolution = FDELTA Hz)*/
float
*noise, /* array of received noise psd samples (resolution = FDELTA Hz) */
int
baud, /* PAM baud rate expressed in units of FDELTA (frequency resolution) */
int
end, /* Maximum number of frequency samples */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in linear units */
{
int
i = 0, cnt = 0;
float snr, temp = 0.0;
while(i<end && i < baud) {
if( in_dB == 1 ) {
if (i+3*baud < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
pow(10.0,0.1*(signal[i+2*baud]-noise[i+3*baud]))+
pow(10.0,0.1*(signal[i+3*baud]-noise[i+3*baud]))+
1.0);
} else if (i+2*baud < end){
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
pow(10.0,0.1*(signal[i+2*baud]-noise[i+2*baud]))+
1.0);
} else if (i+baud < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
1.0);
} else {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);
}
} else {
if (i+2*baud < end){
temp += log(signal[i]/noise[i] +
signal[i+baud]/noise[i+baud]+
signal[i+2*baud]/noise[i+2*baud] +1.0);
} else if (i+baud < end){
temp +=log(signal[i]/noise[i] +
signal[i+baud]/noise[i+baud]+1.0);
} else {
temp += log(signal[i]/noise[i] +1.0);
}
}
cnt++;
i++;
}
temp /= (float) cnt;
snr=10.0*temp*log10(exp(1.));
return(snr);
/* dB */}
66
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table A.5 - Matlab code to set-up ADSL margin computation
-----------------------------------------------------------------------------% ADSL_margin.m
% This program shows how to set up the various parameters required to compute
% the DMT margin as per the spectrum management standard.
% Define_Xmit_PSD, Define_Loop_Function and Define_NEXTFEXT_Noise are user
% supplied functions
% Emphasis has been put on code portability rather than code efficiency
% set the direction, bit rate and loop length of the ADSL system
Direction = ’DN’; % either UP or DN
BitRate = 5184e3; % in bps
LoopLength = 9000; % in feet
% assumes that all signal computations are in the linear domain for the margin
% computation. This parameter should not be changed
in_dB = 0;
% set up the number of carriers and specifies the xmit PSD.
% The Xmit PSD will be the template corresponding to SMC5.
if (strcmp(upper(Direction),’DN’))
% carriers 33 to 255
CarrierStart = 33;
CarrierEnd = 255;
Carriers = [CarrierStart:CarrierEnd];
XmitPsd=’ADSL-DN’;
elseif (strcmp(upper(Direction),’UP’))
% carriers 6 to 31
CarrierStart = 6;
CarrierEnd = 31;
Carriers = [CarrierStart:CarrierEnd];
XmitPsd=’ADSL-UP’;
else
error([’Invalid direction,’ direction])
end;
% set up the Coding gain, min and max number of bits per carrier, number of bits
% per symbol and frequency separation of the ADSL carriers
CODING_GAIN = 3.0;
MINBITS = 2;
MAXBITS = 14;
CarrierSpacing=4312.5;% Hz takes into account cyclic prefix
NPointsPerCarrier=4; % number of frequency points per carrier
BitsPerSymbol= BitRate/(4000/ NPointsPerCarrier); % baud rate is 4000 symbols/second
Deltaf=CarrierSpacing/NPointsPerCarrier;
% define frequency vector using NPointsPerCarrier frequency points distributed
% uniformly over each carrier
if (rem(NPointsPerCarrier,2)==0) % check remainder of a division by 2
% NPointsPerCarrier is even
StartFreq=CarrierStart*CarrierSpacing - (NPointsPerCarrier-1)*Deltaf/2;
EndFreq = CarrierEnd*CarrierSpacing + (NPointsPerCarrier-1)*Deltaf/2;
else
% NPointsPerCarrier is odd
StartFreq=CarrierStart*CarrierSpacing - (NPointsPerCarrier-1)/2*Deltaf ;
EndFreq = CarrierEnd*CarrierSpacing + (NPointsPerCarrier-1)/2*Deltaf ;
end;
Freq=StartFreq:Deltaf:EndFreq;
% define Xmit PSD
% psd_xmit is a vector that contains the value of the PSD corresponding
% to each frequency point defined in Freq. The units should be watts/Hz
% i.e. 10*log10(psd_xmit)+30 is in dBm/Hz.
This is a draft document and thus, is still dynamic in nature.
67
T1E1.4/2000-002R6
ANSI T1.417
% The Xmit PSD of ADSL should be the template of SMC5 either UP or DN
% i.e. -40 dBm/Hz for example in the passband of the downstream direction
psd_xmit = Define_Xmit_PSD(Freq,XmitPsd)
% define cable function
% loop is a vector that contains the magnitude squared of the insertion loss
% corresponding to each frequency point defined in Freq. The loop function
% should be on a linear scale.
% The termination impedance should be 100 Ω.
% See clause A.2.1 for insertion loss
% See Table A. 1 for cable parameters
loop = Define_Loop_Function(Freq,AWG26_length,termination_impedance)
% signal PSD is xmit PSD attenuated by loop
signal = psd_xmit .* loop;
% background noise at -140 dBm/Hz converted to a linear scale of Watts/Hz
Background_Noise = 1e-3.*(10.^(-140/10));
% NEXTFEXTNoise is a vector that contains the value of the NEXT plus FEXT PSD
% corresponding to each frequency point defined in Freq. The units should be watts/Hz
% see clause 5.3 and associated tables and figures for a description of the SM classes.
NEXTFEXTNoise = Define_NEXTFEXT_Noise(Freq,etc,...)
% noise = NEXT + FEXT + Background noise
noise = NEXTFEXTNoise + Background_Noise;
[snr_margin resolution]= dmtmrgnTA3(signal, noise, BitsPerSymbol, 1,length(Freq), in_dB, CODING_GAIN, MINBITS,
MAXBITS);
% display the results used in the margin computation
fout=1; % redirect to std output
fprintf(fout,’\nADSL
PARAMETERS:
C.G.:
%ddB
Carriers:
%d-%d
MinBits:
%d
%d’,CODING_GAIN,Carriers(1),Carriers(length(Carriers)),MINBITS,MAXBITS);
fprintf(fout,’\nSIMULATION PARAMETERS:’);
fprintf(fout,’\nNPointsPerCarrier: %d Freq. Resolution: %5.2f Hz’,NPointsPerCarrier,Deltaf);
fprintf(fout,’\nADSL Direction: %s BitRate: %dbps LoopLength: %dfeet’,Direction,BitRate ,LoopLength );
fprintf(fout,’\nADSL MARGIN (resolution: %3.2fdB) : %2.1fdB\n’,resolution,snr_margin);
68
This is a draft document and thus, is still dynamic in nature.
MaxBits:
T1E1.4/2000-002R6
ANSI T1.417
Table A. 6 - Matlab Code to compute a DMT margin
function [snr_margin, MARGIN_STEP]= dmtmrgnTA3(signal, noise, rate, f_start, f_end, in_dB, CODING_GAIN,
MINBITS, MAXBITS)
% signal -> psd of signal
% noise -> psd of noise
% rate -> # bits per symbol
% f_start -> index of starting freq.
% f_end -> inded of ending freq.
% in_dB = 1 if PSD is in dB or in_dB = 0 if PSD is in linear units
% CODING_GAIN -> self explanatory
% MINBITS -> min number of bits per carrier
% MAXBITS -> max number of bits per carrier
% snr_margin <- computed DMT margin with a resolution of MARGIN_STEP
% Assumes that the margin is MAXIMUM_VALUE - MARGIN_STEP
% Starts a brute force search from this point downward
% Computes the capacity and compares to the target bit per symbol rate
% If not enough, decrease the margin by MARGIN_STEP
% if enough, then has found the correct margin
% In case the margin is greater than MAXIMUM_VALUE - MARGIN_STEP,
% will add 10 dB to MAXIMUM_VALUE until finds an initial guess that
% is larger than the margin and proceeds as above.
% Emphasis has been put on code portability rather than code efficiency.
% To achieve greater speed, one can vectorize the various loops and use a root
% finding algorithm such as, for example, fzero in matlab
MAXIMUM_VALUE =7.1;
% resolution of the margin computation will affect the speed
MARGIN_STEP = 0.1;
SNRGAP = 9.75 - CODING_GAIN;
snr_margin = MAXIMUM_VALUE;
firstpass = 1;
totcap = 0;
while (totcap < rate)
snr_margin = snr_margin - MARGIN_STEP;
% compute capacity
totcap = 0;
for j = f_start:1:f_end,
if (in_dB)
snr = signal(j) - noise(j);
else
snr = 10*log10(signal(j) / noise(j));
end;
delcap = log(1.+10.^((snr - snr_margin - SNRGAP)/10))/log(2);
if (delcap > MAXBITS) delcap = MAXBITS; end;
if (delcap < MINBITS) delcap = 0; end;
totcap = totcap + delcap;
end;
if ((totcap > rate) & (firstpass == 1))
snr_margin = snr_margin +10;
totcap = 0;
else
firstpass = 0;
end;
end;
This is a draft document and thus, is still dynamic in nature.
69
70
6.
5.
1.
2.
3.
4.
N /A
1 1 .5
9 .0
1 0 .5
N /A
6 .5
7 .5
1 3 .5
N /A
1 5 .5
1 4 .5
1 3 .5
1 3 .5
1 3 .0
1 2 .5
1 2 .0
1 1 .5
1 1 .0
1 0 .5
1 0 .0
9 .5
9 .5
9 .0
8 .5
8 .5
8 .0
7 .5
7 .0
6 .5
6 .0
5 .5
5 .0
6 .0
SM 1
SM 2
SM 3
SM 4
SM 5
SM 7
SM 8
SM 9
S D S L ( < 3 0 0 k b ps )
S D S L ( 3 2 0 k b ps )
S D S L ( 3 3 6 k b ps )
S D S L ( 3 8 4 k b ps )
S D S L ( 4 0 0 k b ps )
S D S L ( 4 1 6 k b ps )
S D S L (46 4 k b ps )
S D S L (52 8 k b ps )
S D S L (59 2 k b ps )
S D S L (65 6 k b ps )
S D S L (72 0 k b ps )
S D S L (78 4 k b ps )
S D S L (84 8 k b ps )
S D S L (91 2 k b ps )
S D S L (97 6 k b ps )
S D S L (1 0 4 0 k bp s )
S D S L (1 1 0 4 k bp s )
S D S L (1 16 8 - 1 23 2 k b ps)
S D S L (1 29 6 - 1 42 4 k b ps)
S D S L (1 48 8 - 1 68 0 k b ps)
S D S L (1 74 4 - 1 87 2 k b ps)
S D S L (1 93 6 - 2 12 8 k b ps)
S D S L (2 19 2 - 2 25 6 k b ps)
S D S L (2 3 2 0 k bp s )
T1
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
V o ic e
A bs
15 .0
N /A
49
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P -P h on e
A bs
1 5 .0
N /A
49
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
DDS
R el
1 4.1
SM1
49
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
IS D N
Rel
1 7 .7
SM1
49
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
HDSL
R el
9 .0
SM3
49
P ASS
P ASS
P ASS
P ASS
P ASS
6.3
7.5
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
P ASS
8.9
8.7
8.5
8 .2 - 7 .9
7 .7 - 7 .3
7 .2 - 6 .8
6 .6 - 6 .4
6 .2 - 5 .8
5 .6 - 5 .4
5.2
6.2
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
10 .9
10 .4
9 .9
9 .6
9 .3
PASS
PASS
9 .0
10 .6
PASS
B a s is S ystem
ADSL
ADSL
A
B
9.0
11 .5
SM3
SM2
24
24
This is a draft document and thus, is still dynamic in nature.
SDSL @ 784 kbps and HDSL have equivalent results.
T1 defined in T1.413-1998.
1 3 .7
PASS
1 5 .4
1 4 .7
1 3 .6
1 3 .4
1 3 .2
1 2 .5
1 2 .1
1 1 .6
PASS
PASS
1 1 .7
ADSL
C
1 5 .5
SM1
24
Table A. 7 - Spectral compatibility with the basis systems
SM 1 results are equivalently achieved by DDS (T1.410-1992), BRI (as defined in T1.413-1998) and SDSL < 300 kbps.
SM 4 results are equivalently achieved by HDSL2 (T1.418-2000).
SM 5 results are equivalently achieved by non-overlapped ADSL (T1.413-1998, Appendix F).
SM 9 results are equivalently achieved by overlapped ADSL (T1.413-1998).
Notes:
EW L
PSD
B a sis S ystem
P e rfo rm a nc e L e ve l
E valu atio n L oo p 2 6 A W G
R e fe re n ce D is turbe r
# D isturbe rs
T1E1.4/2000-002R6
P AS S
P AS S
P AS S
P AS S
P AS S
8 .3
8 .3
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
8 .9
8 .8
8 .7
8 .6
8 .6
8 .6
8 .6
8 .6
P AS S
HD SL2
Rel
9 .0
M ixe d
2 4 /2 4
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
SDSL
1552
7 .0
SM7
49
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
P AS S
SDSL
1 04 0
8 .5
SM8
49
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
PASS
SDSL
40 0
1 3.5
SM2
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Table A. 8 - HDSL2_delta (in dB) for various test crosstalk combinations
(a)
(b)
(c)
(d)
Downstream
49 New Technology
24 G.shdsl_1848+ 24 New
24 New + 24 SMC 4
12 G.shdsl_1848 + 12 SMC 4 +
24 New
0.0
0.0
0.0
0.0
Upstream
49 New Technology
24 SMC 3 + 24 New
24 New + 24 SMC 5
12 SMC 3 + 12 SMC 5 +
24 New
0.3
0.3
0.0
0.4
Table A. 9 - 2B1Q SDSL data rate and associated spectrum management classes
2B1Q SDSL data rate (kbps)
data rate ≤ 288
288 < data rate ≤ 528
528 < data rate ≤ 784
784 < data rate ≤ 1168
1168 < data rate ≤ 1568
SMC
1
2
3
8
7
This is a draft document and thus, is still dynamic in nature.
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1
Simplified Model for 49 Disturbers
-6
10
80
24
49
-5
1% NEXT Loss - dB
-4.5
70
-4
60
slope
slope
-15
50
40
100
-14
-14
-14
-15
10000 20000
1000
100000
Frequency - Hz
Notes:
Z 0 at each frequency
1.
Terminated with cable characteristic impedance
2.
NEXT disturbers in the same cable binder unit of 50 pairs
Figure A. 1 – Two-piece NEXT model and simplified NEXT model
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Ref-C
Ref-R
TU-C
TU-R
New-C
New-R
Z kft, 26 gauge
Figure A. 2 – Crosstalk into a Basis System: NEXT and FEXT
New
NEXT
Reference
NEXT
FSAN
TU - C
Z kft
TU - R
FSAN
New
FEXT
Z kft H(f)
& Coupling
Reference
FEXT
Z kft H(f)
& Coupling
AWGN Noise
Figure A. 3 – Simulation Model for Reference and New Crosstalk into Downstream
Receiver
This is a draft document and thus, is still dynamic in nature.
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Go to Step 8
Test PSD against basis
system performance level
Yes
1. Up & downstream:
calculate target margin of
basis system with
reference disturbers only
2. Up & downstream:
calculate margin with new
technology and mix of
reference disturbers
3. New margin >
(target margin - delta) or
(fixed margin)?
No
9. Go to new
performance
level
4. Reduce loop length
11. Go to new
basis system
No
8. All
performance level
scenarios for this basis
system calculated?
No
Yes
6. New margin >
(target margin - delta) or
(fixed margin)?
Yes
No
10. All
basis systems
calculated?
7. Reduce loop length
5. Up & downstream:
calculate margin with new
technology and mix of
reference disturbers
Yes
New system is
considered
spectrally
compatible
If steps 4 to 7 are employed for any performance scenario, the new
system is considered spectrally compatible only when deployed on
loops Z’ or shorter, where Z’ is the shortest loop length required for
any basis system performance level scenario. All deployment
guidelines round Z’ to the nearest 500 ft increment.
Figure A. 4 - Process flow for spectral compatibility calculations
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Figure A. 5 – 2B1Q SDSL PSD at several data rates
Figure A. 6 - 2B1Q SDSL PSD at several data rates
This is a draft document and thus, is still dynamic in nature.
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Figure A. 7 - Minimum return loss for 784 kbps 2B1Q SDSL system
Figure A. 8 - Longitudinal balance for 784 kbps 2B1Q SDSL system
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Annex B: Loop Information
(Informative )
B.1
General
B.1.1 The loop environment
There are more than 700 million metallic twisted pair cables delivering communications services to customers
around the world. It is predicted that most of this embedded base of copper wire will eventually be replaced with
wider bandwidth transmission media such as optical fiber and coaxial cable. However, twisted pair copper wire will
be the main method of delivery for several years to come13.
Recent years have seen the increase in demand for customer information bandwidth escalate dramatically from 3
kHz analog voice services to digital services requiring several megabits per second. Advances in integrated circuit
density, digital signal processing techniques and information compression algorithms are resulting in the
introduction of ever-higher bandwidth twisted pair transmission systems that can transport these new services.
These new systems must fit into an outside loop transmission environment with several existing transmission
systems and other systems that may be introduced later. For the voice frequency services the twisting of the wire
pairs and construction of the cables such that no two pairs traveled together for very long, helped to control crosstalk
coupling. Interference between pairs was held to acceptable levels. As signal bandwidth increases, the crosstalk
coupling between pairs increases at the same time as the transmission loss increases making the circuits more
susceptible to interference. Interference can come from other transmission systems of the same kind or from
different type systems that overlap the signal spectrum.
The 1.5 Mbps T1 line system was originally developed for application in the interoffice cable plant whose
construction is very carefully controlled. To control crosstalk interference between T1 systems, T1 signals in the
two directions were placed on separate pairs located in different binder groups that had shields between them. T1
repeaters were spaced and placed to minimize differences in signal levels. When T1 systems began to be deployed
in the customer outside loop plant, the situation became much more challenging. As will be described later, the loop
plant is designed and constructed to deliver voice services to customers at acceptable quality and minimum cost. In
recent years, many T1 lines have been deployed in the outside plant to deliver 1.5 Mbps services to business
customers. The engineering design and construction of these lines is a challenge in minimizing interference and
cost.
Over the years several high-bandwidth analog carrier systems were also deployed in the outside plant with mixed
results for compatibility and interference.
The use of the loop plant to transport high rate digital signals was not envisioned at the beginning. Indeed, for over
100 years the loop plant has been optimized for the reliable delivery of voice frequency services at lowest cost and
acceptable quality. In the last several years, the design of new loop plant has been modified slightly to ease the
introduction of digital transmission.
As newer digital transmission systems have been developed for the loop plant, each one has been subjected to hard
scrutiny for potential interference with like systems. BRI systems had to account for other DSL systems in the cable
and existing systems like T1 lines and DDS. In turn, HDSL had to show compatibility with BRI, T1 and DDS.
ADSL had to account for all of the above.
Development and deployment of these new transmission systems is very costly in time and money. It would be very
desirable to predict how a system will perform before it is actually built. Testing a system against all reasonable
cases of interferers on all reasonable loop configurations is not feasible. To test the performance of a system and the
mutual interference with other systems, a combination of analysis, simulation, laboratory and field testing is done.
Analysis and computer simulations are the first steps in developing new systems. Needed are accurate models for:
the transmission systems (the new proposed system as well as the other possible conflicting systems), the primary
13
Not all telephony wire is made up of twisted pairs. Some utilize non-staggered twist (NST) cables which do not use
different lengths of twist among the constituent pairs and are prohibited for use as T1 lines by telephone company
practices. Some single-pair aerial/overhead drop wire uses parallel/flat/non-twisted conductors for lengths up to 700
feet. Also, not all of the wire is copper. Copper-clad steel and copper-cadmium mixes are used where strength is
needed in drop cables. Aluminum wires have also been deployed in some areas.
This is a draft document and thus, is still dynamic in nature.
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transmission constants of the cables, and the crosstalk coupling for the frequency spectrum, representative set(s) of
test loop topologies, a reasonable set of interferer systems (types, numbers and combinations), broadband
background (thermal) noise models, impulse noise models, etc.
B.1.1.1 Background noise
According to one study, the residence background noise level in the band of interest could be at a level of around 140 dBm/Hz. This background noise level is higher than that achievable by a receiver front-end electronic circuit.
On the other hand, attention still has to be paid to make the receiver front-end circuit noise level below the assumed
-140 dBm/Hz level.
B.1.1.2 Impulse noise
Impulse noise is of major concern for high-speed twisted wire-pair systems, especially due to the higher subscriber
loop loss. Compared with the very weak received signal, a majority of measured impulses from one study would
cause receiver detection error. It has been shown that forward error correction coding is effective at minimizing the
impact of impulse noise. The effect of impulse noise needs to be included in transmission performance simulation.
Forward error correction codes are typically used to handle impulse noise.
B.1.1.3 Radio frequency interference (RFI)
In addition to coupling within a cable, radio frequency interference (RFI) also becomes a concern as the signal
frequency increases, the wavelength shortens and approaches the dimensions of the cable structure components, and
overlaps radio services. Radio frequency energy may radiate from a wire pair and interfere with radio services
(egress). Radio frequency energy may enter a wire pair and interfere with the wire pair transmission system
(ingress).
Modern wire pair systems operate with signals in a "metallic" mode where currents in the two conductors are equal
and opposite in direction thus tending to reduce radiation either entering or exiting. Currents that travel on both
conductors in the same direction are said to be in a "longitudinal" mode. These longitudinal currents are much more
likely to radiate. External radio frequency fields tend to couple to the pair in the longitudinal mode. The balance of
the individual wire pair conductors and the connecting circuitry relative to the environment determines the
conversion of the normal metallic signal conduction to longitudinal currents and the conversion of longitudinal
currents to metallic signals.
B.1.1.4 Structural cable faults
Structural cable faults (degraded splices, shorts, opens, grounds, crossed pairs, conductor pair reversals, etc.) do
occur and will prevent a transmission system from working. Such mechanical cable faults are beyond the scope of
this document.
B.1.1.5 The loop environment
Early digital twisted pair transmission systems needed to have cables with very simple make-ups. T1 carrier system
was originally intended for the interoffice cable plant to replace the loaded cable voice-frequency pairs. Interoffice
voice-frequency cable used only one gauge of wire. No bridged taps were allowed. The T1 repeater spacing
matched that of the loading coils starting at 3000 feet and at intervals of 6000 feet afterwards to take advantage of
the loading coil mounting locations between central offices.
When T1 carrier began to be deployed in the subscriber loop plant for connections to Digital Loop Carrier systems
and for high-capacity digital services to business customers, it encountered a much tougher environment in terms of
cable makeup. Repeater spacing had to be reduced to 4000 or 3000 feet. Bridged taps had to be removed.
Modern digital twisted pair transmission systems are intended to deliver digital information to average households
and businesses through the copper loop plant as it exists without modifying the makeup. Depending on the desired
information rate and noise environment, the serving distance from a central office (CO) or a remote terminal (RT)
could be different. To have a low overall cost, the deployment procedure for a new digital transmission system
should be as simple as possible. In other words, it would be ideal if the system terminals could simply be installed
on the selected loop and turned on. Additional engineering work such as field trips and loop qualification should be
avoided. From the telephone company point of view, a service using the transmission system as a delivery vehicle
should be pre-qualified for a known type of loop plant, such as resistance design range, CSA, etc. Any loop
qualification should be on a bulk area basis, not for each individual loop.
B.1.1.6 Telephone cable and subscriber loop structures
This section describes the nature of the structure of the outside loop plant in the US.
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A subscriber loop consists of sections (typically 500 feet long) of copper twisted 14 pairs of different gauges. A
section of a subscriber loop could be aerial (hung on poles), buried (directly in the ground), or underground (pulled
inside protective conduit). Electrical joints called splices, for cable sections could be made on a telephone pole for
aerial cables or in a manhole for underground cables. These splices are not soldered as in most electronic circuits,
but are made with some form of compression technique. For many years the most common splice was made by
stripping the insulation from the wire ends, hand twisting the bare wire ends together and covering the splice with
tape. Modern splices use connectors that use a hand compression tool to generate the force to penetrate the
insulation and make a solid connection. Properly performed, the compression splice results in a metal to metal
connection that is impervious to liquid or gas.
Twisted pair cables have large cross sections near the central office. There could be 12, 13, 25, 50 or 100 pairs in a
cable binder group and up to 50 binder groups per cable. Binder groups are combined to form cables of from 50
pairs to several thousand pairs. Cables share a common electrical and physical structure, with metallic electrical
sheathing and plastic covering. Cables intended for application near the customer premises may have fewer pairs.
Functionally, a subscriber loop can be divided into portions that belong to feeder cable, distribution cable, and drop
wire15. Wiring inside the customer premises that connects to the drop wire at the network customer interface does
not count as part of the network loop16. The interface between the network loop and the customer premises wiring is
usually made as close as practical to the point of entry to the premises.
For large multi-tenant buildings and campuses, the network may provide cabling past the minimum point of entry if
permitted by state regulations. Feeder cables provide links from a central office to a concentrated customer area.
Distribution cables then carry on from feeder cables to potential customer sites. Since the loop plant construction is
completed before customer service requests, distribution cables are usually made available to all existing and
potential customer sites. Hence, it is a common practice to connect a twisted pair from a feeder cable with more
than one distribution cable to maximize the probability of reaching a potential customer. These multiple
connections from a feeder or a distribution cable to more than one customer location create open-ended cable stubs
called “bridged taps”. At any one time only one customer is connected and the other taps are left open17. As
customers connect and disconnect service, these bridged tap appearances allow the operating company flexibility in
the use of the wire.
At voice frequency the transmission effects of bridged taps are relatively small and can be controlled within
acceptable limits by design. The loop plant design rules, such as Resistance Design and CSA, limit the total bridged
tap length to minimize adverse effects, mainly loss and spectrum distortion, on POTS transmission. From a
transmission line viewpoint, these bridged taps are open-ended shunts. Above the voice band the transmission
effects become more significant as the frequency increases and the signal wavelengths approach the tap lengths.
Connection points between feeder cables and distribution cables are commonly located in cabinets, called Feeder
Distribution Interfaces (FDI). Connection points in distribution cables are commonly in pedestals for underground
cables or terminals for aerial cables.
Single aerial drop wires often consist of parallel copper-clad steel wires, sometimes called "flat pairs." For new
construction in recent years, multiple (2, 4 or 6) twisted copper pairs are being used, and are buried if possible. The
drop wire is usually short and has a proportionately small effect on the loop transmission characteristics except for
potential radiation effects. A typical rule of thumb was to allow the drop wiring to be less than 700 feet or 25 Ω in
resistance.
The loop and drop wire potentially could pick up other high frequency radiation noises. It could also radiate signals
to other high frequency electronic devices.
B.1.2 Loop plant design rules: resistance design
Most of the embedded outside loop plant in the US has been constructed using the guidelines called Resistance
Design or one of its variations.
14
An exception to twisted pair cable is single-pair aerial drop wire.
The term "drop" refers to the drop downward from a pole to a house. Today, most "drop" wires in new construction
are buried.
16
Any significant lengths of customer premises wiring included before a transmission system terminal would
contribute to transmission effects. Customer premises wiring can vary from a few feet to thousands of feet in length.
The analysis here assumes that the terminal is at the interface.
17
Party lines had all the customers on the line tied to the same loop back to the Central Office.
15
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POTS loop plant design must accomplish three goals: ensure that there is sufficient direct current flow from the
network battery plant to operate station sets, allow dc/low-frequency call process signaling (dialing, ringing), and
limit transmission loss and frequency roll-off to acceptable levels. As mentioned, telephone cables are designed
with different gauges of wire from 26-AWG (thin, with higher resistance) to 19-AWG (thicker with lower
resistance). These different gauges are designed to have close to the same capacitance between conductors per unit
length (nominal 0.083 µf/mile). It happens that limiting the maximum dc resistance also controls the maximum
voice frequency loss and roll off with frequency.
For modern switching systems a maximum loop resistance (DC resistance) of 1500 Ω18 meets powering, signaling
and transmission objectives. The maximum transmission loss at 1004 Hz is about 9 dB with a roll-off of 6 dB at
2804 Hz. From survey data, the average loop has a dc resistance of 600 Ω with 4 dB of loss at 1 kHz. Since copper
resistance has a strong dependency to temperature, the maximum operating temperature is normally used to
calculate the maximum distance that can be covered by a specific wire gauge. This temperature is usually assumed
to be 100°F for buried installations and 140°F for aerial installations.
Since distances from a central office to each customer are different, distribution cables of different gauges are
utilized to keep the amount of copper (and dollars) used to a minimum while meeting design guidelines. To reduce
overall loop resistance the end sections of a long subscriber loop tend to have coarser twisted pairs, whereas finer
gauge twisted pairs are used closer to the central office in order to reduce the diameter of cables in crowded ducts
and minimize cost.
However, some customers are so far away from the central office that even while using proper wire gauge such as to
keep the dc resistance within the specified 1500 Ω, the voice-frequency loss at 1 kHz and roll-off across the voicefrequency band is such that it would result in poor voice channel service quality. A procedure of installing loading
coils and coarser gauge cables has been used to extend the central office serving distances for the voice channel.
Inductive loading results in a loop with reduced loss within the voice band for a given gauge of cable and acts as a
low pass filter above 3000 Hz19. The original rule for loading cable was 18-kft working length excluding any
bridged taps.20 Under the current rules (revised resistance design), loading coils are installed for cables with a total
length exceeding 18-kft including bridged tap. For the most common loading plan, called "H88" with 88 mH
inductors, the first loading coil is installed at 3-kft from the central office. Loading coils are installed every 6-kft
thereafter. There may be no bridged taps between loading coils. Bridged tap on the end sections at the central office
and customer ends may be left connected, up to a total tap length of 6-kft.
B.1.3 Loop plant design rules: carrier serving area (CSA)
DLC systems were originally developed to serve POTS customers beyond the Resistance Design range. Early DLC
systems are based on copper twisted pairs using the 1.5 Mbps T-carrier, T1-Line technology. Twenty-four voice
channels are carried on one T1-line by use of time division multiplex. With the use of outside plant digital
repeaters/regenerators it is possible to reach out 100 miles. Fiber based DLC systems are now more popular.
Depending on the cost of DLC electronics, it becomes more economical to serve customers with DLC systems
beyond a certain distance. This "prove in" distance has been decreasing as DLC electronics costs have come down.
The concept of Carrier Serving Area (CSA) engineering guidelines was originally developed in the early 1980’s to
support 56 kbps Digital Data Service (DDS) delivery to customers served by DLC systems. The concept was then
revised very slightly and has been used as the guide for voice grade special services and POTS deployment from the
18
Different types of switching equipment have different dc loop resistance limits, depending on the battery feed
voltage, the feed resistance, the typical set resistance and the desired minimum current. Step-by-step (SXS)
switches with nominal 48 Vdc batteries (and a 41.5 V emergency minimum) typically have a 1300 Ω design limit to
achieve a minimum of 23-mAdc through a rotary dial 500-type station set with about 150 Ω dc resistance. Thus,
references are common to "1300 Ω Resistance Design" even though SXS switches have been retired in all major
operating companies. The revised resistance design rules allow an increase of the 1300 Ω dc resistance limit to
1500 Ω. Newer electronic switches typically have a 1500 Ω loop design limit from a nominal 52 Vdc battery, for a
minimum of 20 mAdc through 400 Ω dc Touchtone station sets.
19
Transmission analysis shows that loss is independent of frequency for a particular ratio, termed Heaviside’s
condition for a distortionless line, of resistance, conductance, capacitance and inductance. Normal cable has a small
inductance relative to resistance and capacitance. Lumped inductive loading achieves close to the ratios for flat
cable loss within the voice band.
20
The length of the cable that connects directly from the network to the customer, excluding any bridged taps, is
called the "working length." The working length of the cable corresponds to the dc resistance path from the network
battery to the customer interface. Bridged taps are open-circuited to dc flow.
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DLC remote terminal. A CSA is roughly defined as a serving distance of 9-kft for 26-gauge loops and 12-kft for 24gauge loops from a DLC remote terminal. The term is also applied to loops that originate from a central office as
well if they meet CSA guidelines. Short loops around a central office may be consistent with CSA rules even
though constructed using Resistance Design rules. A recent (1991) survey shows that over 60% of DLC loops meet
the CSA guidelines. (References in this document to "CSA" loops or "CSA-type" loops mean wire pairs that meet
CSA design guidelines whether they originate from a central office or from a network remote terminal site.)
Carrier serving area wire pairs from the remote terminal of a DLC system to the network interface on the customer’s
premises are expected to meet the following design guidelines.
a) Non-loaded cable only
b) Multi-gauge cable is restricted to two gauges (excluding short cable sections used for stubbing or fusing).
c) Total bridged tap length may not exceed 2.5-kft. No single bridged tap may exceed 2.0-kft.
d) The amount of 26-gauge cable (used alone or in combination with another gauge cable) may not exceed a
total length of 9-kft including bridged tap.
e) For single gauge or multi-gauge cables containing only 19, 22 or 24-gauge cable, the total cable length
including bridged tap may not exceed 12-kft.
f)
The total cable length including bridged tap of a multi-gauge cable that contains 26-gauge cable may not
exceed
12 −
3(L26)
kft
9 − LBTAP
where L26 is the total length of 26-gauge cable in the cable (excluding any 26-gauge bridged tap) and
LBTAP is the total length of bridged tap in the cable. All lengths are in kilofeet (kft).
The limits defined above are the maximum permissible outer bounds for a CSA. Nothing in the CSA concept
prohibits the restriction of CSA cables to shorter lengths.
CSA guidelines do not include central office wiring on the switch side of the protector frame, drop wire or customer
building wiring. Detailed statistics for central office wiring or customer premises are not available. Central office
cabling is typically 24-or 26-gauge and may be up to 1-kft long. Customer premises wiring is typically 26-gauge
and may be up to 1-kft or more. In some installations, some drop and wiring inside the customer premises may be
part of the network. Although not a transmission requirement, it is suggested that no more than two gauges of cable
be used. Note: all wire gauge references in this document are American Wire Gauge (AWG).
B.1.4 Distribution area (DA)
A CSA is often further divided into 1 to 6 Distribution Areas (DA). A DA is characterized by a single Feeder
Distribution Interface (FDI) where cross-connects are located. A DA typically serves about 500 customers. The
cable pair group from a RT to all DAs could have different service capacity than that of all DAs combined.
Distribution cables emanating from an FDI usually have a 1.5 to 2 pairs for all potential customer living units. On
the other hand, cable pairs from RT to FDI are installed based on the number of real customer lines with a smaller
spare ratio. This strategy is aimed at an overall minimized installation cost. The average serving distance of each
DA is usually significantly shorter than that of a CSA.21 A recent (1991) survey shows that most DA distribution
loops are less than 6 to 8-kft in length (26-and 24-gauge respectively) or about 2/3 of the maximum CSA lengths.
B.1.5 Loop statistics
The Resistance Design and Carrier Serving Area design do not define how much of each type of cabling is actually
used. Major surveys of loop topology in the old Bell System were conducted in 1976 and 1983.
B.2 AWG and metric cable: diameters and DC resistance and capacitance
T1E1.4 and ETSI have developed test loop sets for AWG and metric cables for BRI, HDSL, ADSL and VDSL. It is
sometimes useful for interested parties who are familiar with one set of cables, but not the other, to make a rough
judgment on which cable in one set compares to which cable in the other set, if any. One can get into the right
ballpark or at least out of the wrong one, by comparing conductor diameters, DC (0 Hz) resistance and low
frequency capacitance and insulation materials. Table B.1 summarizes this data for the most common types of
21
However, the maximum serving distance of a DA might still be very close to that of a CSA.
This is a draft document and thus, is still dynamic in nature.
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metric and AWG telephony cables. Non-telephony 18 and 20-AWG gauges are also included for comparison
because their conductor diameters are close to 0.8 mm and 1.00 mm metric cables.
Attenuation versus frequency data (say at 1 kHz, 10 kHz, 100 kHz, 1 MHz, 10 MHz and 30 MHz) would allow
further contrasts and comparisons. Polyethylene is the most common insulation for feeder and distribution cables.
Polyethylene is a very good dielectric whose properties change very little with frequency. PVC is the most common
insulation for single-pair, overhead/aerial drop wires exposed to the external environment. PVC dielectric properties
vary much more with frequency than those of polyethylene.
Only 26-AWG PIC and metric 0.40 PE are reasonably close in transmission characteristics.
B.3 Cable primary constants (RLCG) characterization
It is not feasible to perform laboratory or field tests to represent all likely environments that a transmission system
will encounter. Computer simulation provides a means to test schemes against anything that can be quantified
numerically. Fundamental to the simulation of wire systems are accurate models of the transmission characteristics
of the wire itself versus frequency and temperature.
The primary constants of resistance (R, Ω/km), inductance (L, µH/km), capacitance (C, nF/km), and conductance
(G, Mho/km) are used to model most transmission lines. Secondary parameters such as impedance, attenuation and
phase or the chain parameters ABCD may be calculated from the primary constants. These "constants" actually vary
in value with frequency, temperature and humidity. To a first order, signal attenuation increases as the square root
of the frequency. Variation of the "constants" and inductive reactance becoming larger with frequency relative to
resistance and capacitance result in the actual attenuation versus frequency curve being more complex.
Chain parameters ABCD allow cascading of models of two port electronic devices such as wire pairs. Complex
loop topologies with changes of gauge and bridged taps can be constructed with ABCD matrices. See Figure B.1.
The existing primary constant RLCG models of the common AWG PIC cables were based on careful measurements
and curve-fitting in the early 1970s. They were believed to be valid to 10 MHz and to represent nominal values for
expected manufacturing variations. VDSL and newer proposed schemes may well have spectral components to 30
MHz. It is vital to have models that reflect the transmission behavior of the cables in the real world to the frequency
and temperature ranges needed.
The primary constant data can be presented in either as R, L, C and G values versus frequency or as parameters to
equations that have been curve fitted to measured data.
B.3.1 Transmission-Line Characterization
This section directly addresses the transmission characteristics of twisted-pair phone lines.
Most twisted-pair phone lines can be well modeled for transmission at frequencies up to at least f < 30 MHz by
using what is known as two-port modeling or “ABCD” theory. Such ABCD theory is well covered in basic
electromagnetic texts, but is often not in a form convenient for use in DSL systems. Werner presented essential
results of such translation to DSL systems in a 1991 JSAC paper “The HDSL Environment” (August 1991) and this
section essentially repeats that effort, but provides more detail along with updates based on various studies in
standards bodies that have led to DSL characterization to at least 30 MHz.
Section B.3.1.1 first describes ABCD modeling in general before section B.3.1.2 specializes to the case of twistedpair transmission lines. Section B.3.1.5 considers the special case of bridged taps before section B.3.1.6 shows how
to compute the transfer characteristics of a subscriber loop consisting of many sections. Section B.3.1.7 shows how
to measure RLCG parameters for loop characterization as well as lists models for several popular twisted-pair types.
B.3.1.1 “ABCD” modeling
Figure B. 2 shows a general two-port linear circuit. There is a voltage at each port and a current on the upper path
on each port. The voltages and currents will depend on the source (port 1) and load (port 2) impedance and voltage
source(s), but nevertheless always relate to each other by the matrix relationship:
V1   A B  V2 
V2 
 =
 ⋅ I  = Φ ⋅ I 
I
C
D
  2
 1 
 2
or
V1 = AV2 + BI 2
I1 = CV2 + DI 2
where Φ is a 2 × 2 matrix (nonsingular in all but trivial situations not of interest) of 4 possibly frequency-dependent
parameters, A, B, C, and D, which all depend only on the network and not on external connections. The quantities
have circuit definitions as in the table below:
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A
open-load voltage ratio
B
shorted-load impedance
C
open-load admittance
D
shorted-load current ratio
The transformation is reversed by Φ −1 so that
V2 
1
 =
 I 2  AD − BC
 D − B  V1
−1 V1

 ⋅ I  = Φ ⋅ I  .
− C A   1 
 1
When Φ = I , that is an identity, the network is a trivial connection of the upper path and lower path across the
network, essentially meaning there is no network. A relationship of interest is the complex frequency response
T (f ) =
V2
V1
where the frequency dependence is shown explicitly for T(f), but not for the other voltages to simplify notation.
V
This ratio depends on the load impedance attached at port 2, ZL = Z 2 = 2
.
I2
T (f ) =
1
A+B
=
ZL
ZL
A ⋅ ZL + B
can be related to a transfer function between an input voltage supply VS (with finite internal impedance ZS ) and
the output voltage VL = V2 (across a load ZL = Z 2 ):
VL ( f ) VL ( f ) V1 ( f )
Z1
=
⋅
=
⋅T( f ),
VS ( f ) V1 ( f ) VS ( f ) Z1 + Z S
V
where Z1 = 1
is the input impedance of the terminated two-port. Z1 must be computed as in the second equation
I1
below and is the ratio of input voltage to current when load ZL is attached at the output.
A cascade of two-ports has a two-port matrix that is the product, in order, of the two ports
V1
VN 
VN 
  = Φ1 ⋅ Φ 2 ...⋅ ΦN −1 ⋅   = Φ ⋅  
 I1 
 IN 
 IN 
,
which allows the calculation of transfer functions and insertion losses of more complicated networks as long as a
two-port model can be found for each subsection in the cascade. The inverse is found by reversing the order and
taking the product of the inverse matrices. The input impedance of the two-port is
V
Z1 = 1 =
I1
B
AZL + B
ZL
.
=
D
CZL + D
C+
ZL
A+
Two-port networks are very useful in the analysis of twisted-pair transmission lines as in the next several sections.
In these sections, the transmission line is modeled as a cascade of two ports that are characterized by resistance,
inductance, capacitance, and conductance per unit length, and by the length of the transmission-line segment.
B.3.1.2 Transmission-line RLCG characterization
The two-port characterization of a transmission line derives from the per-unit length two-port model in Figure B.3.
The R, L, C, and G parameters represent resistance, inductance, capacitance, and conductance per unit length of the
transmission line.
A segment of transmission line can be viewed as a cascade of such sections that are infinitesimally small in length.
At any point x, the two-port voltages and currents relate through the differential equations
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dV
= (R + jωL ) ⋅ I
dx
dI
−
= (G + jωC ) ⋅ V
dx
−
at any given frequency ω = 2πf . V and I are phasor quantities representing peak amplitudes of sinusoids at
frequency f (or amplitudes of the complex exponential e j 2πft ). The R, L, C, and G parameters themselves can vary
with frequency, but are presumed constant with respect to length at any given frequency in the analysis to follow.
This set of differential equations is equivalent to the pair of second-order differential equations
d 2V
dx 2
d 2I
= γ 2 ⋅V
,
= γ2 ⋅ I
dx 2
where
γ = α + jβ =
(R + jωL ) ⋅ (G + jωC ) =
Z ⋅Y
is the frequency-dependent propagation constant for the twisted pair, and characterizes the segment of transmission
line. The impedance per unit length, Z, and the admittance per unit length, Y, are also defined in Figure B.3. The
attenuation constant is α and the phase constant is β. The attenuation constant is very important for twisted-pair. As
can be inferred from equations to come, the attenuation of a twisted-pair is approximated by 8.686 α dB per unit
length at the frequency of interest. The phase constant is related to speed of propagation on the twisted pair: At each
frequency ω = 2πf , a sinusoid propagates on the twisted pair with phase given by
θ(ω, x ) = ωt − βx
and has envelope amplitude attenuated as e − αx . The wavelength is the length (at fixed frequency and time) over
which the sinusoid undergoes a full cycle and is thus given by
λ=
2π
.
β
Remembering that β is tacitly a function of frequency, different frequencies thus have different wavelengths. The
sinusoidal wave at frequency appears to propagate along the twisted pair at phase velocity
vp =
ω
,
β
and the phase delay per unit length at this same frequency is τ p = 1v = β ω . When β is a linear function of
p
frequency, the channel is said to have linear phase and the phase velocity and delay are constant over all frequencies.
An example is the case where R=G=0, and then β = ω LC - and (when L and C are constant with respect to
frequency) means that all frequencies move at the same phase velocity v p = 1
. Such a transmission line is
LC
said to be dispersionless. Note that it is possible to subtract out the linear (proportional) part of β without
introducing error to the time domain response of a cable pair only when there is no reflected wave being propagated
along the pair. Such a condition (no reflection) occurs only when the impedance of the load is matched to the
characteristic impedance of the cable pair. This may be particularly important when modeling bridged taps. In
practical DSL systems, dispersionless transmission never occurs and different frequencies travel with different
velocities, leading to dispersion of signal energy (and to the intersymbol interference). For a dispersive transmission
line, it is of interest to investigate the speed at which a group of frequencies centered around propagates. To
understand this concept of “group” or “envelope” velocity, suppose one investigates the differing speeds of the two
frequencies, ω ± ∆ω ,where the offset or difference is small and the corresponding values of β ± ∆β , but both have
the same amplitude. The resultant sum waveform is
A cos[(ω + ∆w )t − (β + ∆β )x ] + A cos[(ω − ∆w )t − (β − ∆β)x ] = 2 A cos[∆w ⋅ t − ∆β ⋅ x ] ⋅ cos[ωt − βx ] .
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The right side of this equation is an “envelope-modulated” sinusoid, a product of two sinusoids. When the phase
velocity is constant and there is no dispersion, the phase velocity of the first term on the left side is the same as that
of the second term, and the phase velocity equals the group velocity. However, when phase velocity is not constant,
the first term moves at a different (often much slower) speed given by ∆ω / ∆β . This slower speed is the group
velocity and in general computed by the inverse of the group delay
τg =
dβ
dω
.
Group delay in essence measures the spread in delay between the fastest and slowest moving frequencies in the
immediate vicinity of ω . The greater the group delay, the greater the dispersion in the transmission line.
The solution to the set of differential equations is easily modeled as the sum of two opposite-direction
voltage/current waves:
V (x ) = V0+ ⋅ e − γx + V0− ⋅ e γx
.
I (x ) = I0+ ⋅ e − γx + I 0− ⋅ e γx
By insertion of either of these solutions into the appropriate first-order voltage/current differential equations, the
ratio of the positive-going voltage to the positive-going current, as well as the (negative of the) ratio of the negativegoing voltage to the negative-going current is equal to a constant characteristic impedance of the transmission line
V+
V−
R + jωL
Z
.
Z0 = 0 = − 0 =
=
+
−
G + jωC
Y
I0
I0
One easily verifies that the R, L, C, and G parameters are equal to
R = ℜ{γ ⋅ Z0 }
1
ℑ{γ ⋅ Z0 }
ω
1  γ 
C = ℑ  .
ω  Z0 
L=
 γ 
G = ℜ 
 Z0 
For twisted-pair transmission and DSL, it is rare that any of these 4 parameters are zero and so simplifications in
textbooks or other developments that lead to so-called “lossless transmission lines” or “dispersion-less” transmission
are not of interest for DSL. Furthermore, these parameters are frequency-dependent for transmission lines and are
best determined by measurement as in section B.3.1.7.1.
A segment of transmission line of length d has solution VL = Vd and I L = Id and thus
VL = V (d ) = V0+ ⋅ e − γd + V0− ⋅ e γd
I L = I (d ) = I 0+ ⋅ e − γd + I 0− ⋅ e γd
.
Since the two voltage waves in each direction are related to the same-direction current waves by the common ratio
Z0 , one can solve the above two equations for V0+ and V0− to get:
(VL + IL ⋅ Z0 ) ⋅ e γd
.
V0− = 21 (VL − I L ⋅ Z0 ) ⋅ e − γd
V0+ =
1
2
By substituting these constants back into the solution in general and evaluating for the voltage and currents at x=0 in
terms of those at x=d , one obtains the following two-port representation
Z0 ⋅ sinh(γd )
 cosh(γd )
V (0 ) 
 ⋅ V (d ) .
1
=

 
cosh(γd )   I (d ) 
 I (0 )   Z ⋅ sinh(γd )
 0

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The ABCD entries can be read from the matrix, or equivalently, can be computed from the R, L, C and G values
through the relations for γ and for Z0 . Then, for a given length of transmission line d, the engineer may model
that transmission line as a single “lumped” two-port, replacing the distributed model in Figure B.3.
Knowing the load impedance so that V (d ) / I (d ) = ZL , the complex frequency response then becomes
T=
1
sech (γd )
=
.
 Z0 
 Z0 
 ⋅ sinh (γd ) 1 + 
 ⋅ tanh (γd )
cosh(γd ) + 
Z
Z
 L
 L
The input impedance of the two-port is
Z1 = Z 0 ⋅
V (0) / I (0) or
Z L + Z 0 ⋅ tanh (γd )
.
Z 0 + Z L ⋅ tanh (γd )
The input impedance of a very long line reduces to Z1 = Z0 , since tanh(γd ) → 1 for large d .
The transfer function relating the output voltage
VL VS =
Z1
T=
Z1 + Z S
VL to the input source voltage VS becomes
Z 0 ⋅ sech (γd )
.

 Z0

 Z0
ZS ⋅ 
⋅ tanh (γd )
+ tanh (γd ) + Z 0 ⋅ 1 +
Z
Z
L



 L
Thus, this type of model applies to the upper example in Figure B.5. Note also there the two-port models that
characterize the source and load. Thus, general principle of multiplying matrices when cascading two-ports can be
directly applied. If several transmission line segments with different R, L, C, and G were cascaded, then each would
have its own two-port model. This situation corresponds to connection of twisted pairs (splicing) with different
gauges.
B.3.1.3 Power for transmission lines
A sinusoid at any frequency on a transmission line represented by the phasor voltage V and phasor current I has
average (rms) power
P (f ) =
[ ]
1 ℜ VI *
2
.
Figure B.4 shows a simple phase circuit having input current I and voltage V across a load with impedance
ZL = RL + jX L . From basic circuit theory, a sinusoidal current with peak amplitude I delivers power
P (f ) =
2
1
2
2
I RL =
1
2
V
RL =
ZL
[ ]
1 ℜ VI *
2
,
thus providing interpretation for the relation in the previous equation.
Maximum power is transferred from the power supply to the load when the source impedance is the conjugate of the
load impedance in Figure B.4, ZS,opt = ZL* = RL − jX L . This corresponds to one-half the total power of the source
being dissipated in the load. An example of the use of this maximum-power-transfer result is when one investigates
the termination of a twisted-pair transmission line. To transfer maximum power from the line to the load, the load
impedance should be designed to be the conjugate of the line impedance viewed going back into the line. When the
line is long, this impedance will be the characteristic impedance of the line itself, meaning the best loading is
ZL,opt ≅ Z0* ,
meaning half the power in the line is transferred to the load (with the other half dissipated within the line itself).
Similarly, the optimum driving impedance is the conjugate of the line impedance, which again for long lines is the
characteristic impedance, so
ZS,opt = ZL,opt ≅ Z0*
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Again, half the source power will be delivered to the line. For a lossless transmission line, the half of the source
power delivered to the line is the same half of power delivered to the load. At very high frequencies, the best load
and source impedance become resistive and equal to the (real) characteristic impedance of the line.
The condition for maximum power transfer is not the same condition for elimination of reflections (see next subsection) unless the line is lossless.
B.3.1.4 Reflection coefficients
When the load impedance is equal to the characteristic impedance (and not the conjugate of the characteristic
impedance), the negative-going wave constant
V0− equals zero in the above equations. There is then no reflected
wave and all the above relationships simplify somewhat. In practice, such matching is not likely to occur, and the
solution for the differential equation at x=d has general ratio of positive-going wave to negative-going wave as
V − ⋅ e − γd
Z − Z0
.
ρ= 0
= L
+
γd
Z
V0 ⋅ e
L + Z0
This reflection coefficient is clearly zero when the transmission line is “matched” or terminated in its own
impedance, ZL = Z0 . The return loss is defined as the inverse of the reflection coefficient for any interface to a
two-port, and usually expressed as a positive quantity in decibels. This situation prevents “bouncing” of signals on
a transmission line and thus reduces the dispersion (relative delay) of signals on the line. In this case of ZL = Z0 ,
the input impedance is then also Z1 = Z0 . When the transmission line impedance is approximately real, then the
situation of no bouncing corresponds also to maximum energy transfer in section B.3.1.3 from the line into the
matched load. However, when (as usual for twisted pairs), the line characteristic impedance is complex, then
maximum energy transfer occurs when the load is the conjugate of the characteristic impedance, and thus
elimination of bouncing does not guarantee maximum energy transfer for lossy lines. On many lines as the
frequency increases, the R and G terms become negligible relative to L and C and so for these frequencies,
maximum energy transfer and elimination of bouncing occur when the load impedance is matched to
ZL = Z0 ≈ L
C
.
A similar source reflection coefficient can be written as
ρS =
ZS − Z 0
.
ZS + Z 0
This source reflection coefficient measures the reflected positive-going wave amplitude with respect to a negativegoing wave that flows into the source impedance. The return loss at the interface to between the source and line is
therefore the inverse (in dB) of the source reflection coefficient. Note that the source impedance that leads to
maximum power transfer into the line ZS = Z1* again is not necessarily the same as that leading to no reflection at
the source end. A wave launched from a source will traverse the loop with phase and group velocities, will be
reflected at one end, reflected again at the source end, and so on. This series of reflections leads to a transient on the
loop, unless the loop is terminated in load impedance equal to the characteristic impedance of the line. Again when
the line can be approximated over the used frequency range as lossless, and thus having real characteristic
impedance, then the maximum energy transfer and reduction of bouncing objectives coincide.
Formally the return loss of a transmission line is the inverse ratio of reflected power to incident power on the load
(or next section of circuitry). This return loss is simply the square of the reflection coefficient, thus
return loss = 10 log10
1
ρ
2
dB.
B.3.1.5 Characterization of a bridged tap section – a three-port
For modeling of loops, a bridged tap is a three-port section, but one of the ports appears as load impedance to the
line, between the two sections on each side of the bridged tap. Such a situation can be modeled by the two-port with
ABCD matrix shown in the last example of Figure B.5.
The impedance of the tap section Zt is computed according to the formula above for the input impedance of a
section of transmission line terminated with an open circuit ( ZL = ∞ ), which simplifies to
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ANSI T1.417
cosh(γd )
.
sinh(γd )
If the tap were not terminated in an open circuit, then the general formula for the input impedance Z1 (above) of the
section should be used.
Circuits with bridged tap on bridged tap have an impedance that is calculated by working backwards from all open
taps to points of the taps, modeled as the two-tap section impedance in parallel. The resultant impedance then
becomes a termination (load) impedance for the next section working backwards towards the main transmission pair
of interest. While perhaps tedious, the calculation process is straightforward and recursive.
B.3.1.6 Computation of transfer function
The computation of the transfer functions for twisted-pair transmission lines with multiple sections then simply
becomes a process of multiplying in cascade the corresponding two-port ABCD matrices for each section. Some
examples are provided in Figure B.5, with the corresponding two-port matrices below each example. The matrices
are multiplied left to right in the natural order of appearance in the figure. That is the overall two port is just
Φ = Φ 0 ⋅ Φ1 ⋅ K ⋅ Φ N
where the source voltage divider is modeled by the two-port
1 ZS 
Φ0 = 
.
0 1 
The final output voltage and current are related by the usual VL = IL ⋅ Z L , which allows the transfer function to be
computed from the ratio VL VS .
In the upper example of Figure B.5, a simple section of twisted pair with characteristic impedance Z0 and
propagation constant γ is modeled by the cascade of a two-port matrix description Φ1 for a length d and the source
two-port matrix Φ 0 . This upper example is straightforward application of the two-port theory. The lower example
additionally has a bridged tap section with Z02 and γ 2 of length d 2 and a second section of the transmission line
with yet a third characteristic impedance and propagation constant. The two sections of transmission line are
modeled as usual, where the impedance and propagation constant can be computed for each frequency from the
known RLGC parameters for each section. The bridged tap section is modeled as a parallel (shunt) impedance that
is computed according to the formula for an open-ended transmission line of length d 2 (if the tap were terminated,
the impedance shown need only be replaced by the more general expression for the inverse of the input impedance
of that section). The overall two-port matrix is simply the product of the 4 two-port matrices shown.
A variety of simplifications are sometimes studied assuming each section is very long and so appears to be
terminated in its own characteristic impedance, leading to expressions for the transfer function and input impedance
in various situations. While sometimes useful for interpretation, with modern day signal processing analysis tools
(for instance, Matlab, etc.), it is often easier to compute the transfer function without simplifying assumptions and
then analyze the corresponding results.
B.3.1.7 Relationship of transfer function and “insertion loss”
Transmission engineers directly measure the characteristics of a transmission line at several frequencies. It is hard
to measure the transfer function directly because of loading effects, but it is possible to measure easily the insertion
loss, from which the transfer function can be computed if load impedance and source impedance for the
measurement are known.
The insertion loss is computed using the configuration in Figure B.5 by first measuring the voltage Vno across the
load impedance Z L with the transmission line removed, and then inserting the transmission line and again
measuring VL , the voltage across the load. The insertion gain transfer function is defined as the ratio of these two
voltages:
H (f ) =
VL ( f )
Vno (f )
.
The insertion loss in dB is defined as
IL( f ) = −20 log10 H (f )
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The transfer function relating output voltage to input source voltage is
VL Vno VL
ZL
=
⋅
=
⋅ H (f ) .
VS VS Vno Z S + Z L
Note that when Z1 = ZL , meaning the line is terminated in its own impedance as often in practice, then this equation
can be rewritten in terms of the T(f) as
VL V1 VL
Z1
=
⋅
=
⋅T (f ) ,
VS VS V1 Z S + Z 1
which also then shows that in the matched-termination case, T (f ) = H (f ) . In most cases of interest in DSL, the line
is long and so the source impedance is matched to the characteristic impedance (which equals the input impedance
of the line when the line is long) and all impedances are real over the higher frequencies used for DSL transmission.
In this case, the VL VS transfer function loss (in dB) is 6 dB higher than the insertion loss.
A crucial point of note: When the VL VS transfer function is computed for a circuit using RLCG parameters, then
the insertion loss may be computed from the VL VS transfer function loss and is roughly 6 dB lower under the
approximations above. The insertion point is exactly the point at which a transmitted power constraint applies.
Thus for instance, input voltage levels computed from a power constraint for a DSL (for instance, in performance
calculation or SNR computation) undergo a channel loss that is the insertion loss, and not the VL VS transfer
function loss. A common mistake is to compute data rates and performance as if the transmitted voltage were 6 dB
lower by incorrectly using the VL VS transfer function loss instead of the insertion loss.
Transmission lines are characterized in this Appendix by four parameters, the Resistance R in Ω/km, the Inductance
L in Henrys/km, the Capacitance C in Farads/km, and the Conductance G in Mhos/km22.
The RLCG parameters in this appendix were provided by the following measurement and curve-fitting procedures.
B.3.1.7.1
Measurement Procedure
The open-circuit impedance, ZOC , and short-circuit impedance, ZSC , for a length, l , of twisted-pair transmission
line are measured versus frequency. An l = 10 m length is used for measurements below 2 MHz and an l = 1 m
length is used for measurements between 2 MHz and 30 MHz. The characteristic impedance and propagation
constant are computed from the measured impedance according to:
characteristic impedance:
Z0 = ZOC ⋅ ZSC
propagation constant:
 ZSC 
1

γ = tanh −1
 ZOC 
l


From the characteristic impedance and propagation constant, RLCG parameters can be computed as:
R = ℜ(γZ0 )
L=
1
ℑ(γZ0 )
ω
C=
1
ω
 γ 

ℑ

 Z0 
 γ 
 .
G = ℜ

 Z0 
22
Conductance can also be expressed in units of Siemens. One Siemen equals one Mho.
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Curve-fitting
Because of error in practical measurements of the impedance, the RLCG values may not follow smooth curves with
frequency so parameterized (smooth) models of RLCG are then fit to the measured values. The models are:
1
R( f ) =
1
4r4
0c
+ ac ⋅ f 2
+
1
4r4
0s
+ as ⋅ f 2
where r0c is the copper DC resistance and r0s is steel DC resistance, while ac and as are constants
characterizing the rise of resistance with frequency in the “skin effect.”
 f 
l 0 + l ∞  
 fm 
L( f ) =
b
 f 
1 +  
 fm 
b
where l 0 and I ∞ are the low frequency and high-frequency inductance, respectively and b is a parameter chosen
to characterize the transition between low and high frequencies in the measured inductance values.
C ( f ) = c∞ + c0 ⋅ f − ce
where c ∞ is the “contact” capacitance and c 0 and c e are constants chosen to fit the measurements.
G( f ) = g 0 ⋅ f g e
where g 0 and g e are constants chosen to fit the measurements.
Further information on smoothing of test measurements can be found in ASTM D 4566.
B.3.2 26-AWG Multi-pair Cable
Parameters found in Table B. 2 are computed using the models in section B.3.1.7.2 and the measurement data. The
primary constants produced using the parameters are given in Table B. 3. Additional measurements listed in Table
B. 4 and Table B. 5, have indicated that results for 26-AWG PIC cables have found strong agreement with the model
values.
B.3.3 24-AWG multi-pair cable
Parameters found in Table B. 6 are computed using the models in section B.3.1.7.2 and the measurement data.
Primary constants are found in Table B. 7. Additional measurements listed in Table B. 8 have indicated that results
for 24-AWG PIC cables are in strong agreement with the model values.
B.3.4 22-AWG Twisted Pair PIC Cable
Measurements for 22-AWG twisted pair are found in Table B. 9.
B.3.5 FP
FP is representative of ETSI 1.14 mm flat (no twists) phone-line twisted pair. Parameters computed using methods
in section B.3.1.7.2 are found in Table B. 10. Primary constants are found in Table B. 11.
B.3.6 Category-5 Twisted Pair
Cable parameters are given in Table B. 12 and Table B. 13 gives primary constants computed using methods in
section B.3.1.7.2 for cables that meet or exceed EIA/TIA Category 5 twisted-pair specifications.
B.3.7 Two-Pair Twisted Drop
Measurement data are given for a two-pair twisted service drop cable where the tip and ring are twisted for each
pair, and the two pairs are then twisted together. The conductor gauge is 22-AWG, and the tested cable was of
length 228.6 m = 750 ft. The values for R, L, C and α are averaged over the two pairs since they exhibited a high
degree of symmetry.
The measurements were made with a HP-3577A Network Analyzer connected to a HP-356711A S-Parameter test
set. Measurements at equally spaced log frequencies between 772 kHz and 40 MHz were obtained.
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C and α were directly measured with the network analyzer. The short and open complex impedances, Zsc and
Zoc of the drop cable were also measured with the network analyzer, and the characteristic impedance, Z0 ,
calculated in the usual fashion.
It was not feasible to obtain accurate measurements of the conductance G due to a lack of a precise measurement of
impedance angle. This is a result of measurement equipment limitations and the transformer baluns used to perform
the impedance conversion. Additionally, the drop wire jackets directly contact the pair insulation, hence altering the
effective dielectric constant and tan delta. Moreover, the capacitance is not flat over the entire frequency range.
Fortunately, at high frequencies, G is of little importance for transmission.
Using the relationship
Z0 =
L
C
,
which holds when G<< ω C and R<< ω L, the inductance values are calculated.
Using the relationship
α = 4.34 ∗ R / Z 0
,
the resistance values over the range 0.772 - 40 MHz are evaluated. Results are given in Table B. 14.
B.3.8 Two-Pair Quaded Drop
Measurement data are given for a two-pair quaded service drop cable where the four conductors comprising the two
pairs are twisted together as a unit. The conductor gauge is 22-AWG, and the tested cable was of length 228.6 m =
750 ft.
Test equipment, measurement setup and the equations used to perform the calculations are identical to those used in
section B.3.7 on Two-Pair twisted drop. Results are found in Table B. 15.
B.3.9 Flat-Pair Drop
Measurement data are given for a flat-pair service drop cable where the tip and ring conductors of a single pair are
parallel. The conductor gauge is 18-1/2 AWG, and the tested cable was of length 291 m = 954 ft. Test equipment,
measurement setup and the equations used to perform the calculations are identical to those used in section B.3.7 on
Two-Pair twisted drop. Results are found in Table B. 16.
B.3.10 Additional Models
For additional models for European and other types of cable, see ETSI STC TMC6 Permanent Document #
TM6(97)02.
B.4 Cable crosstalk models
Accurate models of crosstalk coupling between pairs in typical cable structures are as vital as the primary RLGC
constant models to system simulation. For the DSL family of transmission systems the limiting factor on loop range
has been crosstalk coupling of signal energy from like or unlike transmission systems on other pairs in the cable and
not from the end-to-end attenuation of the signal.
The current crosstalk models were developed in the 1980s based on computer simulations of the physical structure
of the cables and later compared with measurements. Quantitative crosstalk models for less than full binder groups
or small cables are not available.
B.4.1 Near end crosstalk, NEXT
Telephone twisted pairs are organized in binder groups of 12, 13, 25, 50 and 100 pairs. Many binder groups share a
common physical and electrical shield in a cable. Due to capacitive and inductive coupling, there is crosstalk
between each twisted pair even though pairs are well insulated at DC. The crosstalk in voice frequency band is
minimal, i.e. one can hardly hear the voice energy from an adjacent pair because the crosstalk loss is usually more
than 80 dB, compared with a voice channel loss of less than 20 dB.
In general the effect of cable crosstalk is minimized not only by the use of good insulation material between pairs
but also by adapting different twist distances among different pairs in a binder group. The binder groups are also
twisted such that no two groups are adjacent for long runs. For digital communication via digital subscriber line
technology, where the signal bandwidth reaches into the MHz range, the crosstalk is a limiting factor to the
achievable throughput.
This is a draft document and thus, is still dynamic in nature.
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Near-end-crosstalk (NEXT) is defined as the crosstalk effect between transmitter and receiver pairs at the same end
of a cable section. In other words, NEXT is a measure of the crosstalk effect between a transmitter and a receiver at
the same end of a twisted pair cable. See Figure B. 6. NEXT is usually considered for full duplex digital subscriber
line systems such as BRI and HDSL where the transmitted and receive spectra at each end are the same (or overlap).
NEXT is strongest on the cable at the point where the transmitter of the crosstalking signal puts the signal on the
pair. Any receivers near to this transmitter will receive NEXT as well as the intended signal. The NEXT path
attenuates the unintended signal greatly, but the relevant issue is the signal to noise ratio between the intended signal
and the NEXT. Therefore, NEXT becomes a problem if the intended signal is attenuated enough. Symmetrical
systems such as BRI have transmitters at both ends of every pair on which it is installed. The worst case NEXT is
then usually the NEXT produced by a binder group full of similar collocated transceivers. The NEXT received from
similar systems, i.e. BRI to BRI, HDSL to HDSL, or T1 to T1, in this way is called “self-NEXT.”
For BRI and HDSL, full duplex communication on a single pair is achieved by the use of the echo cancellation
technique. This requires transmitted and received signal paths be as fully separated as practical with signal
processing techniques23 even though transmitted and received signals share the same frequency spectra. However,
transmitted signals in other adjacent pairs are not available to the particular receiver. Thus, any energy coupled into
a pair used by a transmission system can not be effectively removed from the received signal.
For the T1 line system, bipolar (AMI) encoding of the 1.544 Mbps signal results in a transmitted spectrum centered
on 772 kHz. This 772 kHz signal is much higher in frequency than voice signals and crosstalk coupling is much
higher. For T-carrier or T1 system, the full duplex communication is based on two separate twisted pairs. In the
interoffice cable plant, special cables are used with a shield between the binder groups. T1 signals going in one
direction from all T1 systems are placed in single binder groups. All signals going the other direction are in the
other binder group with the shield between them. This binder group separation of transmit and receive pairs and
shielding greatly reduce, but does not eliminate, the NEXT effects.
In the outside customer loop plant, the special cables are not readily available. Binder group separation is practiced
as much as possible for T1 in the loop plant. Shorter repeater spacing and very careful attention to placing of
repeaters relative to other T1 systems helps compensate somewhat for the much more severe crosstalk environment
in the loop plant.
There could be 1225 different NEXT values at a particular frequency for a 50 pair binder group, assuming pair-topair NEXT is symmetrical. The measured NEXT can be approximated with a gamma or a truncated Normal
distribution on log scale. The truncated Normal distribution has a better physical meaning since the number of
NEXT pairs is limited. In practice, we might be concerned about NEXT from more than one disturber. We need to
calculate a power sum for multiple disturbers. We have 50, 3.16 × 1015 and 4.1 × 1011 different power sum NEXT
values for 49 disturbers, 24 disturbers, and 10 disturbers respectively in a 50 pair binder group. The manipulation of
large numbers of power sums for 24 and 10 disturbers is not easy. Hence, NEXT power sum values are
approximated by a statistical distribution function rather than using individual values.
NEXT is dependent on frequency as well as on the relative location of the pairs in the binder group. However,
location is not relevant for a full binder group. Cables differ from one another with respect to NEXT due to the
cable design and manufacturing variations. The NEXT loss at any given frequency, is usually stated as the power
sum of crosstalk from signals in all other pairs of the cable binder group. The NEXT model used for studies such as
the one reported here is stated as expected 1% worst case power sum crosstalk loss as a function of frequency. This
means that on average, 1% of the pairs tested are subjected to power sum crosstalk loss worse (less) than assumed
by the model at the given frequency. Such a model is a smooth curve Vs frequency, in which the loss decreases at
about 15 dB per decade of frequency. The statistical model ensures that the 1% worst case NEXT power sum level
follows the curve over frequency but this noise level may appear on different pairs at different frequencies. The 49NEXT power level for one specific pair is not a smooth curve if plotted over frequency and the deviation from the
statistical model increases as the number of interferers used in the power sum is reduced. The power sum loss for
less than a full binder group depends on the distribution of the pairs on which the crosstalking signal appears.
Measurements for a 25-pair binder group of a 24-AWG PIC cable are given in Figure B.7.
The study of transmission issues related to T1 systems established a first step in dealing with NEXT modeling for
simulations. The study not only tried to model NEXT loss with mean and standard deviation but also initiated the
use of 1% worst case NEXT value for overall system requirements. The reason is that people were expecting better
23
Two-to-four-wire hybrid circuits that act as balanced bridge networks perform the first level of separation between
transmit and receive signals. Ten to twenty dB of isolation can be achieved with active and passive analog
compromise balance impedance networks. Digital echo cancellers can provide 30 to 40 dB of additional isolation.
92
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than 95% satisfactory T1 service at an error ratio of less than 10-6. The use of the 1% worst case for transmission
engineering would allow multiple spans of T1 systems in an end-to-end service connection and also provide room
for some unforeseen impairment.
The same better than 95% satisfactory service objective also applies to other digital subscriber line systems such as
BRI, HDSL, and ADSL. The 1% worst case NEXT model has also been used for BRI and HDSL simulation
studies and test procedures. The piece-wise linear (log-log scale) NEXT models used for BRI and HDSL have loss
values of 57 dB, 61 dB, and 67 dB for 49 disturbers, 10 disturbers, and 1 disturber, respectively, at a frequency of 80
kHz. A simplified 49 disturber NEXT model that has 57 dB of loss at 80 kHz and a linear (log-log scale) slope of 15 dB/decade has been used by Working Group T1E1.4 and can be expressed by
NEXT49 = xn × f 3 / 2
where
xn = 8.818 × 10−14 × (n 49 )0.6 , f = frequency in Hz and n is the number of disturbers. Experimental
results for a 25-pair binder group of a 24-AWG PIC cable, support this model. In Figure B.8, those results are
shown fitted to the model. The difference between the fitted results and the simplified model can be explained as
the difference between a 50-pair binder group of 22-AWG PIC cable (simplified model) and a 25-pair binder group
of 24-AWG PIC cable.
B.4.2 Far end crosstalk, FEXT
Far-end crosstalk (FEXT) is defined as the effect of crosstalk due to adjacent transmitters. In other words, FEXT is
due to crosstalk from adjacent transmitters at the transmitter end that couples to the receiver of another system. See
Figure B.9. FEXT loss is similar but not equal to the combination of NEXT and the subscriber loop channel losses
over the coupling length. FEXT was also considered during T1 transmission engineering efforts but was classified
as a minor factor compared with NEXT. The effect of FEXT for BRI and HDSL is very small and, hence, has been
omitted in test procedures.
The effect of ADSL system self-FEXT can not be simply ignored. At high frequencies and for upstream transmitter
disturbers on short loops, ADSL self-FEXT noise power can exceed that of HDSL NEXT and white background
noise combined. A simplified FEXT model has been used by Working Group T1E1.4 and is expressed by
2
FEXT49 = H (f ) × klf 2
where H ( f ) is the magnitude of the loop insertion gain transfer function, k = 8 × 10-20 × (n 49 )0.6 , n = number of
disturbers, l = the working length in feet subject to FEXT coupling, and f = frequency in Hz. Experimental results
for a 25-pair binder group of a 24-AWG PIC cable, support this model. In Figure B.10, those results are shown
fitted to the model. The difference between the fitted results and the simplified FEXT model can be explained as the
difference between a 50-pair binder group of 22-AWG PIC cable (simplified FEXT model) and a 25-pair binder
group of 24-AWG PIC cable.
The simplified FEXT model assumes the channel transfer function and length of the coupling path match those of
the disturbed system or more simply that the disturber system FEXT sources (transmitters) are co-located with the
transmitter of the disturbed system. In the upstream direction, this underestimates the FEXT where the disturbers
are closer to the central office than the victim signal transmitter.
B.4.3 FSAN method: combining crosstalk from unlike disturber types
B.4.3.1 Base models for NEXT and FEXT
The modeling of interference contributions to an access DSL system due to crosstalk from other DSL systems in the
same cable is a fundamental part of spectral compatibility studies. The widely accepted base models due to work by
Werner and others for near end crosstalk (NEXT) and far end crosstalk (FEXT) which are commonly used (see
B.4.1 and B.4.2) for this modeling are of the form:
NEXT [f , n ] = S [f ] X N f
3
2
n0.6
FEXT [ f , n, l ] = S [ f ] H [ f ]
2
X F f 2 l n 0.6
where f = frequency in Hz, n is the number of disturbing systems, l = length of the cable in feet, X N and X F
are scalar constants, S [f ] is the PSD of the interfering systems and H [ f ] is the loop insertion gain transfer function.
The above base models are stated as expected 1% worst case power sum crosstalk noise. This means that on the
average, 1% of the cables tested have power sum crosstalk noise worse (greater) than the base models at any given
This is a draft document and thus, is still dynamic in nature.
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frequency, number of disturbers and loop length. There is an implicit assumption in these models that all the pairs
involved are in the same binder group of the same cable and have a common length and also that all the interferers
are of the same type.
There is a counter-intuitive aspect of these models relating to the n 0.6 term. Intuitively it would be expected for the
interference power to be proportional to the number of disturbers (since the disturbers are independent) but instead
there is the n 0.6 factor. This is due to the fact that the quantity being dealt with is not an average value or an
expectation of any sort, but a 1% worst case.
If the proximity of pairs in a cable segment is maintained along its length, certain pairs (usually the proximate ones)
contribute much more to the interference in a given pair than others do. When there are few interferers (n small) if a
single member is one of the proximate pairs the contribution to interference is disproportionately increased. For this
reason the model has to be biased for small numbers of interferers and this is the reason for the exponent of n being
less than unity.
A difficulty arises when modeling complex access network scenarios though, where there may be many types of
interferer. Suppose for example that the NEXT from n1 systems of spectrum S1[ f ] and n2 systems of spectrum
S2 [ f ] is considered. The obvious way of extending the model to cope with this is to add the crosstalk power
contributions according to the base model for each:
NEXT [f ] = S1[f ] X N f
3
2
n10.6 + S 2 [f ] X N f
3
2
n20.6
The difficulty here is that each term in this expression is pessimistic enough for the 1% worst case, but their joint
probability is much lower, so the combined model is excessively pessimistic. This can be seen by taking this
expression and allowing S2 [ f ] = S1[ f ] (the interferers have become of the same type). In this case the expression
can be simplified to:
NEXT [f ] = S1[f ] X N f
3
2
(n
0.6
1
+ n20.6
)
whereas the base model would in this case predict the lesser interference of:
NEXT [f ] = S1[f ] X N f
3
2
(n1 + n2 )0.6
This annex describes the recommended method for calculation of NEXT and FEXT contributions from groups of
unlike disturbers. The method avoids making an over pessimistic calculation of total crosstalk contribution which
arises when assuming that all sub-groups of n interfering systems are using the worst n pairs in a multi-pair cable.
It does so without treating any sub-group differently so that there is only one way of making the computation. The
computation is such that in various limiting or trivial cases it converges asymptotically to the base model for the
reduced state. Also it never predicts a lower crosstalk level when more disturbers are added.
The method is equally applicable to the calculation of NEXT and FEXT models.
B.4.3.2 Combining crosstalk from mixed disturber types
Instead of directly adding the crosstalk power terms, each term is first arbitrarily raised to the power 1/0.6 before
carrying out the summation. Then, after the summation, the resultant expression is raised to the power 0.6. There is
no simple physical justification for this process but it has been shown both analytically below and elsewhere by
means of Monte Carlo simulations that the method has many sound and realistic properties.
B.4.3.3 Application to two NEXT terms
Take the example from B.4.3.1. The combined NEXT power would take the form:
(

3
NEXT [f ] =  S1[f ] X N f 2 n10.6

)
1
0.6
+
( S [f ] X
2
N
f
3
2
n 2 0.6
)
1

0.6 
0.6


The first sound property is that if either inner term vanishes the model returns to the base model. Suppose for
example that S 2 = 0 or n 2 = 0 . In this case the second term would vanish. This would leave the two arbitrarily
introduced exponents acting on a single expression, so that they cancel out, returning the expression to the base
model.
The second sound property arises when S2 = S1 . In this case the common factors S1[ f ] X N f 3 / 2 can be taken out
of the two inner terms, and further brought outside the enclosing brackets, leaving:
94
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T1E1.4/2000-002R6
ANSI T1.417
NEXT [f ] = S1[f ] X N f
( )
2
 0.6
 n1


2
(n1 + n2 )0.6
3
1
0.6
(
+ n 20.6
)
1

0.6 
0.6


This in turn quickly collapses to:
NEXT [f ] = S1[f ] X N f
3
which is identical to the base model for the case of n1 + n 2 identical disturbers.
The same process can be applied to collections of more than two interference contributions.
B.4.3.4 Application to FEXT terms
The same process can also be applied to collections of FEXT interferers.
Take the case of three sources of FEXT at a given receiver. In this case there are n1 systems of spectrum S1[ f ] at
range l 1 , a further n 2 systems of spectrum S2 [ f ] at range l 2 and yet another n 3 systems of spectrum S3 [ f ] at
range l 3 .
The expected crosstalk is built in exactly the same way as before, taking the base model for each source, raising it to
power 1/0.6, adding these expressions, and raising the sum to power 0.6:
1
1



  S [ f ] H [f ] 2 X f 2 l n 0.6  0.6 +  S [ f ] H [f ] 2 X f 2 l n 0.6  0.6
1
1
2
2
F
1
1
F
2
2





FEXT [ f ] = 

1
2

0.6  0.6 
2

[
]
X F f l 3 n3 
+ S3 [ f ] H 3 f






In this case it is assumed that
0.6
H 1 [ f ] is the loop insertion gain transfer function of the length l 1 , etc.
Even in this more complex case the same sound properties appear.
The first sound property is that if any of the inner terms vanishes the model returns to the simpler case until when
there is only one inner term left it returns to the base model. For FEXT though there are many more ways in which
a term can disappear. Instead of just S 2 = 0 or n 2 = 0 there are also the possibilities l 2 = 0 and l 2 → ∞ . The
2
latter arises because the product l 2 H 2 [ f ] → 0 as l 2 → ∞ . In any of these cases the second term would vanish,
and the equation is exactly as it would appear if the second crosstalk subgroup had not been considered in the first
place. If in addition the third term disappears, for example because n 3 = 0 , the resulting equation is easily reduced
to the base model for just the first subgroup of interferers.
The second sound property arises when for example
S 2 = S1 and l 2 = l 1 . This means that the first two terms
actually relate to identical system types causing FEXT at the same location. As l 2 = l 1 it can be assumed that
2
2
H 2 [ f ] = H 1[ f ] . In this case the common factors  S1[ f ] H1[f ] 2 X F f 2 l 1 
1
0.6


can be taken out of the
first two inner terms, leaving the expression:
(

FEXT [ f ] =  S1[ f ] H12 [f ] X F f 2 l 1


)
)
 (n
1 
0.6 

0.6
1
1
0.6
(
+ n 20.6
)
1

0.6 
(
2
0.6
2
 + S3 [ f ] H 3 [f ] X F f l 3 n 3

)
1

0.6 
0.6


The exponents around n1 and n 2 now collapse to yield the sum n1 + n 2 which can then be taken back inside the
common factor to yield:

FEXT [ f ] =   S1[ f ] H1[ f ]


2
XF f
2
1
0.6  0.6
l 1 (n1 + n 2 )


+  S 3 [ f ] H 3 [ f ]

2
XF f
2
l 3 n3
1

0.6  0.6 


0.6


This is exactly the form that would be obtained if the new method were applied to the simplified modeling situation
(of an increased number of identical disturbers at the same location) in the first place.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R6
ANSI T1.417
In addition if the terms subscripted with 3 were to vanish, for example because n 3 = 0 , then the expression would
further simplify to the base model for the remaining interferers.
B.4.3.5 Crosstalk is non-decreasing
It will be apparent that the exponentiation operations, which are applied in this process, are applied to quantities of
dimension power. This means of course that they are applied to real positive functions. After exponentiation the
functions are still real and positive. As adding more disturbers is modeled by adding together these real positive
functions and then applying a monotonic mapping to the sum (the subsequent exponentiation with exponent 0.6) it
follows that adding more disturbers always increases the crosstalk.
B.4.3.6 All disturbers are treated equally
It should be apparent from the absolute symmetry of the method that all disturbers are treated equally. It does not
matter what order the disturbers are taken in the resulting expression is the same.
B.4.3.7 Adding NEXT and FEXT
The method should be separately applied to the NEXT terms and the FEXT terms to arrive at separate NEXT and
FEXT disturbance power spectra. These power spectra should then be added.
The method should not itself be used for adding NEXT to FEXT. This is because it is perfectly feasible for the same
proximate disturbing pair to contribute both NEXT and FEXT powers from different disturbing transceivers,
whereas it cannot contribute two lots of NEXT or two lots of FEXT from different disturbing transceivers.
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Table B.1 – American wire gauge (AWG) and metric wire
Metric & AWG Wire Gauges: R & C (0Hz/DC, 20°C or 70°F)
nF/k
m
40
40
40
nF/mi
Ω/km
Ω/mi
Ω/kft
64.4
64.4
64.4
677
494
409
1090
792
658
206
150
125
50
51.6
40.0
80.5
83.0
64.3
280
273
213
451
439
343
84.4
83.1
64.9
20.10 mil
50.9
55
50
51.6
82.0
88.5
80.5
83.0
181
179
179
172
291
288
288
276
55.1
55.6
54.5
52.3
25.35 mil
40
45
51.6
64.4
72.4
83.0
123
113
108
198
182
174
37.5
34.5
32.9
40
64.4
90
145
27.5
40
64.4
69
66.6
111
107
21.0
20.3
55.5
55
53.8
141
89.3
88.5
86.6
227
16.9
16.8
16.4
43
41.2
41.9
66.3
67.4
12.6
12.8
mm
AWG
mils
(0.253)
0.30 PE
0.32 PE
30
10.03 mil
28
12.64 mil
0.40 PE
(0.405)
(0.455)
26 PIC
25 MAT
15.94 mil
0.50 DW10
0.50 DUG
0.50 PE
(0.511)
24 PIC
0.60 PE
0.63 PE
(0.644)
22 PIC
0.70 PE
0.80 PE
(0.812)
20
31.96 mil
0.90 DW12
0.90 PE
(0.912)
19 PIC
(0.965)
18 ½ 1)
1.0 DW8
(1.024)
18
DW1
DW3
DW5
DW6
NOTES:
40.3 mil
51.1
40
51.6
82.3
64.4
83.0
72118
116190
22.7
36.5
NOTES
Loss
Metro. Area Trans., PIC
for comparison, not
telephony
35.89
PVC, copper-clad steel,
parallel
PVC, copper &cadmium
for comparison, not
telephony
28.0 45.1
63.5
102
19.4
PVC, copper & cadmium
24.4 39.3
266
428
81.0
PVC, copper-clad steel
29.3 47.2
258
415
78.6
PVC, copper-clad steel
27.9 44.9
200
322
61.0
PVC, copper-clad steel
( ) = AWG conductor diameter → (mm) = not a normal metric size
PE = metric Polyethylene insulated cable
PIC = AWG Polyethylene insulated cable, sometimes called "plastic insulated cable"
as contrasted to older pulp or paper insulated cable.
PVC = Polyvinyl chloride insulated cable
DW = European drop wire, overhead/aerial
DUG = European underground drop cable
1)
F Drop Wire, AT-8668, aerial, parallel (flat, not twisted) 18 ½ AWG copper-clad steel conductors,
solid black PVC insulation, oval cross section, conductor diameter = 0.038 inch, 43 Ω/kft = 227 Ω/mi,
C = 0.116 µf/mi dry, C = 0.190 µf/mi wet (US Drop wire limits: 700 feet or 25 Ω)
This is a draft document and thus, is still dynamic in nature.
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Table B. 2 - Cable model parameters for 26-AWG twisted pair cable
Resistance
r0c
r0s
(value)
Inductance
286.176 Ω/km
l0
∞ Ω/km
l∞
(value)
Capacitance
(value)
Conductance
675.369 µH/km
c∞
49 nF/km
g0
488.952 µH/km
c0
0.0 nF/km
ge
(value)
43 nMho/km
.70
ac
0.14769620
b
0.929
ce
0.0
as
0.0
fm
806.339 kHz
Table B. 3 - Primary constants for 26-AWG twisted pair cable
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
10.5e6
30.e6
98
Resistance
(Ω/km)
286.215
286.333
286.804
290.036
300.775
626.851
1.9606e3
2.0090e3
3.3955e3
Inductance
(H/km)
673.728e-6
672.268e-6
669.552e-6
662.286e-6
651.941e-6
572.869e-6
505.334e-6
504.669e-6
495.205e-6
Capacitance
(F/km)
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
Conductance
(Mho/km)
16.701e-6
27.131e-6
44.075e-6
83.704e-6
135.978e-6
681.504e-6
3.4156e-3
3.5343e-3
7.3698e-3
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table B. 4 - Cable parameters for 26-AWG PIC air core
MHz
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
2.959
3.225
3.509
3.78
4.119
4.482
4.827
5.26
5.724
6.165
6.718
7.31
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
20.95
R
G
L
C
α
(Ω/km) (µMho/km) (mH/km) (nF/km) (dB/km)
354.02
95
0.579
49.61
14.26
358.78
103
0.579
49.87
14.50
364.39
112
0.577
50.09
14.78
372.51
123
0.576
50.29
15.16
380.64
133
0.576
50.46
15.52
392.35
145
0.575
50.60
16.04
405.63
158
0.574
50.71
16.63
420.44
170
0.573
50.82
17.27
440.01
186
0.571
50.89
18.11
462.03
203
0.569
50.92
19.06
485.14
218
0.568
50.96
20.05
512.91
238
0.565
50.93
21.25
542.09
258
0.562
50.88
22.50
568.12
278
0.559
50.83
23.63
596.53
303
0.556
50.74
24.87
622.32
329
0.552
50.64
26.01
643.03
354
0.550
50.60
26.94
665.05
385
0.547
50.55
27.93
687.17
419
0.544
50.53
28.94
709.85
451
0.542
50.56
29.97
741.08
492
0.539
50.56
31.37
776.22
535
0.536
50.56
32.95
809.18
576
0.534
50.57
34.43
844.21
628
0.531
50.54
36.03
873.33
683
0.528
50.53
37.38
898.41
736
0.526
50.56
38.56
936.27
803
0.523
50.59
40.30
978.03
874
0.521
50.60
42.23
1014.30
942
0.519
50.63
43.91
1051.68
1026
0.516
50.65
45.67
1094.11
1118
0.514
50.68
47.66
1135.57
1205
0.512
50.73
49.59
1181.04
1314
0.509
50.76
51.74
1229.07
1430
0.507
50.80
54.01
1274.64
1542
0.505
50.85
56.15
1326.70
1682
0.503
50.89
58.63
1383.81
1831
0.501
50.92
61.34
1433.71
1975
0.499
50.98
63.71
1496.64
2153
0.497
51.01
66.71
1555.32
2345
0.495
51.05
69.53
1610.79
2529
0.494
51.11
72.20
1679.19
2758
0.492
51.15
75.49
1747.18
3003
0.490
51.19
78.77
1809.56
3238
0.489
51.25
81.79
1883.19
3532
0.487
51.29
85.37
1959.05
3846
0.485
51.34
89.07
2030.18
4148
0.484
51.40
92.53
2113.47
4524
0.483
51.44
96.60
2204.55
4926
0.481
51.48
101.05
2285.22
5313
0.480
51.55
105.00
2381.34
5793
0.479
51.58
109.73
2484.60
6309
0.477
51.62
114.80
2577.25
6803
0.476
51.69
119.36
β
Z0 (Ω)
(rad/km)
10.23
11.05
12.05
13.13
14.16
15.44
16.80
18.10
19.71
21.41
23.04
25.04
27.16
29.18
31.67
34.32
36.86
40.03
43.44
46.71
50.78
55.11
59.24
64.34
69.81
75.08
81.63
88.62
95.29
103.60
112.51
121.02
131.59
142.92
153.76
167.24
181.65
195.47
212.61
230.98
248.56
270.44
293.84
316.25
344.13
373.99
402.58
438.13
476.19
512.65
557.99
606.52
653.04
108.05
107.71
107.37
107.07
106.84
106.59
106.35
106.16
105.94
105.72
105.53
105.32
105.09
104.90
104.67
104.43
104.22
103.98
103.73
103.51
103.25
102.98
102.76
102.49
102.22
101.99
101.71
101.44
101.21
100.94
100.67
100.44
100.18
99.92
99.70
99.44
99.19
98.98
98.74
98.50
98.30
98.07
97.84
97.66
97.44
97.23
97.06
96.86
96.66
96.50
96.32
96.14
95.99
This is a draft document and thus, is still dynamic in nature.
99
T1E1.4/2000-002R6
ANSI T1.417
Table B. 4 (concluded)
MHz
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
R
G
L
C
(Ω/km) (µMho/km) (mH/km) (nF/km)
2691.58
7419
0.475
51.72
2811.07
8079
0.474
51.76
2915.22
8711
0.473
51.82
3058.27
9499
0.472
51.85
3189.29
10343
0.470
51.89
3320.49
11153
0.470
51.94
3469.12
12160
0.469
51.98
3606.84
13077
0.468
52.03
α
(dB/km)
125.00
130.90
136.07
143.12
149.65
156.16
163.58
170.45
β
(rad/km)
710.86
772.74
832.08
905.83
984.79
1060.47
1154.59
1240.11
Z0 (Ω)
95.82
95.65
95.52
95.36
95.21
95.09
94.95
94.83
Table B. 5 - Cable parameters for 26-AWG filled PIC
100
MHz
R
(Ω/km)
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
2.959
3.225
3.509
3.78
4.119
4.482
4.827
5.26
5.724
6.165
6.718
7.31
397.8
398.7
399.5
400.3
401.6
403.9
407.3
413.9
423.1
437.7
454.6
478.8
506.4
533.3
565.9
595.1
616.4
635.4
649.9
665
688.1
721.9
758
796
821.3
840
871.4
913.4
948.7
979.9
1018
1057
1100
1153
1196
1243
1300
1347
1403
1467
G
L
C
(µMho/km) (mH/km) (nF/km)
48.3
52
56
60.2
64.9
69.7
75.1
80.8
86.9
93.6
101
108
117
125
135
145
156
168
181
195
209
226
243
261
281
302
326
351
377
406
436
471
506
545
586
630
679
730
786
846
0.685
0.682
0.68
0.677
0.676
0.674
0.672
0.671
0.67
0.668
0.667
0.665
0.663
0.661
0.658
0.654
0.652
0.649
0.646
0.644
0.643
0.641
0.639
0.637
0.634
0.633
0.632
0.63
0.628
0.627
0.625
0.624
0.623
0.621
0.62
0.619
0.618
0.617
0.616
0.614
46.44
46.77
47.09
47.38
47.64
47.9
48.15
48.37
48.59
48.78
48.95
49.07
49.14
49.18
49.15
49.08
49.05
48.98
48.95
48.97
48.99
49
48.99
48.93
48.86
48.86
48.85
48.82
48.81
48.77
48.75
48.75
48.72
48.69
48.68
48.66
48.63
48.62
48.6
48.57
α
(dB/km)
β
(rad/km)
14.267
14.376
14.488
14.592
14.701
14.854
15.042
15.336
15.734
16.328
17.005
17.959
19.045
20.1
21.376
22.528
23.378
24.153
24.758
25.376
26.309
27.65
29.074
30.583
31.607
32.377
33.646
35.321
36.73
38.004
39.56
41.111
42.837
44.956
46.712
48.622
50.907
52.81
55.087
57.697
10.767
11.618
12.677
13.813
14.902
16.259
17.715
19.112
20.851
22.712
24.487
26.678
29
31.197
33.902
36.772
39.512
42.923
46.595
50.143
54.578
59.311
63.801
69.341
75.261
80.973
88.12
95.73
102.987
112.019
121.726
131.005
142.545
154.887
166.675
181.378
197.094
212.146
230.88
250.927
This is a draft document and thus, is still dynamic in nature.
Z0
(Ω)
121.409
120.79
120.134
119.54
119.076
118.579
118.127
117.77
117.386
117.032
116.751
116.446
116.164
115.937
115.689
115.457
115.27
115.064
114.871
114.713
114.539
114.374
114.239
114.089
113.947
113.83
113.699
113.575
113.472
113.358
113.248
113.158
113.056
112.959
112.879
112.789
112.702
112.631
112.551
112.473
T1E1.4/2000-002R6
ANSI T1.417
Table B. 5 (concluded)
MHz
R
(Ω/km)
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
20.95
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
1519
1581
1650
1712
1785
1872
1943
2024
2129
2206
2307
2431
2513
2636
2759
2886
2996
3149
3301
3470
3671
G
L
C
(µMho/km) (mH/km) (nF/km)
909
980
1054
1134
1222
1312
1415
1522
1637
1763
1894
2042
2196
2363
2545
2734
2947
3170
3411
3673
3946
0.614
0.613
0.612
0.611
0.61
0.609
0.608
0.608
0.607
0.606
0.605
0.605
0.604
0.603
0.603
0.603
0.602
0.601
0.601
0.6
0.6
48.57
48.54
48.52
48.52
48.5
48.48
48.48
48.46
48.43
48.44
48.42
48.4
48.4
48.38
48.36
48.37
48.35
48.33
48.35
48.33
48.34
α
(dB/km)
β
(rad/km)
59.805
62.34
65.163
67.691
70.701
74.235
77.176
80.519
84.794
87.994
92.156
97.247
100.693
105.808
110.912
116.164
120.82
127.189
133.523
140.573
148.837
270.094
293.981
319.541
344.03
374.474
407.112
438.268
477.129
518.684
558.535
608.048
661.129
711.891
774.997
842.743
907.614
988.173
1074.618
1157.406
1260.329
1353.774
Z0
(Ω)
112.41
112.338
112.269
112.212
112.148
112.086
112.035
111.978
111.923
111.877
111.826
111.776
111.736
111.69
111.646
111.609
111.568
111.529
111.496
111.46
111.429
Table B. 6 - Cable model parameters for 24-AWG twisted pair cable
Resistance
r0c
r0s
ac
as
(value)
Inductance
174.559 Ω/km
l0
∞ Ω/km
l∞
0.05307
b
0.0
fm
(value)
Capacitance
617.295 µH/km
c∞
478.971 µH/km
c0
1.15298
ce
553.76 kHz
(value)
Conductance
50 nF/km
g0
0.0 nF/km
ge
0.0
(value)
234.875 pMho/km
1.38
Table B. 7 - Primary constants for 24-AWG twisted pair cable
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
(Ω/km)
174.621
174.808
175.548
180.486
195.447
482.061
1.51788e3
2.62895e3
Inductance
(H/km)
616.690e-6
615.957e-6
614.353e-6
609.159e-6
600.416e-6
525.440e-6
483.722e-6
480.344e-6
Capacitance
(F/km)
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
Conductance
(Mho/km)
29.882e-9
77.774e-9
202.422e-9
716.828e-9
1.86568e-6
44.7544e-6
1.07358e-3
4.88950e-3
This is a draft document and thus, is still dynamic in nature.
101
T1E1.4/2000-002R6
ANSI T1.417
Table B. 8 - Cable parameters for 24-AWG PIC air core
MHz
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
2.959
3.225
3.509
3.78
4.119
4.482
4.827
5.26
5.724
6.165
6.718
7.31
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
102
R
G
L
C
α
β
(Ω/km) (µMho/km) (mH/km) (nF/km) (dB/km) (rad/km)
269.87
280.59
291.94
302.05
310.67
321.11
333.53
346.45
362.84
378.68
391.83
405.72
420.66
436.52
455.25
472.06
487.37
507.85
527.94
545.55
568.5
590.64
611.79
636.98
662.83
686.62
715.83
744.89
772.18
804.85
838.52
869.26
905.99
944.23
978.86
1021.06
1064.19
1104.2
1152.18
1201.2
1246.16
1300.4
1356.48
1407.6
1469.55
1533.1
1591.69
1662.07
1735.64
1802.69
1886.11
1973.16
98
105
115
125
134
146
159
172
188
204
220
240
262
282
308
335
362
394
429
463
504
549
592
645
702
756
823
896
965
1051
1143
1231
1341
1458
1570
1709
1859
2002
2179
2369
2551
2778
3020
3252
3540
3849
4144
4512
4905
5282
5750
6251
0.581
0.578
0.574
0.57
0.567
0.563
0.559
0.556
0.552
0.548
0.545
0.54
0.537
0.534
0.53
0.526
0.524
0.521
0.518
0.515
0.513
0.51
0.508
0.506
0.504
0.502
0.5
0.498
0.497
0.495
0.494
0.493
0.492
0.49
0.489
0.488
0.487
0.487
0.486
0.485
0.484
0.484
0.483
0.482
0.482
0.481
0.481
0.48
0.48
0.48
0.479
0.479
51.3
51.22
51.13
51.07
51.09
51.12
51.19
51.29
51.36
51.41
51.45
51.48
51.53
51.62
51.66
51.69
51.74
51.78
51.8
51.84
51.86
51.87
51.9
51.9
51.9
51.92
51.9
51.88
51.9
51.87
51.84
51.84
51.81
51.77
51.77
51.72
51.68
51.67
51.63
51.58
51.57
51.52
51.47
51.46
51.41
51.37
51.35
51.3
51.26
51.25
51.2
51.15
11.05
11.51
12.01
12.46
12.86
13.34
13.92
14.51
15.27
16.01
16.63
17.29
18.01
18.76
19.64
20.45
21.18
22.15
23.11
23.95
25.03
26.09
27.09
28.28
29.5
30.62
32
33.37
34.65
36.19
37.77
39.21
40.93
42.73
44.35
46.33
48.35
50.22
52.47
54.77
56.88
59.43
62.06
64.46
67.37
70.36
73.12
76.44
79.91
83.08
87.03
91.14
10.42
11.19
12.14
13.16
14.14
15.36
16.67
17.92
19.47
21.12
22.68
24.63
26.71
28.72
31.2
33.85
36.38
39.54
42.91
46.14
50.15
54.44
58.54
63.65
69.11
74.34
80.84
87.78
94.44
102.72
111.57
120.05
130.59
141.86
152.67
166.1
180.48
194.25
211.37
229.7
247.25
269.08
292.44
314.84
342.67
372.47
401.04
436.54
474.55
510.99
556.28
604.76
This is a draft document and thus, is still dynamic in nature.
Z0
(Ω)
106.39
106.24
105.99
105.67
105.35
104.96
104.53
104.16
103.71
103.27
102.89
102.46
102.03
101.69
101.29
100.92
100.62
100.28
99.96
99.7
99.42
99.16
98.95
98.72
98.51
98.35
98.17
98
97.87
97.73
97.61
97.51
97.41
97.31
97.24
97.16
97.1
97.05
96.99
96.95
96.91
96.88
96.85
96.82
96.8
96.78
96.77
96.76
96.75
96.75
96.74
96.74
T1E1.4/2000-002R6
ANSI T1.417
Table B. 8(concluded)
MHz
R
G
L
C
α
β
(Ω/km) (µMho/km) (mH/km) (nF/km) (dB/km) (rad/km)
20.95
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
2049.06
2145.43
2251.42
2338.22
2453.81
2573.37
2678.96
2816.8
2935
6732
7329
7968
8581
9342
10157
10938
11909
12792
0.479
0.478
0.478
0.478
0.477
0.477
0.477
0.477
0.477
51.14
51.09
51.05
51.04
51
50.95
50.94
50.9
50.9
94.75
99.32
104.34
108.48
113.98
119.68
124.73
131.31
136.97
Z0
(Ω)
96.74
96.75
96.75
96.76
96.76
96.77
96.78
96.79
96.79
651.24
709.04
770.89
830.23
903.98
982.89
1058.57
1152.64
1238.16
Table B. 9 - Cable parameters for 22-AWG PIC air core
MHz
R
G
L
C
α
β
(Ω/km) (µMho/km) (mH/km) (nF/km) (dB/km) (rad/km)
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
2.959
3.225
3.509
3.78
4.119
4.482
4.827
5.26
5.724
197.3
202.3
207.61
214
220.28
227.97
236.78
246.08
258.3
271.25
285.09
301.4
317.76
333.2
349.26
363.23
374.44
386.35
399.1
412.51
431.41
452.73
471.26
492.08
508.82
525.41
552.67
580.94
602.38
621.3
650.94
679.27
704.13
733.72
765.09
795.07
835.86
96
104
113
123
133
145
159
171
187
204
220
240
262
282
308
335
361
394
429
462
504
549
592
645
701
756
824
896
965
1052
1144
1233
1342
1460
1573
1713
1862
0.537
0.537
0.536
0.534
0.533
0.531
0.528
0.527
0.524
0.521
0.519
0.516
0.513
0.51
0.507
0.504
0.502
0.499
0.497
0.495
0.493
0.491
0.49
0.488
0.486
0.485
0.483
0.482
0.481
0.479
0.478
0.477
0.476
0.475
0.475
0.474
0.473
50.27
50.33
50.41
50.51
50.64
50.76
50.89
51.03
51.15
51.27
51.38
51.45
51.51
51.56
51.58
51.59
51.64
51.67
51.72
51.79
51.83
51.86
51.89
51.89
51.89
51.93
51.94
51.92
51.92
51.9
51.9
51.9
51.87
51.85
51.85
51.82
51.78
8.33
8.55
8.79
9.09
9.38
9.74
10.16
10.59
11.16
11.76
12.41
13.17
13.93
14.66
15.42
16.09
16.64
17.23
17.85
18.5
19.4
20.42
21.3
22.3
23.11
23.91
25.21
26.55
27.57
28.5
29.91
31.25
32.45
33.87
35.36
36.8
38.74
9.92
10.69
11.65
12.67
13.64
14.86
16.15
17.39
18.93
20.56
22.13
24.06
26.11
28.07
30.5
33.09
35.58
38.68
42.01
45.21
49.19
53.43
57.47
62.49
67.88
73.06
79.49
86.35
92.89
101.06
109.84
118.21
128.6
139.78
150.44
163.73
177.94
Z0
(Ω)
103.38
103.28
103.08
102.83
102.57
102.25
101.89
101.58
101.21
100.83
100.52
100.15
99.79
99.5
99.16
98.84
98.58
98.29
98.02
97.8
97.56
97.34
97.15
96.96
96.78
96.63
96.48
96.33
96.22
96.1
95.99
95.9
95.81
95.73
95.67
95.6
95.54
This is a draft document and thus, is still dynamic in nature.
103
T1E1.4/2000-002R6
ANSI T1.417
Table B. 9(concluded)
MHz
R
G
L
C
α
β
(Ω/km) (µMho/km) (mH/km) (nF/km) (dB/km) (rad/km)
6.165
6.718
7.31
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
20.95
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
861.63
900.91
937.53
974.62
1016.13
1057.87
1097.92
1145.72
1191.83
1236.98
1290.75
1349.41
1403.08
1464.42
1534.46
1591.96
1668.08
1737.61
1808.44
1889.94
1994.92
2070.38
2175.93
2278.32
2006
2184
2375
2558
2786
3029
3262
3552
3862
4159
4529
4925
5304
5775
6280
6763
7364
8008
8624
9391
10212
10999
11977
12866
0.472
0.471
0.471
0.47
0.47
0.469
0.469
0.468
0.468
0.468
0.467
0.467
0.467
0.466
0.466
0.466
0.465
0.465
0.465
0.465
0.465
0.465
0.464
0.464
51.78
51.75
51.71
51.71
51.67
51.63
51.63
51.59
51.55
51.54
51.51
51.47
51.46
51.42
51.38
51.38
51.34
51.3
51.3
51.26
51.23
51.23
51.19
51.19
39.99
41.87
43.63
45.41
47.41
49.42
51.35
53.66
55.9
58.08
60.69
63.54
66.14
69.14
72.54
75.36
79.08
82.51
85.99
90.02
95.14
98.9
104.11
109.13
191.54
208.49
226.63
243.97
265.58
288.69
310.85
338.4
367.9
396.17
431.33
468.97
505.05
549.89
597.91
643.91
701.15
762.44
821.18
894.26
972.48
1047.44
1140.68
1225.38
Z0
(Ω)
95.5
95.45
95.41
95.37
95.34
95.31
95.29
95.27
95.26
95.24
95.23
95.23
95.22
95.22
95.21
95.21
95.21
95.22
95.22
95.22
95.23
95.23
95.24
95.25
Table B. 10 - Cable model parameters for FP (1.14 mm flat cable)
Resistance
r0c
r0s
(value)
Inductance
41.16 Ω/km
l0
∞ Ω/km
(value)
Capacitance
1000 µH/km
c∞
22.68 nF/km
g0
53 nMho/km
911 µH/km
c0
31.78 nF/km
ge
.88
(value)
Conductance
(value)
104
l∞
ac
.001218
b
as
0
fm
1.195
ce
.1109
174.2 kHz
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table B. 11 - Primary constants for FP (1.14 mm flat cable)
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
(Ω/km)
41.269
41.59
42.805
49.316
62.285
186.924
590.762
1.0232e3
Inductance
(H/km)
998.73982e-6
997.16583e-6
993.76481e-6
983.62766e-6
969.66713e-6
920.40732e-6
911.20963e-6
910.69563e-6
Capacitance
(F/km)
35.041871e-9
34.127572e-9
33.280903e-9
32.257008e-9
31.548702e-9
29.550852e-9
28.003118e-9
27.392833e-9
Conductance
(Mho/km)
95.360709e-6
175.49949e-6
322.98493e-6
723.38496e-6
1.3312998e-3
10.098942e-3
76.608308e-3
201.43854e-3
Table B. 12 - Cable model parameters for category 5 twisted pair
Resistance
r0c
r0s
(value)
Inductance
176.6 Ω/km
l0
∞ Ω/km
l∞
(value)
Capacitance
1090.8 µH/km
c∞
48.55 nF/km
g0
1.477 nMho/km
504.5 µH/km
c0
0.0 nF/km
ge
.91
(value)
Conductance
(value)
ac
.050008
b
0.705
ce
0.0
as
0.0
fm
32.570 kHz
Table B. 13 - Primary constants for category 5 twisted pair
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
(Ω/km)
176.657
176.827
177.501
182.020
195.899
475.172
1.49548e3
2.59014e3
Inductance
(H/km)
967.308142e-6
913.078780e-6
847.551900e-6
753.691218e-6
687.417012e-6
552.634084e-6
514.663928e-6
509.228994e-6
Capacitance
(F/km)
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
Conductance
(Mho/km)
3.430086e-6
6.445287e-6
12.110988e-6
27.880784e-6
52.389261e-6
425.835904e-6
3.461324e-3
9.406382e-3
This is a draft document and thus, is still dynamic in nature.
105
T1E1.4/2000-002R6
ANSI T1.417
Table B. 14 - Cable parameters, two-pair twisted drop
MHz
0.772
0.819094
0.869062
0.922077
0.978327
1.038008
1.101329
1.168514
1.239797
1.315428
1.395673
1.480813
1.571148
1.666992
1.768684
1.876579
1.991056
2.112517
2.241387
2.378118
2.523191
2.677113
2.840425
3.0137
3.197545
3.392605
3.599564
3.819149
4.052129
4.299321
4.561593
4.839864
5.13511
5.448368
5.780735
6.133378
6.507533
6.904512
7.325709
7.772599
8.246752
8.749829
9.283595
9.849923
10.4508
11.08833
11.76475
12.48244
13.2439
14.05182
14.90903
106
R
(Ω/kft)
L
(mH/kft)
C
(nF/kft)
α
(dB/kft)
112.6412
116.4074
120.112
124.0752
128.3016
132.5838
137.0843
142.5063
147.1853
151.9907
157.6941
163.022
168.7727
174.1886
180.6494
187.1776
194.3419
201.1778
209.5537
216.4363
223.9826
232.2693
240.6053
250.1631
259.1278
268.8572
279.5037
290.5045
300.6599
311.7282
323.8352
336.0791
349.7153
363.3367
377.6032
393.3168
407.97
426.5291
443.104
461.1752
479.3825
499.3066
518.4216
539.9883
562.2894
586.0891
610.2553
635.0644
662.0816
691.5593
720.1421
0.143576
0.141203
0.140489
0.140184
0.139989
0.139838
0.139706
0.139581
0.139459
0.139336
0.13921
0.137695
0.138949
0.138813
0.138676
0.138536
0.138396
0.138254
0.138115
0.137974
0.137835
0.137699
0.137566
0.137436
0.137309
0.137186
0.137066
0.13695
0.136837
0.136727
0.13662
0.136515
0.136413
0.136312
0.136212
0.136114
0.135994
0.135918
0.135701
0.135721
0.135621
0.13552
0.135417
0.135312
0.135205
0.135096
0.134984
0.13487
0.134752
0.134631
0.134508
14.15747
13.94644
13.89817
13.88958
13.89123
13.89671
13.90334
13.91021
13.91676
13.92271
13.92782
13.79328
13.93559
13.93821
13.94017
13.94142
13.94219
13.94235
13.94233
13.94175
13.94097
13.94009
13.93908
13.9381
13.93699
13.93593
13.93491
13.9339
13.93283
13.93179
13.93077
13.92969
13.92858
13.92734
13.92597
13.92445
13.92044
13.92074
13.90626
13.91594
13.91307
13.90983
13.90623
13.9022
13.89775
13.89286
13.88752
13.88174
13.87545
13.86865
13.86144
4.85443771
5.02088326
5.18481854
5.36006449
5.54683894
5.73618274
5.93512422
6.17415048
6.38116738
6.59381465
6.84559689
7.0812544
7.3354461
7.57525379
7.86066953
8.14921318
8.46564642
8.7679671
9.13761936
9.44235495
9.77621976
10.142589
10.5113113
10.9336262
11.3302283
11.7604684
12.2310533
12.7173706
13.1668983
13.6566036
14.1920396
14.7337063
15.3366472
15.939193
16.5702788
17.2651257
17.9136803
18.7340146
19.4674802
20.266952
21.072676
21.9541433
22.8003059
23.7545736
24.7414418
25.7945582
26.8641089
27.962258
29.1579437
30.462329
31.7276262
Z0
(Ω)
100.7043
100.6214
100.5408
100.4626
100.3867
100.313
100.2415
100.172
100.1046
100.0391
99.97558
99.91388
99.85398
99.79583
99.73937
99.68456
99.63135
99.5797
99.52955
99.48086
99.43359
99.3877
99.34315
99.2999
99.25792
99.21715
99.17758
99.13916
99.10186
99.06565
99.0305
98.99637
98.96324
98.93107
98.89984
98.86953
98.84009
98.81152
98.78378
98.75685
98.7307
98.70532
98.68068
98.65676
98.63353
98.61098
98.58909
98.56784
98.54721
98.52718
98.50774
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table B. 14 (concluded)
MHz
15.81852
16.7835
17.80735
18.89365
20.04622
21.2691
22.56658
23.94321
25.40382
26.95353
28.59778
30.34234
32.19331
34.1572
36.2409
38.4517
40
R
(Ω/kft)
L
(mH/kft)
C
(nF/kft)
α
(dB/kft)
749.6236
783.4812
817.8867
851.7503
890.3597
925.5264
969.5871
1010.435
1060.989
1104.402
1152.924
1191.673
1244.775
1319.738
1367.755
1441.865
1488.43
0.134382
0.134252
0.13412
0.133985
0.133846
0.133706
0.133561
0.133416
0.133265
0.133115
0.132962
0.132811
0.132654
0.132489
0.132332
0.132167
0.132058
13.85377
13.84553
13.83685
13.82779
13.81817
13.80831
13.79778
13.78702
13.77562
13.76416
13.75225
13.74044
13.72787
13.71443
13.70164
13.68794
13.67882
33.0328382
34.5312289
36.0541353
37.5534985
39.2624684
40.819977
42.7701295
44.5789511
46.816403
48.7392081
50.8878273
52.6053964
54.9569524
58.2741802
60.4020976
63.682754
65.7446239
Z0
(Ω)
98.48886
98.47053
98.45274
98.43547
98.4187
98.40242
98.38661
98.37127
98.35637
98.34191
98.32787
98.31424
98.30101
98.28816
98.27569
98.26358
98.25571
This is a draft document and thus, is still dynamic in nature.
107
T1E1.4/2000-002R6
ANSI T1.417
Table B. 15 - Cable parameters, two-pair quad drop
MHz
0.772
0.82
0.87
0.923
0.979
1.038
1.103
1.169
1.241
1.317
1.396
1.483
1.572
1.668
1.771
1.877
1.993
2.114
2.243
2.381
2.523
2.68
2.843
3.016
3.201
3.393
3.604
3.822
4.055
4.304
4.562
4.845
5.139
5.453
5.788
6.133
6.515
6.91
7.331
7.782
8.247
8.76
9.291
9.857
10.46
11.09
11.78
12.49
13.25
14.07
108
R
(Ω/kft)
Pair 1
125.4
130.1
135
139.5
143.4
148.3
153
158.1
163.8
170.3
176.3
180.7
187.2
193.5
200.4
207.1
214
221.6
229.6
237.2
245.6
254.9
263.5
273.1
283.3
293.1
303.9
314.7
326.9
338.9
351.8
364.6
377.6
391.6
407
424.1
439.6
456.8
474.7
494.7
513.7
533.8
553.8
578.8
596.9
620.5
646.7
672.1
701.4
728.4
Pair 2
129.2
134.7
139.9
145.4
149
154.1
158.3
164.2
170.3
176.6
182.6
188.3
194.3
202.3
209.2
215.6
223
232.1
239.2
248.3
257.7
267.4
276.3
287.4
297.2
307.7
318.5
331.8
343.3
355.5
370.7
384.7
400.6
413.7
429.1
445.9
463.9
482.3
501.6
520.4
542.6
565.7
594.5
610.9
635.1
661.2
685.5
712.6
740.6
772.8
L
(mH/kft)
Pair 1
0.156
0.14
0.145
0.146
0.146
0.146
0.146
0.142
0.146
0.146
0.146
0.146
0.146
0.146
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.143
0.143
0.143
0.142
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.143
Pair 2
0.178
0.141
0.148
0.149
0.15
0.151
0.151
0.151
0.148
0.153
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.15
0.149
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
C
(nF/kft)
Pair 1
12.53
11.28
11.67
11.75
11.79
11.81
11.82
11.5
11.84
11.84
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.77
11.79
11.83
11.83
11.83
11.83
11.83
11.83
11.83
11.83
11.83
11.82
Pair 2
13.69
10.83
11.38
11.51
11.57
11.6
11.62
11.63
11.43
11.75
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.65
11.59
11.46
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
α
(dB/kft)
Pair 1
4.879
5.065
5.259
5.439
5.592
5.786
5.971
6.174
6.402
6.658
6.896
7.07
7.33
7.579
7.851
8.118
8.39
8.691
9.01
9.312
9.642
10.01
10.35
10.74
11.14
11.53
11.96
12.39
12.87
13.34
13.85
14.36
14.88
15.43
16.04
16.72
17.34
18.02
18.73
19.52
20.27
21.07
21.87
22.85
23.57
24.51
25.55
26.55
27.72
28.79
Pair 2
4.923
5.132
5.331
5.54
5.676
5.87
6.031
6.256
6.487
6.729
6.957
7.175
7.404
7.71
7.969
8.214
8.497
8.843
9.116
9.46
9.819
10.19
10.53
10.95
11.32
11.72
12.13
12.64
13.08
13.54
14.12
14.66
15.26
15.76
16.35
16.99
17.68
18.38
19.11
19.83
20.67
21.55
22.65
23.28
24.2
25.19
26.12
27.15
28.22
29.44
This is a draft document and thus, is still dynamic in nature.
Z0
(Ω)
Pair 1 Pair 2
111.5
113.9
111.5
113.9
111.4
113.9
111.3
113.9
111.3
113.9
111.2
113.9
111.2
113.9
111.1
113.9
111.1
113.9
111
113.9
111
113.9
110.9
113.9
110.9
113.9
110.8
113.9
110.8
113.9
110.7
113.9
110.7
113.9
110.6
113.9
110.6
113.9
110.6
113.9
110.5
113.9
110.5
113.9
110.5
113.9
110.4
113.9
110.4
113.9
110.4
113.9
110.3
113.9
110.3
113.9
110.3
113.9
110.2
113.9
110.2
113.9
110.2
113.9
110.2
113.9
110.1
113.9
110.1
113.9
110.1
113.9
110.1
113.9
110
113.9
110
113.9
110
113.9
110
113.9
109.9
113.9
109.9
113.9
109.9
113.9
109.9
113.9
109.9
113.9
109.9
113.9
109.8
113.9
109.8
113.9
109.8
113.9
T1E1.4/2000-002R6
ANSI T1.417
Table B. 15 (concluded)
MHz
14.91
15.84
16.8
17.82
18.92
20.05
21.29
22.58
23.96
25.43
26.95
28.63
30.37
32.22
34.2
36.24
38.5
40
R
(Ω/kft)
Pair 1
757.8
789.1
821.2
854.9
892.9
925.2
964.6
1012
1053
1104
1142
1194
1229
1306
1352
1395
1482
1530
Pair 2
804.1
837.8
872.1
906.3
945.9
991.2
1028
1078
1120
1169
1226
1283
1319
1378
1439
1507
1555
1622
L
(mH/kft)
Pair 1
0.143
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
Pair 2
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
C
(nF/kft)
Pair 1
11.82
11.82
11.82
11.82
11.82
11.82
11.82
11.81
11.81
11.81
11.81
11.81
11.81
11.81
11.8
11.8
11.8
11.8
Pair 2
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
α
(dB/kft)
Pair 1
29.95
31.2
32.47
33.81
35.32
36.6
38.16
40.03
41.66
43.69
45.19
47.26
48.65
51.71
53.53
55.26
58.69
60.6
Pair 2
30.64
31.92
33.23
34.53
36.04
37.77
39.18
41.08
42.67
44.56
46.71
48.9
50.26
52.49
54.83
57.4
59.24
61.81
This is a draft document and thus, is still dynamic in nature.
Z0
(Ω)
Pair 1 Pair 2
109.8
113.9
109.8
113.9
109.8
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.7
113.9
109.6
113.9
109.6
113.9
109.6
113.9
109.6
113.9
109.6
113.9
109.6
113.9
109.6
113.9
109
T1E1.4/2000-002R6
ANSI T1.417
Table B. 16 - Cable parameters, flat-pair Drop
MHz
0.772
0.82
0.87
0.923
0.979
1.038
1.103
1.169
1.241
1.317
1.396
1.483
1.572
1.668
1.771
1.877
1.993
2.114
2.243
2.381
2.523
2.68
2.843
3.016
3.201
3.393
3.604
3.822
4.055
4.304
4.562
4.845
5.139
5.453
5.788
6.133
6.515
6.91
7.331
7.782
8.247
8.76
9.291
9.857
10.46
11.09
11.78
12.49
13.25
14.07
14.91
15.84
110
R
(Ω/kft)
L
(mH/kft)
C
(nF/kft)
α
(dB/kft)
170.5
173.7
177.6
182.4
188
194.5
201.8
209.9
219
228.9
239.8
251.5
264.2
277.8
292.3
307.8
324.2
341.6
359.9
379.1
399.4
420.6
442.8
466
490.1
515.3
541.4
568.5
596.6
625.7
655.8
686.9
719.1
752.2
786.3
821.4
857.5
894.7
932.8
972
1012
1053
1095
1139
1183
1228
1274
1322
1370
1419
1469
1521
0.129
0.13
0.131
0.132
0.133
0.135
0.136
0.138
0.139
0.141
0.142
0.144
0.145
0.147
0.148
0.149
0.151
0.152
0.153
0.155
0.156
0.157
0.158
0.159
0.16
0.161
0.162
0.163
0.164
0.165
0.166
0.167
0.168
0.169
0.169
0.17
0.171
0.171
0.172
0.173
0.173
0.174
0.175
0.175
0.176
0.176
0.177
0.177
0.178
0.178
0.178
0.179
10.5
10.43
10.39
10.36
10.34
10.33
10.33
10.32
10.33
10.34
10.35
10.36
10.38
10.39
10.41
10.42
10.43
10.45
10.46
10.47
10.49
10.49
10.5
10.51
10.52
10.52
10.53
10.54
10.54
10.54
10.55
10.55
10.55
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.55
10.55
10.55
10.54
10.55
6.681
6.76
6.87
7.012
7.185
7.389
7.625
7.892
8.19
8.519
8.88
9.273
9.696
10.15
10.64
11.16
11.7
12.28
12.9
13.54
14.21
14.92
15.66
16.43
17.23
18.06
18.92
19.81
20.74
21.69
22.68
23.7
24.75
25.83
26.95
28.09
29.27
30.47
31.71
32.98
34.28
35.62
36.98
38.38
39.8
41.26
42.75
44.27
45.82
47.41
49.02
50.67
Z0
(Ω)
110.8
111.5
112.2
112.9
113.6
114.2
114.8
115.5
116.1
116.6
117.2
117.7
118.3
118.8
119.3
119.7
120.2
120.7
121.1
121.5
121.9
122.3
122.7
123.1
123.5
123.8
124.2
124.5
124.9
125.2
125.5
125.8
126.1
126.4
126.6
126.9
127.2
127.4
127.7
127.9
128.1
128.3
128.6
128.8
129
129.2
129.4
129.6
129.7
129.9
130.1
130.2
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Table B. 16 (concluded)
MHz
R
(Ω/kft)
L
(mH/kft)
C
(nF/kft)
α
(dB/kft)
16.8
17.82
18.92
20.05
21.29
22.58
23.96
25.43
26.95
28.63
30.37
32.22
34.2
36.24
38.5
40
1573
1626
1680
1736
1792
1849
1908
1967
2027
2089
2151
2214
2279
2344
2410
2455
0.179
0.18
0.18
0.181
0.181
0.181
0.182
0.182
0.182
0.182
0.183
0.183
0.183
0.184
0.184
0.184
10.55
10.54
10.55
10.54
10.54
10.54
10.53
10.53
10.53
10.53
10.53
10.52
10.51
10.52
10.51
10.5
52.34
54.05
55.79
57.57
59.37
61.2
63.07
64.96
66.89
68.85
70.84
72.86
74.92
77
79.12
80.55
Z0
(Ω)
130.4
130.6
130.7
130.9
131
131.1
131.3
131.4
131.5
131.7
131.8
131.9
132
132.1
132.2
132.3
This is a draft document and thus, is still dynamic in nature.
111
T1E1.4/2000-002R6
Z in =
AZL + B
CZL + D
Zout =
DZS + B
CZS + A
Vin = VS
Z in
ZS + Z in
Vout = VL = VS
Pin =
ANSI T1.417
ZL
AZL + B + CZS ZL + DZS
 1 
1
| Vin |2 Re 

2
 Z in 
Pout =
 1 
1
| Vout |2 Re  
2
 ZL 
Vout
ZL
=
Vin
AZL + B
Loop Insertion Loss = − 20 log 10
ZL + ZS
AZ L + B + CZ S Z L + DZ S
1
Mean Squared Loss (MSL) = − 10 log 10 
 N
Pout ( fi ) 

in ( fi ) 
i =1

N
∑P
Figure B.1 – Loop ABCD parameters, impedance and voltages
112
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
I1
ZS
Vs
I2
+
V1
–
+
–
A B 
C D 


Source
+
V2=VL
–
ZL
Load
Two-port Network
Figure B. 2 – Two-port network model.
I(x)
I(x+dx)
+
V(x)
–
Rdx
Ldx
Cdx
Gdx
+
V(x+dx)
–
Z = R + jωL
Y = G + jωC
Two-port Network
Figure B.3 – Incremental section of twisted-pair transmission line.
ZS = RS + jX S
+
–
VS
+
V
I
ZL = RL + jX L
–
Figure B.4 – Simple load circuit for power analysis
This is a draft document and thus, is still dynamic in nature.
113
T1E1.4/2000-002R6
ANSI T1.417
Source
ZS
VS
+
Line Length d
Z0 , γ
+
–
VL
ZL
–
1 ZS 
Φ0 = 

0 1 
Load
 cosh(γd )
Z0 ⋅ sinh( γd )

Φ1 =  1
⋅ sinh( γd )
cosh( γd ) 
 Z0

Source
ZS
VS
Line Length d2
Z02 , γ2
+
–
+
VL
Line Length d3
Z03 , γ3
Line Length d1
Z01 , γ1
1 ZS 
Φ0 = 

0 1 
 cosh(γ1d1) Z01⋅ sinh(γ1d1)

Φ1 =  1
 ⋅ sinh(γ1d1) cosh(γ1d1) 
Z
 01


1
0
Φ2 =  1 ⋅ tanh(γ d ) 1


2 2

 Z02
ZL
–
Z03 ⋅ sinh(γ 3d3) Load
 cosh(γ 3d3)


1
Φ3 =
cosh(γ 3d3) 
 ⋅ sinh(γ 3d3 )
Z
 03

Figure B.5 – Examples of two-port cascades for twisted-pair transmission line configurations
Same Binder Group
Transmit
NEXT
Receive
Figure B. 6 – Near end crosstalk (NEXT)
114
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Pair 1
Pair 2
Pair 3
Pair 4
Pair 5
Pair 6
Pair 7
Pair 8
Pair 9
Pair 10
Pair 11
Pair 12
Pair 13
Pair 14
Pair 15
Pair 16
Pair 17
Pair 18
Pair 19
Pair 20
Pair 21
Pair 22
Pair 23
Pair 24
Pair 25
1% Case
NEXT POWER SUM LOSS(dB)
1000 FT, 24 AWG PIC
70
N
E
X
T
60
50
40
P
S 30
L
(dB) 20
10
0
0.1
1
10
100
FREQUENCY(MHz)
Figure B.7 – NEXT power sum losses for 25 pairs of PIC cable binder group
1% NEXT POWER SUM LOSS
1000 FT, 24 AWG PIC
70
.3 - 40MHz fit
N 60
E
X 50
T
40
P
30
S
L 20
(dB)
10
Model
1.5 - 30MHz fit
0
0.1
1
10
100
FREQUENCY(MHz)
Figure B.8 – Comparison of Simplified Model NEXT with Measured NEXT
This is a draft document and thus, is still dynamic in nature.
115
T1E1.4/2000-002R6
ANSI T1.417
Same Binder Group
Transmit
FEXT
Receive
Figure B.9 – Far end crosstalk (FEXT)
1% FEXT POWER SUM LOSS
1000 FT, 24 AWG PIC
70
F
E
X
T
60
50
40
.3 - 40MHz fit
P 30
S
L 20
(dB)
10
Model
1.5 - 30MHz fit
0
0.1
1
10
100
FREQUENCY(MHz)
Figure B.10 – Comparison of Model FEXT with Measured FEXT
116
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R6
ANSI T1.417
Annex C: Probability of error estimation
(Informative)
As in all digital transmission, the most common measure of DSL performance is the probability of error. Usually,
probability of bit error is desired, but sometimes probability of symbol error is also of interest. In either case, the
engineer attempts to measure the probability of error by observation of a system’s performance – this is usually
achieved with transmission hardware through the use of a BERT. Typically, the BERT allows a test engineer to
select one of a number of different bit streams (typically various lengths of pseudo-random patterns). The BERT
essentially synchronizes to the receiver output bit pattern and compares it to the input pattern, while counting the
number of error positions. The number of errors is periodically divided by the total number of bits measured to
estimate the probability of bit error. As time increases, this average bit error rate should converge to the actual
system value (when the system is not time varying). A reset button allows bit error counts to be restarted at zero
when necessary.
In general, bit error rate measurements become more reliable with time. The designer then needs to know how long
the bit error rate needs to be observed before any derived bit error rate is sufficiently accurate. This is a basic
statistical problem that involves measurement of a distribution. Let us suppose that bit (or symbol) errors are made
with some unknown, but fixed, probability p. One measures p by counting errors in successive observations of the
channel output. Let the kth experiment be denoted by pk where
error measured ( p )
1
.
pk = 
0 no error measured (1 − p )
Then, an estimate of the probability of error, based on N independent measurements, is
pˆ (N ) =
1 N −1
⋅ ∑ pk .
N k =0
This estimate has an average value
E [pˆ (N )] = p
and a variance about this average of
σˆ 2p = var [pˆ (N )] =
p
(1 − p ) ≈ p .
N
N
Clearly, this estimate converges to the true probability of error as N gets large. However, N can be much larger than
sometimes expected. For instance, the standard deviation is the square root of the variance. Thus, for a system
where p=10-7, for the probability of error estimate to have a standard deviation of 10% of the value of p, then
10 −8 =
10 −7
N
or
N = 10 9 .
In fact, a single standard deviation may not be sufficient to guarantee good accuracy of measured probability of
error.
The distribution of the random variable pˆ (N ) has a binomial distribution given by
N 
f pˆ (k ) = Pr {N ⋅ pˆ (N ) = k } =  (1 − p )N −k p k .
k 
p
from
L
the true value with a high degree of confidence. Let us say that we desire (1 − δ ) (equal say 90%) confidence that
The test engineer desires to ensure that the probability of error estimate deviates less than an amount ε =
the measurement deviates less than ε from the true value. Corresponding to this value of ε is a range of values for
the index k (ε ) such that the estimate is close enough, mathematically stated precisely as
This is a draft document and thus, is still dynamic in nature.
117
T1E1.4/2000-002R6
{
ANSI T1.417
}
Pr p̂ (N ) − p < ε > 1 − δ =
N 
∑  k (1− p )N −k p k .
k (ε )
Clearly, just to have a non-trivial set for k (ε ) , then N ≥ L / p . Evaluation of the sum can be excessively intensive
and so a rough use of the central limit theorem is applied to the distribution to say that for large N, the distribution is
approximately Gaussian and so the probability is then approximated by
{
}
p
⌠L
Pr p̂ (N ) − p < ε ≈ 
⌡− p
L
1
2π
−
e
x2
2pL
 Np 
 = 1− δ ,
dx = 1 − 2Q 
 L 


or then
 Np  δ
= .
Q
 L  2


For 90% confidence, δ = .1 that the error is less than p/L%, then the above equation produces
N≥
14.8 ⋅ L2
,
p
so, for instance, 10% accuracy at p = 10 −7 , requires that nearly 3 billion bits be tested. Thus, at a speed of 10
Mbps, this takes about 300 seconds, or approximately 5 minutes. For 1 Mbps transmission, the test would require
3.5 hours. The measurement time can be reduced most easily by reducing L to 2, which corresponds to only about a
.2 dB SNR difference. Even then, 1 Mbps DSL transmission at 10 −7 error rate may take 2 minutes for a
measurement, while a lower speed of 100 kbps would take 20 minutes. Such measurement intervals are typical in,
for instance, performance comparison tests sponsored by standards groups like ANSI.
C.1 Effect of input bit sequence
Clearly, the input bit sequence will need to be periodic for any practical implementation of a BERT. The period of
this sequence should be such that it exceeds the memory of the transmission system significantly. Such sequence
length is necessary to ensure that all possible channel output conditions are excited. Given that DSL transmission
systems may have long memory, a standard pseudo-random sequence of length 223-1 bits and running through all
23-bit sequences once and only once per period. Some sequences with greater lengths will not have equal likelihood
of occurrence and can bias probability of error measurements, but DSL engineers usually presume this effect small.
C.2 Period of injected “Gaussian” noise
Working Group T1E1.4 studies note that most commercial line simulators make use of pseudo-random noise in
generating Gaussian noise measuring DSL performance. An unfortunate consequence is that the peak noise samples
generated do not accurately follow the Gaussian distribution tails, thus biasing probability of error measurements in
an optimistic direction. Typically, line simulators generate noise by using some internal analog noise source and
adding digitally generated noise to it. If the period of the latter digital “Gaussian” noise is M, then the peak value of
the noise in a set of M samples is also Gaussian with mean
 1 
µ = 1− Q 

 Mσ 
and variance
µ2
σ 2peak =
(M − 1) ⋅ 2πσ2 ⋅ e σ2
.
M3
To eliminate an optimistic bias, the tester would need M >107, which complicates line simulator design. For the
more typical value of M=8192, the bias is optimistic by 2.4 dB, meaning that lab measurements for M = 8192 are
then optimistic by 2.4 dB and should be reduced by such for field performance.
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C.3 dB margin and importance sampling
To avoid long measurement times, importance sampling is a method used by test engineers to test only the worstcase situations by increasing the occurrence of peak noise samples with respect to Gaussian noise. Such importance
sampling must be very carefully applied for informative results. However, DSL engineers use a form of importance
sampling in the concept of margin. Recalling that DSL systems are specified to have a probability of bit error of
10 −7 with a 6 dB margin. This means that the actual probability of error would be below 10 −24 , requiring
centuries of measurement time. Instead, testing is executed with noise increased by 6 dB so that reasonable
measurement times can be used. The margin concept is one mechanism for importance sampling. DSL engineers,
however, prefer the supposed practical interpretation that unforeseen noise disturbances of a temporarily nature will
not cause an error with such a large margin, although justification for such unforeseen noises at a level of 6 dB is
difficult (either the noise change is much smaller for crosstalk changes or much larger for impulse or temporary RF
disturbances).
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Annex D: Additional spectrum management topics currently under study
by the formulating committee of this standard
(Informative )
The formulating committee of this standard has considered several additional topics for which specific requirements
or recommendations could not be finalized in the short time available for the development of the first edition of this
standard. These topics include, but are not limited to, the following items which the formulating committee feels are
important and should be addressed in future editions of this standard:
120
−
Revision of non-DSL out-of-band metallic and longitudinal signal power limits to provide an adequate
level of protection for DSL systems.
−
Addition of VDSL to the basis systems list.
−
Extension of spectrum management class 5 upstream band to lower frequencies.
−
Methods for optimizing PSDs, maximizing throughput and binder group capacity.
−
Trade-offs between loop length guidelines and spectral characteristics.
−
The susceptibility of some deployed systems to short term stationary crosstalk.
−
Spectral compatibility with T1.419 (splitterless ADSL) basis systems.
−
Spectral compatibility of systems using intermediate TU.
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Annex E: Time varying, user data-dependent crosstalk from T1 and DDS
services
(Informative)
Both T1 AMI and 56 kbps DDS are established and growing services in North America. Thus, it is important that
the effects of these services on other services are properly considered. Specifications for T1 AMI can be found in
T1.403-1992. Specifications for DDS can be found in T1.410-1992 and AT&T Technical Reference PUB 54075.
Neither T1 nor DDS include scrambling of user data.24 As a consequence, both the spectral behavior and the time
domain behavior of T1 and DDS line signals are highly dependent on the user data being transmitted at any moment.
This behavior thus manifests itself in the crosstalk interference of T1 and DDS into other services. Such behavior
strongly contrasts with traditional stationary crosstalk models used to analyze and test subscriber loop technologies.
T1 and DDS services host many types of user data and protocols. A consequence is that user data content and data
patterns cannot be predicted nor controlled and it should be assumed that any pattern can be transmitted, that the
duration of a pattern is indeterminable and that changes from one pattern to another can occur at any moment.
Examples include bursts of “random” user data followed by idle periods consisting of HDLC flags or Ones.
One consequence of data dependency is that the transmitted power spectral density (PSD) and the signal energy in a
given frequency band can vary greatly as user data patterns change. The time duration of each PSD variant is
caused solely by the time change of user data content, and thus the time duration of each PSD variant may vary from
less than a millisecond to many hours. Changes from one PSD to any other may occur at any moment.
Figure E. 1 and Figure E. 2 show examples of stationary PSD variants for T1 and DDS. Figure E. 3 and Figure E. 4
show examples of how the power in frequency bands can vary with time. (It is cautioned that these are but
examples and are not inclusive of all possible PSD variants. Note also that other DDS data rates exist.)
Several conclusions may be drawn regarding crosstalk from DDS and T1:
1) Crosstalk can exceed that commonly modeled based on a random data assumption for T1 and BRI in certain
frequency bands by as much as 20 dB.
2) Crosstalk should be considered to be time varying. The time duration and time change of each PSD variant
is neither predictable nor controllable. It is caused by user data content.
3) Crosstalk in a wide band (for example, tens of kilohertz) can change at least 25 dB .
4) Crosstalk in a narrow band (for example 3 kHz) can change at least 45 dB.
5) The above affects all frequency bands from near DC to the highest range of T1.
24
An option for scrambling is defined in T1.410. Currently, however, it is not widely used.
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A ll Z E R O s
Q R R S “R an dom ” D ata
-2 5
-3 5
dB
-4 5
-5 5
T 1 P ow er Spectral D ensity V ariations
A ll O N E s
-95 dB m /H z
20
kH z
12 0
kH z
2 20
kH z
32 0
kH z
4 20
kH z
52 0
kH z
6 20
kH z
72 0
kH z
8 20
kH z
92 0
kH z
Figure E. 1 - Examples of T1 power spectral density variations
D D S P ow er Spec tral D e nsity V ariations
-25 dB m /H z
-35 dB m /H z
-45 dB m /H z
2047 R andom ,
all O N E s,
H D L C flag patterns
-55 dB m /H z
-65 dB m /H z
-75 dB m /H z
20
kHz
120
kHz
220
kHz
320
kHz
420
kHz
520
kHz
620
kHz
720
kHz
820
kHz
920
kHz
Figure E. 2 - Examples of DDS power spectral density variations
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Data Dependent Power Change in the 20 - 420 kHz Band
5
Random Data Level
0
-5
-10
dB
-15
-20
Any
Duration
Any
Duration
All ONEs Level
-25
-30
Time
Figure E. 3 - Data dependent power changes in a wide band due to T1 data patterns
Data Dependent Power Change in a Narrow Band at 193 kHz
-30
ZEROs Data Level
-40
-50
dBm/Hz
-60
Random Data Level
-70
ONEs Data Level
-80
-90
Time
Figure E. 4 - Data dependent power changes in a narrow band due to T1 data patterns
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Annex F: Non-continuous CO signaling events
(Informative )
F.1 Ringing
Ringing in North America is an AC voltage superimposed on a DC bias. Many installations in the US use nonsinusoidal 20 Hz ringing with a nominal 90 Vrms at the ringing source. Other frequencies in use range from 16 2/3
to 66 2/3 Hz with voltages from 85 to 135 V 25. One standard sets the maximum voltage limit to 150 Vrms26 and
notes cases where it can attain 175 Vrms.
Ringing is a non-continuous disturber. At the beginning of each ringing burst there is a transition from –48 V battery
feed to –48 V with superimposed AC ringing. Nominal interrupts are 2 seconds on and 4 seconds off. Custom
ringing cadences with multiple ringing, such as triple cadences, are common. The ringing waveform is ideally a sine
wave with its axis of symmetry shifted –48 V from zero. The ringing burst can be characterized in terms of 100's of
milliseconds as shown in Figure F. 1. In this depiction, the sine wave starts and stops in unity with the DC bias and
represents the best case relative to instantaneous power changes as a result of ring application and trip.
Elements of synchronization are related to the application of ringing in many applications, such as the use of a
common ringing bus serving hundreds of lines. Central office implementations, in many cases, simultaneously ring
multiple lines with concurrent cadence. As such, the application and withdraw of ringing is generally without regard
to the phase angle of AC energy. The peak voltage when ringing is tripped can be the sum of the DC and greatest
AC or approximately 170 V as shown in Figure F. 2.
In its worst case, a generated ringing waveform is a trapezoidal shape, which means it has higher frequency
components occurring at 25 ms intervals. Transient energies often result from gap switching in the ringing
generator as shown in Figure F. 3.
Various forms of ringing cadence exist as noted above such as "triple," "double," "long/short," "coded," and "teen
ringing.27 For example, triple ringing bursts three times within 1800 ms as shown in Figure F. 4. These have the
effect of increasing random, ring application and removal impulse effects as shown above.
Telephone Switching systems typically have the capability of ringing as many as one-fourth of the connected lines.
Accordingly, in the worst case, an average of 6 of the 25 pairs in a binder group could be in some phase of ringing
application or removal.
F.2 Supervision (hook flash)
As shown in Figure F. 5, the DC potential is applied to the customer loop through a battery-feed device consisting of
two inductive coils in series with tip and ring. An idle circuit is nominally 48 V with no current flowing.
During service initiation, the customer closes the loop and a transient voltage migration occurs within the cable pair
of greater than 40 V, that is, it drops to 6 V across the telephone set.
A sudden voltage change in the presence of distributed capacitance can couple as not all of it gets cancelled out. A
wave front of the sudden change in loop voltage is unbounded and currently unrestricted. POTS filters for DSL are
only on the pair connected to and adjacent pairs are susceptible to the type of inductive kick as described above.
This exists throughout the network today.
F.3 Dial Pulse
These are periodic transitions from on-hook to off-hook in order to convey numeric values typically at 10 pulses per
second in North America. Usually, 40 ms make (close) versus 60 ms break (open) as there is less time required to
25
GTE Customer Handbook - 500, Issue 1, 1972.
T1.401-1993, "Interface Between Carriers and Customer Installations--Analog Voicegrade Switched Access Lines
Using Loop-Start and Ground-Start Signaling."
27
T1.401.02-1995, "Interface between Carriers and Customer Installations--Analog Voicegrade Switched Access
Lines with Distinctive Alerting Features."
26
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build the magnetic flux versus lose it. As soon as the dial on the phone is turned, all of the resistance in the circuit
(all the handset circuitry) is shunted. There is a solid short in the circuit in order to get ready to go to maximum
current.
The shorter the loop the higher the current but the less the cross talk potential. This is just the opposite of longer
loops. These phenomena exist on short and longer loops.
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-138 V
nominal peak
90 Vac rms.
4200 mS
1800 mS
Figure F. 1 - Standard ringing potential with best case start/end
20Hz or 50 mS Peak to Peak
~170V worst case
- 90Vrms ac
2
- 48Vdc
0
+ 48Vdc
Figure F. 2 - Standard ringing potential worst case start/end
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T he freq uenc y co m po ne nts at the tra nsitio n ed ge o f ringing ca n be > 500H z
infinity
Tim e i+ 2
Tim e i
< .5 m S
25 m S
1 mS
Tim e i+ 1
Figure F. 3 - Ringing waveforms (worst case generalization)
1800 mS
Figure F. 4 - Triple ringing interval
48Vdc
Figure F. 5 - Simple battery feed arrangement
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Annex G: ADSL Calculated Capacities
(Informative )
The following assumptions were used in the analysis:
•
ADSL basis system performance computations were performed using the parameters defined in Annex A.
•
HDSL, HDSL2, and T1 were assumed to be repeatered. Only TU-R NEXT and TU-C FEXT is considered for
ATU-R downstream evaluations and the repeater NEXT / FEXT affects are not considered.
•
T1 disturber was assumed to be in adjacent binder with 15.5 dB reduction.
•
ISDN BRI PSD defined in T1.413-1998
•
HDSL PSD defined in T1.413-1998
•
HDSL2 PSD defined in T1.418-2000
•
T1 PSD defined in T1.413-1998
The above assumptions reflect a worst-case engineering model of conditions that approximately represent less than
or equal to 1% of the anticipated real-life loop plant. Due to the numerous unknowns including actual deployment
distributions, RFI interference, loop plant parameters, and inter-loop interference, the exact percentage of lines
seeing conditions equal to or worse than this model are not predicted. ADSL calculated capacities shown in Table
G. 1 are not intended to be performance target rates for systems in the real world.
Table G. 1: ADSL Calculated Capacity
Basis System
Performance Level
Evaluation Loop 26 AWG
Reference Disturber
# Disturbers
Performance Level Down / Up (kbps)
Technology PSD
ISDN BRI
HDSL
HDSL2
T1 (Adjacent Binder)
ADSL
A
9.0
SM3
24
4850 / 645
ADSL
B
11.5
SM2
24
3095 / 415
Calculated Capacity Down / Up (kbps)
5639 / 683
3407 / 453
515 / 125
5622 / 666
2592 / 423
59 / 66
5289 / 680
2733 / 435
172 / 69
1525 / 706
905 / 472
193 / 154
Note: All calculated data rates correspond to mixed reference case.
128
ADSL
C
15.5
SM1
24
425 / 105
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Annex H: Technology effects of and on T1 carrier systems
(Informative)
H.1 T1 Carrier
In order to assure compatibility with T1, spectrum management class 5 DSL transmission systems and T1 systems
should be assigned to pairs that are in different binder groups whenever possible.
H.1.1 Compatibility with AMI T1
The test for compatibility with repeatered AMI T1 assumes the following:
−
The repeater section is operating with 3 dB of margin.
−
The margin, after taking the proposed DSL system into account, must be at least 2.0 dB.
−
The loss of the first repeater section out of an office is assumed to be 22.5 dB at 772 kHz.
−
The loss of subsequent sections is assumed to be 32 dB at 772 kHz.
−
Power summing margins, to obtain the 2.0 dB of margin, yields a required minimum margin, due to the DSL
system(s) alone, of 9.0 dB.
It has been found empirically that, on a repeater span having a loss of 22.5 dB at 772 kHz, the maximum noise that
can be tolerated at the repeater input, while maintaining a BER of 10-7, is -27.5 dBm. The maximum noise due to
DSL system(s) on the first repeater section out of an office, then, shall be equal to or less than -36.5 dBm (-27.5 –
9.0). Similarly, the maximum noise for a 32 dB span is -40.5 dBm. The maximum noise due to DSL system(s) for
subsequent repeater sections, then, shall be equal to or less than -49.5 dBm (-40.5 – 9.0).
When evaluating the noise coupled into the repeater, the following equation, developed via curve fitting, shall be
used to model the repeater input filtering:
Gain(dB ) = a5 f 5 + a 4 f 4 + a 3 f 3 + a 2 f 2 + a1f + a0
The frequency
.
f is in MHz, and the coefficients for both the 22.5 and 32 dB sections are shown in Table H. 1.
Defining C(f) as the two-piece NEXT model (see Figure A. 1) and Gain(f) as given above, the following conditions
for compatibility with T1 carrier must be met:
1.544MHz
End Section ( 22.5dB ) :
∫ PSDDisturber ∗10
C ( f ) / 10
∗ 10Gain( f ) / 10 df ≤ −36.5dBm
0
1.544MHz
Mid − Span Section ( 32dB ) :
∫ PSDDisturber ∗10
C ( f ) / 10
∗ 10Gain( f ) / 10 df ≤ −49.5dBm
0
H.1.1.1 Evaluation Loops
Evaluation loops for T1 are 3.3-kft of 26-AWG (22.5 dB at 772 kHz) and 4.7-kft of 26-AWG (32 dB at 772 kHz).
Table H. 1: Coefficients for T1 repeater input filtering gain equation
Coefficient
A0
A1
A2
A3
A4
A5
Value for 22.5 dB section
-12.91476008173899
15.74168401196194
20.75952294972729
-36.60781681972960
13.09484055899603
-0.91231176505002
Value for 32 dB section
-21.84038057235726
40.22938541210919
-2.99965401635352
-31.38386179570797
18.63736172126514
-3.26384215013252
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Annex I: C-code
(Informative)
Table I. 1: C-code for DMT margin computation
float dmtmrgn(
float
*signal, /* array of received signal psd samples (resolution =
FDELTA Hz)*/
float *noise,
/* array of received noise psd samples (resolution =
FDELTA Hz) */
int
rate, /* desired bit rate, expressed in units of bits per second
per FDELTA ) */
int
start, /* start of DMT bandwidth (sample number) */
int
end, /* end of DMT bandwidth (sample number) */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in
linear units */
{
int
float
float
float
j, firstpass;
snr;
snr_margin ;
delcap, totcap ;
snr_margin = MAXIMUM_VALUE;
firstpass = 1;
snr_margin += MARGIN_STEP;
do {
snr_margin -= MARGIN_STEP;
/* Compute capacity */
totcap = 0.;
for (j = start; j < end; j++) {
if (in_dB) snr = sig[j]-noise[j];
else snr = 10.*log10(sig[j]/noise[j]);
delcap = log(1. + pow(10.,
.1*(snr -snr_margin-SNRGAP))) / log(2);
if (delcap > MAXBITS) delcap = MAXBITS;
if (delcap < MINBITS) delcap = 0;
totcap += delcap;
}
if (totcap > rate && firstpass) {
snr_margin +=10.; totcap=0.; }
else firstpass = 0;
} while (totcap < rate);
return (snr_margin);
}
SNRGAP, MAXBITS, MINBITS, are all adjusted based on the DMT system being evaluated.
MAXIMUM_VALUE and MARGIN_STEP are control how fast and how accurately the routine
computes margin.
MAXIMUM_VALUE is the maximum margin of interest, the integration
begins there. MARGIN_STEP defines the accuracy of the result.
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Annex J: Derivation of Equivalent Working Length Definition
(Informative )
This standard defines deployment guidelines in terms of an equivalent working length (EWL) of multi-gauge cable.
EWL is intended to provide equivalence between the length of a multi-gauge loop and that of a straight 26-AWG
loop. It is desirable that the definition chosen for EWL provide a loss at each frequency that is nearly the same for a
specific length of 26-AWG cable and an EWL of coarser gauge cable. It is also desirable to have an EWL definition
that is simple to use. The EWL definition used in this standard is based on selected Carrier Serving Area (CSA)
design guidelines and satisfies both of these objectives.
CSA design includes the following guidelines:
- The amount of 26-AWG cable (used alone or in combination with another gauge cable) may not
exceed a total length of 9-kft including bridged tap.
- For single gauge or multi-gauge cables containing only 19, 22 or 24-AWG cable, the total cable length
including bridged tap may not exceed 12-kft.
Since the deployment guidelines in this standard are intended to limit crosstalk between systems (and not qualify the
transmission parameters of a given loop,) the effects of bridged taps do not need to be represented in the definition
of EWL. When the bridged tap length is set to zero in the CSA guidelines listed above, the maximum CSA 26AWG length limit (i.e., 9-kft) can be considered equivalent to the maximum CSA coarser gauge length limit (i.e.,
12-kft). An equivalent working length (EWL) definition based on these selected CSA guidelines can then be
defined as:
EWL = L 26 +
3( L 24)
kft
4
where L26 is the total length of 26-AWG cable in the loop excluding any bridged tap and L24 is the total length of
all 19, 22 or 24-AWG cable in the loop excluding any bridged tap. All lengths are in kilofeet (kft).
The insertion loss of 9-kft of 26-AWG and 12-kft of 24-AWG are equal at approximately 260 kHz. As shown in
Figure J. 1, the loss associated with “equivalent” 24-AWG cable is somewhat less than the loss of 26-AWG cable
for frequencies less than 260 kHz and is somewhat more than the loss of 26-AWG cable for frequencies above 260
kHz. The impact of this loss difference on a technology such as ADSL is assumed to be that the ADSL tones below
260 kHz would perform somewhat better than the spectrum management model predicts while the tones above 260
kHz would perform somewhat worse than the spectrum management model predicts.
Although an EWL definition based on selected CSA guidelines is not perfect, it meets both of the objectives listed
above. It provides a length of coarse gauge cable that provides a loss at each frequency that is nearly equivalent to
the loss for a specific length of 26-AWG cable and provides an EWL definition that is not overly complicated.
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10.0
20.0
30.0
40.0
Loss (dB)
6 kft of 26 AWG
9 kFt of 26 AWG
12 kFt of 26 AWG
50.0
13.5 kFt of 26 AWG
8 kft of 24 AWG
60.0
12 kft 24 AWG
16 kft 24 AWG
18 kft 24 AWG
70.0
80.0
90.0
100.0
0
100
200
300
400
500
600
700
800
900
1000
1100
Frequency (kHz)
Figure J. 1 - Insertion Loss of 24- and 26-AWG Cable vs. Frequency
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Annex K: Interference Level Evaluation for the Case of Unequal Loop
Length for Interfering and Interfered Systems
(Informative )
K.1 General Methodology
In some instances it may be desired to evaluate the effect of the interference from one system onto another when the
two systems are deployed in the same binder but over a different distance. In this case, crosstalk from the system
with the shorter deployment distance will travel down the cable and affect the system with the longer deployment
distance. The reverse is generally not of concern because the system on the shorter loop benefits from a higher input
signal level.
The configuration in Figure K. 1 should be used when performing a computer simulation of the effect of the system
B NEXT and FEXT interference into the system A downstream receiver while system A has the longer loop reach.
The simulation set-up of Figure K. 1 assumes that all of the head-end transmitters (ATU-C, HTU-C, etc) for both
systems are co-located at a central location. It is also assumed that all of the system B upstream transmitters are colocated at the longest supported loop length. This gives a worst case view of the effect of system B interference
upon the operation of system A.
The first cable section is adjusted to cover the maximum reach distance of system B, and the second cable section is
adjusted to cover the remaining length of the test loop under consideration. The system B FEXT noise generator
should generate FEXT noise equivalent to a system B output signal passed through the FEXT coupling loss, using a
coupling length equal to the first cable section, and attenuated by the whole cable length from the system B
transmitter location to the system A receiver location (sum of first and second sections).
The configuration in Figure K. 2 should be used to simulate the effect of system B interference onto the upstream
operation of system A.
The simulation set-up of Figure K. 2 assumes that all of the head-end receivers (ATU-C, HTU-C, etc) for both
systems are co-located at a central location. In this case, the system B FEXT noise generator should generate FEXT
noise equivalent to a system B output signal passed through the FEXT coupling loss and through the cable section
from the system B transmitter location to the system A receiver location (first cable section only).
The use of a coupling length X in the FEXT computation assumes that the loop on which system B is deployed does
not extend beyond the system B TU-R location. In other words, there is no bridged tap located at the system B TUR location.
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System B
FEXT
X + Y kft H(f)
X kft Coupling
System A
TU - C
X
kft
Y
kft
System B
NEXT
System A
TU - R
AWGN Noise
Figure K. 1 - Configuration for the simulation of the effect of NEXT and FEXT from system
B into system A downstream
System B
FEXT
X kft H(F)
& Coupling
System A
TU - C
X+Y
kft
AWGN Noise
System A
TU - R
System B
NEXT
Figure K. 2 - Configuration for the simulation of the effect of NEXT and FEXT from system
B into system A upstream
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Annex L: Interference Level Evaluation for Applications that locate a TU at
an Intermediate point between the CO and CI
(Informative )
L.1 Applications that use Intermediate Repeaters/Amplifiers
In some instances it may be desired to evaluate the effect of the interference from one system to another when the
two systems are deployed in the same binder but one of the systems uses intermediate repeaters/amplifiers between
the CO and CI. In this case, crosstalk from the repeatered/amplified system will travel down the cable and affect the
non-repeatered system. The reverse is generally not of concern because the repeatered/amplified system benefits
from a higher input signal level as a result of the signal being regenerated at an intermediate point between the CO
and CI.
The configuration in Figure L. 1 shows the sources of the crosstalk that are represented in the simulation model for
the basis system downstream receiver. The model in Figure L. 2 should be used when performing a computer
simulation of the effect of the repeatered/amplified system new technology NEXT and FEXT interference into the
basis system downstream receiver. The equations for determining NEXT and FEXT noise are defined in section B.4.
The simulation set-up of Figure L. 2 assumes that all of the head-end transmitters (ATU-C, HTU-C, etc) for both
systems are co-located at a central location. It is also assumed that all of the customer premise transmitters (ATU-R,
HTU-R, etc) for both systems are co-located at the same location. This follows the assumption that binders are
contiguous for the purposes of demonstrating spectral compatibility.
The first cable section is adjusted to cover the distance from the CO based TU-C to the intermediate
repeater/amplifier, and the second cable section is adjusted to cover the remaining length of the test loop under
consideration. The new technology NEW-C FEXT noise is equivalent to a NEW-C output signal passed through the
FEXT coupling loss, using a coupling length equal to the first cable section. The NEW-C FEXT noise is attenuated
by the second cable length. The new technology intermediate TU-C FEXT noise is equivalent to the intermediate
TU-C output signal passed through the FEXT coupling loss, using a coupling length equal to the second cable
section. The new technology intermediate TU-R NEXT noise is determined by the intermediate TU-R output signal.
The intermediate TU-R NEXT noise attenuated by the second cable length.
The configuration in Figure L. 3 shows the sources of the crosstalk that are represented in the simulation model for
the basis system upstream receiver. The model in Figure L. 4 should be used when performing a computer
simulation of the effect of the repeatered/amplified system new technology NEXT and FEXT interference into the
basis system upstream receiver. The equations for determining NEXT and FEXT noise are defined in section B.4.
The simulation set-up of Figure L. 4 assumes that all of the head-end transmitters (ATU-C, HTU-C, etc) for both
systems are co-located at a central location. It is also assumed that all of the customer premise transmitters (ATU-R,
HTU-R, etc) for both systems are co-located at the same location. This follows the assumption that binders are
contiguous for the purposes of demonstrating spectral compatibility.
The first cable section is adjusted to cover the distance from the CO based TU-C to the intermediate
repeater/amplifier, and the second cable section is adjusted to cover the remaining length of the test loop under
consideration. The new technology NEW-R FEXT noise is equivalent to a NEW-R output signal passed through the
FEXT coupling loss, using a coupling length equal to the second cable section. The NEW-R FEXT noise is
attenuated by the first cable length. The new technology intermediate TU-R FEXT noise is equivalent to the
intermediate TU-R output signal passed through the FEXT coupling loss, using a coupling length equal to the first
cable section. The new technology intermediate TU-C NEXT noise is determined by the intermediate TU-C output
signal. The intermediate TU-C NEXT noise is attenuated by the first cable length.
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Ref - C
Ref - R
TU - C
TU - R
New - C
TU-R TU-C
Y kft
New - R
Z-Y kft
Z kft
Figure L. 1 - Crosstalk into Basis System Downstream Receiver: NEXT & FEXT with
Repeater Device
Rptr/Amp
TU-R
NEXT
TU - C
New - R
NEXT
Reference
NEXT
Z-Y
kft
FSAN
Z kft
TU - R
FSAN
New - C
FEXT
Z kft H(f)
Y kft Coupling
Rptr/Amp
TU-C FEXT
Z-Y kft H(f)
& Coupling
Reference
FEXT
Z kft H(f)
& Coupling
AWGN Noise
Figure L. 2 - Simulation Model for Reference and New Crosstalk into Downstream
Receiver with Repeater Device
136
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ANSI T1.417
Ref - C
Ref - R
TU - C
TU - R
New - C
TU-R TU-C
Y kft
New - R
Z-Y kft
Z kft
Figure L. 3 - Crosstalk into Basis System Upstream Receiver: NEXT & FEXT with
Repeater Device
Reference
NEXT
New - C
NEXT
FSAN
Y kft
TU - C
Rptr/Amp
TU-C
NEXT
Z kft
TU - R
FSAN
AWGN Noise
Reference
FEXT
Z kft H(f)
& Coupling
Rptr/Amp
TU-R FEXT
Y kft H(f)
& Coupling
New - R
FEXT
Z kft H(f)
Z-Y kft Coupling
Figure L. 4 - Simulation Model for Reference and New Crosstalk into Upstream Receiver
with Repeater Device
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L.2 Applications that use Intermediate TU-C Devices
In some instances it may be desired to evaluate the effect of the interference from systems that use intermediate TUC devices between the CO and CI to another system when the two systems are deployed with the intermediate TU-C
integrated into the same binder at some intermediate point between the CO and the CI such as may be the case in
DLC deployments. With appropriate power level adjustments made at the injection points, this model can
accommodate variations of architecture such as implementation of power cut-back, introduction of significant cable
length between the intermediate TU-C device and the injection point, etc. In this case, crosstalk from the
intermediate TU-C system will travel down the cable and affect the CO based system. The reverse is generally not
of concern because the intermediate TU-C system benefits from a higher input signal level as a result of the signal
being generated at the intermediate TU-C device.
The configuration in Figure L. 5 shows the sources of the crosstalk that are represented in the simulation model for
the basis system downstream receiver. The model in Figure L. 6 should be used when performing a computer
simulation of the effect of the intermediate TU-C system new technology NEXT and FEXT interference into the
basis system downstream receiver. The equations for determining NEXT and FEXT noise are defined in section B.4.
The simulation set-up of Figure L. 6 assumes that the head-end transmitters (ATU-C, HTU-C, etc) for the two
systems are not co-located at a central location. It is assumed that all of the customer premise transmitters (ATU-R,
HTU-R, etc) for both systems are co-located at the same location. This follows the assumption that the intermediate
TU-C will integrate with the CO based binders at an intermediate point between the CO and the CI such as may be
accomplished by the Serving Area Interface (SAI). For this analysis the binders are contiguous for the purposes of
demonstrating spectral compatibility with the exception of the intermediate TU-C integration.
The first cable section is adjusted to cover the distance from the CO based TU-C to the intermediate TU-C, and the
second cable section is adjusted to cover the remaining length of the test loop under consideration. The new
technology NEW-C FEXT noise is equivalent to the NEW-C output signal passed through the FEXT coupling loss,
using a coupling length equal to the second cable section.
The configuration in Figure L. 7 shows the sources of the crosstalk that are represented in the simulation model for
the basis system upstream receiver. The model in Figure L. 8 should be used when performing a computer
simulation of the effect of the intermediate TU-C system new technology NEXT and FEXT interference into the
basis system upstream receiver. The equations for determining NEXT and FEXT noise are defined in section B.4.
The simulation set-up of Figure L. 8 assumes that the head-end transmitters (ATU-C, HTU-C, etc) for the two
systems are not co-located at a central location. It is assumed that all of the customer premise transmitters (ATU-R,
HTU-R, etc) for both systems are co-located at the same location. This follows the assumption that the intermediate
TU-C will integrate with the CO based binders at an intermediate point between the CO and the CI such as may be
accomplished by the Serving Area Interface (SAI). For this analysis the binders are contiguous for the purposes of
demonstrating spectral compatibility with the exception of the intermediate TU-C integration.
The first cable section is adjusted to cover the distance from the CO based TU-C to the intermediate TU-C, and the
second cable section is adjusted to cover the remaining length of the test loop under consideration. The new
technology NEW-R FEXT noise is equivalent to a NEW-R output signal passed through the FEXT coupling loss,
using a coupling length equal to the second cable section. . The NEW-R FEXT noise is attenuated by the first cable
length. The new technology NEW-C NEXT noise is determined by the NEW TU-C output signal. The NEW TU-C
NEXT noise is attenuated by the first cable length.
138
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Ref - C
Ref - R
TU - C
TU - R
New - C
Y kft
New - R
Z-Y kft
Z kft
Figure L. 5 - Crosstalk into Basis System Downstream Receiver: NEXT & FEXT with
Intermediate TU-C Device
New - R
NEXT
Reference
NEXT
FSAN
TU - C
Z kft
TU - R
FSAN
New - C
FEXT
Z-Y kft H(f)
& Coupling
Reference
FEXT
Z kft H(f)
& Coupling
AWGN Noise
Figure L. 6 - Simulation Model for Reference and New Crosstalk into Downstream
Receiver with Intermediate TU-C Device
This is a draft document and thus, is still dynamic in nature.
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ANSI T1.417
Ref - C
Ref - R
TU - C
TU - R
New - C
Y kft
New - R
Z-Y kft
Z kft
Figure L. 7 - Crosstalk into Basis System Upstream Receiver: NEXT & FEXT with
Intermediate TU-C Device
Reference
NEXT
New - C
NEXT
FSAN
Y kft
TU - C
Z kft
TU - R
FSAN
AWGN Noise
Reference
FEXT
Z kft H(f)
& Coupling
New - R
FEXT
Z kft H(f)
Z-Y kft Coupling
Figure L. 8 - Simulation Model for Reference and New Crosstalk into Upstream Receiver
with Intermediate TU-C Device
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Annex M: Rationale behind General Conformance Criteria in Clause 6.1
(Informative)
M.1 Introduction
The spectrum management (SM) classes were developed to reduce the myriad of potential PSDs on the loop to a few
general classifications, which could then be used in the spectrum management process. In order to develop
appropriate deployment guidelines for each class, a representative PSD, called a template, for each class was
needed. Since this template was used in spectral compatibility calculations involving the SM class, it needed to
accurately represent the worst case crosstalk that could occur when mixtures of various class-compliant transmitters
share the same loop plant. If it is too pessimistic, it will overestimate the effect that crosstalk from class members
have on basis systems; if it is too optimistic, it may under-estimate the interference into basis systems. This
represents a significant challenge, particularly because the spectrum management classes were to be “technology
independent,” that is, several known DSL technologies with significant spectral differences would fit into the same
class. Therefore, the classes would not be optimized for a particular technology (i.e. 2B1Q or QAM), but should
serve both equally well.
Traditionally, compliance to a PSD specification has been through the use of a spectral mask or hard upper limit.
Since several disparate DSL technologies fit into a single class, a PSD mask for a SM class must be fairly loose. If
such a PSD mask were also used in the role of PSD template, the amount of crosstalk into the basis systems would
be significantly over-estimated. Therefore, after much consideration, it was decided that a SM class would have
both a PSD mask and PSD template. The PSD mask would be nominally 2.5 dB above the template, with another 1
dB thrown in to account for equipment variations and measurement inaccuracies. The resulting measurement mask
is a hard limit that is 3.5 dB above the template. This also was consistent with how the PSD mask and nominal
PSDs for ADSL are defined in T1.413-1998.
Since the mask is relatively loose, additional constraints were introduced to ensure that crosstalk from class
members would be accurately modeled by the PSD template. Starting with the sliding window average power
constraint that was used for the high frequency limits in T1.413-1998, an additional “soft” limit was defined for each
SM class. This “soft” limit requires that the average PSD, over a frequency window of 100 kHz, of a class member
should be no larger than the average PSD of the template. This requirement ensures that over any 100 kHz band of
frequencies, the effect of the PSD of a compliant system is approximately the same as that of the PSD template.
The 100 kHz bandwidth was chosen since it was approximately equal to the ADSL upstream bandwidth.
The actual form of the requirement is that the average of the PSD normalized to the template (the average
normalized PSD) should be no greater than 0 dB. As with the PSD mask, 1 dB was added to the limit to allow for
variations in products and measurement accuracy, which is reflected in the equations of clause 6.1.
In addition to the average normalized PSD limit, a second additional requirement on the maximum total transmitted
power was added to ensure that excessive interference into narrow band receivers such as BRI would not occur.
This requirement was necessary since the 100 kHz window would not adequately protect systems with a bandwidth
significantly narrower than 100 kHz. This power limit was chosen to be 14 dBm, the same as the maximum
transmitted power allowed for BRI.
The average normalized PSD requirement is a new concept that was developed as a part of the development of this
standard. Since it is not well known outside of this work, a description of its derivation is provided in this annex.
M.2 Rationale for using the Average Normalized PSD as a “soft” limit
In this section we present the rationale for using the average normalized PSD limit as a requirement for conformance
to a SM class. The development uses the concept of equivalent noise, which is presented first.
M.2.1 Derivation of Equivalent Noise for DFE and DMT receivers
The performance of a majority of modern DSL receivers can be estimated through use of either the optimal decision
feedback equalizer (DFE) margin calculation or capacity calculations for DMT systems, as shown in Annex A. The
optimal DFE margin calculations can be written in the form
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
ln (1 + f _ SNR( f ) )df   − SNR _ reqDB
W


 1
MargindB = 10* log10  exp 
W

∫
(1)
where W is the Nyquist Bandwidth. The DMT capacity can be calculated by using


2


S
(
f
)
H
(
f
)
 df
C=
log 2  1 +
 9.75 −CG + MargindB  

DMTBWUSED


10

 
N( f ) 10


∫
(2)
where the DMTBWUSED does not contain any frequencies where the C ′( f ) is zero (see A.2.6), and thus is the
portion of the available DMTBandwidth that is actually used by data carrying carriers ( CG is the assumed Coding
Gain.) If we approximate the argument of the log function by dropping the “1+” portion, the DMT calculation can
be rewritten in a form that approximates the margin for a given data rate:

1
MargindB ≅  
 DMTBW
USED


10 log10 (SNR( f ) )df   − K 1

DMTBWUSED

∫
(3)
where
K 1 = ( 9.75 − CG ) −
C
10 ln( 2)
ln( 10) DMTBWUSED
(4)
and SNR is defined as in equation (7). K 1 can be thought of as the required SNR for a given data rate C .
We can also rewrite (1) to eliminate the inverse operations of natural logarithm and exponential to produce a much
simpler equation for the DFE margin calculation:
 1
MargindB =  
 W

10 log10 (1 + f _ SNR( f ) )df   − SNR _ reqDB
W

∫
(5)
Now, using (3) and (4) and assuming the folded SNR ( f _ SNR ) is significantly greater than 1 over the Nyquist
band, we can come up with a single equation that approximates the theoretical performance of both DFE based
single carrier transceivers and DMT transceivers:
′ =
MargindB
1
Bandwidth
∫
Bandwidth
10* log10 (SNR( f ) )df − SNRrequired ( dB )
(6)
′ , is the approximate margin in dB, Bandwidth is the Nyquist bandwidth for single carrier
where M arg indB
systems and the effective bandwidth of the utilized carriers for DMT, and
SNR( f ) =
S( f ) H ( f )
N( f )
2
.
(7)
S( f ) is the transmitted PSD, H ( f ) is the loop insertion gain, and N( f ) is the noise PSD present at the receiver.
Since we have assumed that the SNR is large (which eliminated the “1+” term), the signal, loss and noise terms
become separable. We then can define the equivalent transmitted signal as
ES =
1
Bandwidth
∫
10* log 10(S( f ) )df
(8)
20 * log 10( H( f ) )df
(9)
Bandwidth
the equivalent (insertion) loss of the loop as
EL =
−1
Bandwidth
∫
Bandwidth
and the equivalent noise on the loop as
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10* log 10(N ( f ) )df
(10)
MargindB ≈ ( ES − EL − EN ) − SNR required ( dB )
(11)
EN =
1
Bandwidth
∫
Bandwidth
Then, the margin calculation in (5) can be approximated by
M.2.2 Criterion for equivalency of SM Class PSD template and member PSD
From (10) and (11) above, we can see that over a given bandwidth, the composite noise input to a DFE or DMT
receiver affects performance as the average of log of the PSD (or average in dB). We define the average in dB or
AvgdB function in the frequency interval (f , f + ∆f ) as follows:
AvgdBf , ∆f (N (f )) =
1
∆f
f + ∆f
∫ 10* log
f
10
(N (f ′))df ′
(12)
We wish to ensure that crosstalk from members of a spectrum management class and crosstalk resulting from the
template has the same effect on DSL receivers. Our approach here is to ensure that the equivalent noise ( AvgdB )
of the two sources of crosstalk are approximately the same over any frequency interval (f , f + ∆f ) , where ∆f is
chosen based on the bandwidth of the receivers of the basis systems.
We define the effective crosstalk modeling error between these two crosstalk sources as
E f ,f + ∆f




 N



= AvgdBf , ∆f
C n j Pj (f ) − AvgdBf , ∆f  C  N  (PT (f ))


  ∑n j 

 j =1

  j =1 



 
∑ (
)
(13)
where the summation represents the FSAN noise summation method described in B.4.3 (extended to N crosstalk
types), N is the number of distinct PSD types, ni is the number of interferers of type j , Pj ( f ) is the PSD of
interferer type j , PT ( f ) is the PSD of the template, and C m ( f ) is the crosstalk coupling function for m
disturbers.
The first term in this expression is the equivalent noise of the crosstalk that results from a combination of N class
members, while the second term is the equivalent noise of the crosstalk that results from the same number of
“template” interferers. If we substitute the formulas for crosstalk and crosstalk summation (from B.4) into (13), and
then using the rule of logs to eliminate common terms, we obtain the following expression:
E f ,f + ∆f =
1
∆f
∫
f + ∆f
f
 N
1
1 / 0.6 
6 ⋅ log10  ∑ n i Pi (f ′)
 df ′ −
∆f
 i =1

∫
f + ∆f
f

  N  0. 6
10 ⋅ log10   ∑ n i  PT (f ′)  df ′ (14)

  i =1 


This error represents the difference between the effective crosstalk of the combination of actual disturbers vs. that
obtained by substituting the template for the PSD of each disturber. (Note that this equation assumes the FSAN
crosstalk combination method; an equivalent equation can be derived using the mean PSD method.) At this point
we can make one observation: the expression does not include a frequency dependent crosstalk coupling term. This
is a feature of the log function, and points out that exceeding the template at any portion of the averaging interval
has the same effect – the penalty is not frequency dependent.
(This is not to say that an average that includes the term is not without merit. An average of the transmitted PSD
weights the larger values of the PSD more heavily. An average that includes the C ( f ) is averaging the crosstalk
power, and weights the largest crosstalk values more heavily. This type of averaging may be more accurate when
assessing the impact of crosstalk into non-DFE or non-DMT type transceivers. However, we believe that the vast
majority of cases will involve DFE or DMT type transceivers, and are better served by the equivalent noise approach
used here.)
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M.2.3 Developing Compliance Criteria to minimize the error
Equation (14) is the general form of the error using the FSAN summing technique. Unfortunately, it is not very
intuitive. Therefore, in order to gain some insight into what types of compliance criteria will minimize the error, it
is useful to consider a special case, that of non-overlapping spectra. This case is perhaps the worst case situation,
and is good for illustrating the principles involved.
Consider the case where the individual PSDs are non-overlapping (or one is large enough to dominate the others in a
specific frequency interval.) We divide the interval (f , f + ∆f ) up into N sub-intervals, and each of the Pj ( f ′) are
non-zero in only one unique sub-interval.
Then we can reduce the first term of (14) as follows:
1
∆f
N
f j + ∆f j
j =1
fj
∑∫
 N

1
1/ 0.6 
6 ⋅ log 10  ni Pi (f ′)
df ′ ≅


∆f
 i =1

∑
N
f j + ∆f j
j =1
fj
∑∫
(
6 ⋅ log 10 n j Pj (f ′)
1/ 0.6
)df ′
(15)
where the approximation assumes that a different Pj ( f ) dominates in every sub-interval and becomes equality when
the Pj ( f ) are truly non-overlapping. This can be further reduced to
∆f j
1
6 ⋅ log10 (n j ) +
∆f
∆f
j =1
N
∑
N
f j + ∆f j
j =1
fj
∑∫
(
)
10 ⋅ log10 Pj (f ′) df ′
(16)
which we substitute into (14) along with the AvgdB function from (12) to obtain
Ef ,f + ∆f =
1
∆f
∑ ∆f j ⋅ (6 ⋅ log10 (n j ) + AvgdBf ,f +∆f (Pj (f ))) − 6 ⋅ log10  ∑ ni  − AvgdBf ,f +∆f (PT (f ))

N
j
j =1
j
j
N

 i =1

(Note: equality holds in this equation for non-overlapping PSD only)
(17)
Examples:
Let’s first consider the case where the template is a constant over a frequency interval, and the class member PSD
are non-overlapping. Without loss of generality, we set the constant template PSD, PT = 1 , so that
AvgdB( PT ) = 0 dB.
Now, we break the interval up into two non-overlapping sub-intervals, such that
∆f1 + ∆f 2 = ∆f . Now n1 = n 2 = 1, and (17) becomes
E f ,f + ∆f =
∆f1
∆f
P1( dB ) + 2 P2( dB ) − 6 ⋅ log10 (2)
∆f
∆f
(18)
where P1( dB ) and P2( dB ) are the constant PSD levels over their respective intervals in dB/Hz. This situation is
illustrated in Figure M. 1 (amplitude scale is linear).
P1
P2
PT
0
∆f1
∆f2
∆f
Figure M. 1 - Non-Overlapping PSD example
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Candidates for Compliance Criteria: Now we want to determine what kind of constraints on the Pj will keep the
maximum error small. First, let’s consider a limit on the average power in the interval. (We will refer to this as the
“average power constraint.”) In particular, we limit the average power of each interferer to be no larger than that of
the template:
1
∆f
f + ∆f
f + ∆f
∫ P (f ’ )df ′ ≤ ∆f ∫ P (f ’ )df ′
f
j
1
f
T
(19)
for all f . If we let each interferer attain the power equality with the template on the interval, we get the relationship
∆fi ( Pi ( dB ) / 10 )
10
= PT = 1 ,
∆f
(20a)
since all of our PSD are non-zero each on only one sub-interval. Therefore,
 ∆f
Pi ( dB ) = 10 ⋅ log10 
 ∆fi

 .

(20b)
and we can write (18) as
E =
 ∆f  ∆f − ∆f1
 ∆f 
∆f1
 +
 − 6 ⋅ log10 (2)
10 ⋅ log10 
10 ⋅ log10 
∆f
∆f
 ∆f1 
 ∆f − ∆f1 
(21)
∆f1
. We can see that the error has a maximum
∆f
of 1.2 dB, and goes negative as the split point approaches the interval endpoints. Positive values of the error mean
that the template under-represents the actual crosstalk effect by that amount. Negative values mean that the template
over-represents the actual crosstalk. The maximum under-representation occurs when the sub-intervals evenly
divide the interval.
The error for the even sub-division can be extended to equal numbers of N crosstalkers, to give us
since ∆f1 + ∆f 2 = ∆f . Figure M. 2 shows the error as a function of
E = 10 ⋅ log10 (N ) − 6 ⋅ log 10 (N ) = 4 ⋅ log10 (N ) ,
(22)
which grows without bound. For N = 49 , the error is 6.76 dB. This example shows that for a “worst-case”
situation, the average power constraint alone is not sufficient to ensure that the template accurately represents the
class. That is because as the sub-intervals get smaller, the peak PSD rises without limit. To eliminate this problem,
add a second constraint, a peak-PSD limit. This is a limit above the PSD of the template. For this non-overlapped
case, a limit of 3 dB reduces the largest error to that of the 2 disturber case, 1.2 dB, and a limit of 2.5 dB reduces the
largest error to 0.7 dB. However, with the peak-PSD limit, the non-overlapping case no longer allows all of the
transmitters to use the full power available in the frequency interval, so we must return to equation (14) and use
overlapped PSDs to accurately calculate the error under these conditions.
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1.5
1
Error (dB)
0.5
0
-0.5
-1
-1.5
-2
0
0.2
0.6
0.4
0.8
1
Relative split point between two sub-intervals
Figure M. 2 - Error for non-overlapping Spectra
To see the effect of the peak-PSD constraint combined with an average power constraint, we modify our example
slightly. We let the PSD equal the PSD limit on a sub-interval, and set the PSD to a lower value ( PLOW ) elsewhere.
The subinterval is sized so that the average power over the interval equals that of the template. This is illustrated in
Figure M. 3.
P1( f ) = PLIMIT over the interval ( f , f + ∆f1 ) , and P2 ( f ) = PLIMIT over the interval
( f + fOffset , f + fOffset + ∆f1 ) . The offset is varied from 0 (full overlap) to ( ∆f − ∆f1 ) (minimal overlap.)
P1=PLIMIT
P1=PLOW
0
P2=PLIMIT
P2=PLOW
0
fOffset
∆f1
∆f
Figure M. 3 - PSD Limit Example
First, let’s consider the case of the DMT ADSL standard. In that case, the individual subcarriers can be varied from
a low of -14.5 dB to a high of +2.5 dB with respect to the template (-40 dBm/Hz.) So for our simplified case, we set
PLOW = -14.5 dB and PLIMIT = +2.5 dB. Then to achieve an average power equal to the template,
∆f1 ( 2.5 / 10 ) ∆f − ∆f1 ( −14.5 / 10 )
10
+
10
= PT = 1.
∆f
∆f
(23)
Solving this equation, we find ∆f1 ∆f to be .5534. This corresponds to the case where all carriers in the ∆f1 band
are transmitting at the maximum value and all other carriers are transmitting at their minimum values. Figure M. 4,
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Case 1, shows the error as a function of normalized fOffset ( ∆fOffset ∆f ). From the plot, we can see that the
maximum error occurs when the offset is maximized; this is the case where the overlap of the high-PSD regions is
minimized. The maximum value of the error is 0.89 dB, while the average value of the error over all offset values is
-2.1 dB.
2
1
Error (dB)
0
-1
-2
-3
Case 1
Case 2
-4
Case 2A
Case 3
-5
-6
0
Case 4
0.05
0.1
0.15
0.2 0.25 0.3
Normalized fOffset
0.35
0.4
0.45
Figure M. 4 - Error for 4 Example Cases with PSD Limit
Case 2: We next add a third interferer into the mix, this one at the template value for the whole interval. The error
is shown as Case 2 in Figure M. 4. The peak error decreases to 0.64 dB, while the average error increases to 0.21
dB. Case 2A shows the case where the third interferer is at the PSD limit for the center portion of the interval. In
this case, the peak error is 0.75 dB, while the average is -1.4 dB. This gives us reasonable confidence that the worst
case peak error is given by the minimal overlap 2 disturber case.
Case 3: To generalize our results beyond DMT, we next consider the case where PLOW = 0 . With a lower limit of
zero, only a portion of our previous analysis provides meaningful results. If we revisit Case 1, only the maximum
value of fOffset allows the interval to be spanned by a non-zero composite PSD. In that case, the error increases very
slightly (0.03 dB) over the DMT case. When we add in a third interferer at the template value, we can vary fOffset
and the result is shown as Case 3 in Figure M. 4. As with the 2-disturber case, the error is only slightly larger, about
0.037 dB for all offsets. (The allowed offset range is slightly smaller than for Case 2, because ∆f1 is slightly smaller
due to the difference in PLOW .)
The error in case 3 is judged to be acceptable. We now need to see how we do when the PSD template is not flat.
Case 4: Further generalizing our results, we next consider the case where the template is not constant across an
interval. For example purposes, we break the interval into two equal subintervals, and let PT ( f ) be constant on
each sub-interval, with the constant value on the second sub-interval 10 dB lower than on the first. For this case, we
can conjure up the worst case situation for a PSD of the class: Transmit at the PSDLIMIT over the second subinterval, and adjust the PSD on the first to keep a constant average power over the whole interval. For the case of
PLIMIT = PT + 2.5 dB and PLOW = 0 , we can transmit at PLIMIT for 51.9% of the high PSD sub-interval and still
keep our average power over the entire interval the same as the template. For the minimal overlap 2 disturber case,
we calculate the peak error to be 1.63 dB, substantially higher than for the constant PSD case. For the case of
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adding a third interferer transmitting with the template, the error vs. normalized fOffset (normalized to ∆f ) is shown
as Case 4 in Figure M. 4. The peak error for this case is 1.22 dB and the average error is 0.99 dB. (The maximum
value of fOffset is smaller because the offset is only used for the high-PSD sub-interval.)
Average Normalized Power: Since the error from a limit on the average power transmitted in a sliding window can
be larger than desired, we looked to an alternative constraint. The problem of the average power constraint is that it
does not adequately weight the areas where the PSD template is small, compared to the places where the PSD
template is large. To more equally weight all of the PSD areas, we can use the following definition of average
normalized power:
Pavg ,norm (f , ∆f ) =
1
∆f
f + ∆f
∫
f
P (f ′)
df ′
PT (f ′)
(24)
and then limit Pavg ,norm to no more than 0 dB. This criterion weights the power in the class member PSDs relative
to the template values at all frequencies equally. When the PSD template is flat, this criterion is identical to the
average power in a frequency interval constraint (Eq. 19). So in effect, this criterion normalizes non-flat PSD
templates so that the average power constraint can be applied. For example, using this criterion, the error from
Example Case 4 is reduced to that Case 3 – an amount quite similar to the case of the currently specified DMT
transmission.
Since this criterion was consistent with the previously used average power in a sliding window requirement (from
T1.413-1998) and provides a lower equivalent crosstalk modeling error for non-flat PSD templates, it was selected
as the general conformance requirement, along with a PSD mask equal to the template + 2.5 dB. To account for
variations in equipment and measurement errors, both PSD mask and average normalized power requirements were
raised by 1 dB, resulting in the following requirements:
For a given equipment PSD P( f ) and SM class PSD template PT ( f ) ,
(
)
10 log Pavg ,norm (f ,100 kHz ) ≤ 1dB , and
10 log (P (f )) ≤ 10 log (PT (f )) + 3.5 dB for all frequencies of interest.
These requirements are reflected in the text of clause 6.1.
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Annex N: Informative references
(Informative )
[1] TIA/EIA-596, Public Switched Digital Service.
[2] B. J. Dunbar, et. al., “Dataport – Channel Units for Digital Data System 56-kbps Rate”, BSTJ, vol. 61 no. 9,
November 1982.
[3] T. Berger & D. W. Tufts, “Optimum Pulse Amplitude Modulation Part I: Transmitter – Receiver Design and
Bounds from Information Theory,” IEEE Transactions on Information Theory, vol. IT-13, no. 2, April 1967.
[4] Committee T1 Technical Report No. 28, High Bit Rate Digital Subscriber Lines (HDSL)
[5] GTE Customer Handbook - 500, Issue 1, 1972
[6] Transmission Systems for Communications, Bell Telephone Laboratories, Fifth Edition, 1982.
[7] ASTM D 4566, Standard Test Methods for Electrical Performance of Insulations and Jackets for
Telecommunications Wire and Cable, 1994.
[8] T1.403-1999, American National Standard for Telecommunications – Network and Customer Installation
Interfaces - DS1 Electrical Interface.
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