ACTIVE INTEGRATED ANTENNA
WITH SIMULTANEOUS TRANSMIT AND RECEIVE
QI XUEFENG
A project report submitted in partial fulfillment of the
requirements for the award of the degree of
Master of Engineering
(Electrical-Electronics & Telecommunication)
Faculty of Electrical Engineering
Universiti Teknologi Malaysia
MAY 2007
iii
“To my beloved Father, Mothe r, Sister and wife ”
iv
ACKNOWLEDGEMENTS
Firstly, special thanks to go to Dr MOHAMAD KAMAL A RAHIM for giving
this opportunity to work under his supervision and for sharing his great knowledge
and experience with me.
Secondly, I would like to convey my deepest gratitude to Mr. Thelaha Masri
for his guidance to complete this research. Appreciation is also extend to all people
who gave the author heartfelt corporation and shared their knowledge and for giving
some their valuable time.
Finally, my biggest gratitude is to my family, for their endless love,
emotional support and belief in me .Without them I would never come up to this
stage.
v
ABSTRACT
An active antenna with simultaneous transmit and receive function, integrate an
active devices onto a printed antenna to improve its performance or combine
functions within the antenna itself. Such antenna are of increasing interest, as system
designers require more complex functions to be implemented in reduced space. This
paper discusses the integration of active antennas by combining both transmit and
receive functions into one single antenna. Four main components in the design are
circular polarized microstrip patch antenna, rat race coupler, power divider and
amplifiers. All the simulations are done using the Agilent ADS. The circular
polarized antenna resonates at 2. 4 GHz. Two MESFET amplifiers have been used to
transmit and receive the channel separately. The rat race coupler isolates the two
channels and a Tee junction power divider is connected the two channels to the input
and output port. The channels are of the same frequency. The simulation and
measurement results of the microstrip patch antenna for S11 are lower than -10 dB at
frequency of 2.4 GHz. The antenna polarization is confirmed as a circular polarized,
as can be seen in the radiation pattern from the measured and simulated results. The
amplifier biasing circuit is supplied by two power sources; one is the drain voltage
(Vds) which is positive and the other is the gate voltage (Vgs), which is negative.
After integrating all of the components, the radiation pattern is measured for both
transmit and receive. The beamwidth of the antenna is in the range of 60o – 70o for H
plane. The radiation pattern for E plane is smaller compared with the H plane. The
comparison between the passive and the active integrated antenna shows that the
active integrated antenna has 3 dB extra gain compared to the passive antenna for
both transmit and receive. The isolation between transmit and receive is between
20 – 25 dB.
vi
ABSTRAK
Sebuah antenna aktif yang memiliki fungsi serentak ‘menghantar’ dan
‘menerima’, disepadukan dengan komponen-komponen aktif untuk meningkatkan
prestasi dan fungsi dalam antena itu sendiri. Minat terhadap antena yang memiliki
ciri-ciri ini bertambah, dimana para pereka system memerlukan fungsi-fungsi
kompleks bersepadu dalam ruang yang kecil. Thesis ini membincangkan tentang
integrasi antenna aktif, dimana kedua-dua fungsi ‘menghantar’ dan ‘menerima’
digabungkan dalam satu antenna. Empat komponen utama dalam rekabentuk ini
termasuklah sebuah antena microstrip tampal yang berpolarisasi keliling, Gandingan
Rat Race, pembahagi kuasa dan amplifier. Kesemua simulasi dibuat menggunakan
software Agilent ADS. Antenna yang berpolarisasi keliling bergema pada frekuensi
2.4 GHz. Dua unit amplifier ‘MESFET’ telah digunakan untuk menghantar dan
menerima saluran (channel) secara berasingan. Gandingan ‘rat race’ pula
memisahkan kedua-dua saluran ini manakala sebuah pembahagi kuasa berbentuk
‘Tee’ disambung kepada kedua-dua saluran pada port masukan dan port keluaran.
Kedua-dua saluran beroperasi pada julat frekuensi yang sama. Data daripada
keputusan simulasi dan pengukuran untuk antenna mikrostrip tampal ini
menunjukkan julat S11 yang rendah dibawah -10dB untuk frekuensi 2.4GHz.
Polarisasi antenna adalah disahkan berpolarisasi keliling sepertimana dapat di lihat
dalam keputusan penyinaran daripada simulasi dan pengukuran. Pincangan litar
penguat adalah dibekalkan oleh dua kuasa sumber; satu daripadanya ialah voltan
parit (Vds) yang merupakan punca dan satu lagi ialah voltan get (Vgs), yang
merupakan punca negatif. Selepas semua komponen disepadukan, bentuk pancaran
diukur untuk penghantaran dan penerimaan. Lebar alur antena berada dalam julat
60o – 70o untuk satah H. Bentuk pancaran untuk satah E adalah lebih kecil
dibandingkan dengan satah H. Perbandingan di antara antena pasif dan antena aktif
bersepadu, menunjukkna bahawa antena aktif bersepadu mempunyai gandaan 3dB
lebih berbanding dengan antena pasif untuk kedua-dua penghantaran dan
penerimaan. Pengasingan di antara penghantaran dan penerimaan adalah di antara 20
– 25 dB.
vii
TABEL OF CONTENTS
CHAPTER
TITLE
PAGE
TITLE
i
DECLARATION
ii
DEDICATION
iii
ACKNOWLEDGMENT
iv
ABSTRACT
v
ABSTRAKT
vi
TABLE OF CONTENTS
vii
LIST OF APPENDICES
xi
1
INTRODUCTION
1.1
Introduction
1
1.2
Problem Statement
3
1.3
Objective
3
1.4
Scope of Research
4
1.5
Research Methodology
4
1.6
Specification
7
1.7
Thesis Outline
7
1.8
Summary
9
viii
2
LITERATURE REVIEW
2.1
Introduction
10
2.2
Bingchi Luo’s Active Integerated Antenna
10
2.3
Robert
Flynt’s
Active
Integrated
Antenna
14
Transceiver
2.4
M.J. Cryan and P.S. Hall’s AIA with
simultaneous Tx&Rx operation
2.5
17
V.B. Erturk’s Design/Analysis of an Active
Integrated Antenna
2.6
23
S.L. Karode’s Dual Polarized Microstrip Patch
Antenna Using Feedforward Isolation
Enhancement for Simultaneous Transmit receive
2.7
3
Applications
25
Summary
29
SYSTEM DESIGN AND FABRICATION
3.1
Introduction
31
3.2
Microstrip Patch Antenna
32
3.2.1
Antenna Polarization
33
3.2.2
Radiating Microstrip Patch
35
3.2.3
Rectangular
3.3
3.4
Microstrip
Antenna
Design Way
37
3.2.4
Circular Polarized Patch Antenna
39
3.2.5
Microstrip Patch Antenna Desig
41
3.2.6
Liner Polarized Square Patch
41
3.2.7
Circular Polarized Patch Antenna
44
Rat-race Coupler
47
3.3.1
48
Rat-race Coupler Design
Amplifier
3.4.1
50
Amplifier Classes ( Power
amplifier for Tx )
51
3.4.2
Transistor Amplifiers
53
3.4.3
FET Transistor
53
ix
3.4.4
3.5
3.6
3.7
Low Noise Amplifier ( Rx )
Amplifier’s Biasing Circuit
58
3.5.1
Quarter-wavelength
59
3.5.2
Quarter-wave stubs
60
3.5.3
Quarter-wave transformers
61
3.5.4
Microstrip radial stub
62
3.5.6
Amplifier’s Biasing Circuit Design
63
T-junction Power Divider
64
3.6.1
66
T-junction Power Divider Design
Combination Design
3.71
3.7.2
68
Combination of Whole Circuit
except Antenna
Summary
4
68
Combination of Coupler and
Amplifier
3.8
58
69
72
MEASUREMENT SETUP
4.1
Introduction
73
4.2
Biasing Circuit Testing Setup
73
4.3
Radiation Pattern and Gain Measurement Setup
74
4.4
Power Strength Measurement Setup
77
4.5
Summary
79
5
RESULTS COMPARISON AND ANALYSIS
5.1
Introduction
80
5.2
Passive Antenna Analysis Conclusion
80
5.3
Rat-race Coupler Analysis
83
5.4
T-junction Power Divider Analysis
84
5.5
Amplifier’s Biasing Circuit Analysis
86
5.6
Whole Antenna Analysis
86
5.6.1
Antenna Radiation Pattern
Measurement and Analysis
5.6.2
Antenna Comparison with
86
x
Monopole Antenna and Analysis
5.7
6
Summary
89
92
CONCLUSION AND FUTURE WORK
6.1
Conclusion
93
6.2
Future Work
94
References
95
Appendix A
97
Appendix B
98
Appendix C
99
Appendix D
100
Appendix E
101
xi
LIST OF APPENDICES
APPENDIX
TITLE
PAGE
A
Circular Polarized Single Antenna Prototype
97
B
Rat-race Coupler Prototype
98
C
99
D
T-junction Power Divider Prototype
Amplifier Biasing Circuit Prototype
100
E
AIA with Tx & Rx Prototype
101
CHAPTER 1
INTRODUCTION
1.1 Introduction
The active integrated antenna (AIA) has been a growing area of research in
recent years, as the microwave integrated circuit and monolithic microwave
integrated circuit technologies became more mature allowing for high-level
integration. From a microwave engineer’s viewpoint, an AIA can be regarded as an
active microwave circuit in which the output or input port is free space instead of a
conventional 50-Ω interface. In this case, the antenna can provide certain circuit
functions such as resonating, filtering, and duplexing, in addition to its original role
as a radiating element. On the other hand, from an antenna designer’s point-of-view,
the AIA is an antenna that possesses built-in signal and wave-processing capabilities
such as mixing and amplification. A typical AIA consists of active devices such as
Gunn diodes or three-terminal devices to form an active circuit, and planar antennas
such as dipoles, microstrip patches, bowties, or slot antennas.
Looking back in history, the idea of using active antennas can be traced back to
as early as 1928 [1]. At that time, a small antenna with an electron tube was
commonly used in radio broadcast receivers around 1 MHz. After the invention of
2
high-frequency transistors, the study of active antennas received much more attention
and several pioneering works were reported in the 1960s and 1970s. Several
advantages of implementing the active devices in passive radiating elements were
discussed in [2]. For instance, these works include increasing the effective length of
short antenna and increasing antenna bandwidth, decreasing the mutual coupling
between array elements, and improving the noise factor. Over the past decades, the
major driving forces for the research on AIA are the development of novel efficient
quasi-optical power combiners. The original purpose for the quasi-optical power
combining is to combine the output power from an array of many solid-state devices
in free space to overcome combiner loss limitations, which are significant at
millimeter-wave frequencies. Rather, this paper reviews more on the functional
performance of individual AIA Recently, numerous innovative designs based on the
AIA’s concept have been proposed and successively demonstrated. AIA technology
has evolved to a point where practical implementation for use in the latest microwave
and millimeter-wave system is considered feasible. It is currently pursued in a
number of related fields such as power combining, beam steering and switching,
retro directive arrays, as well as high-efficiency power-amplifier designs. These
AIA-based designs are particularly attractive for millimeter-wave systems because
they provide an effective solution to several fundamental problems at these
frequencies, including higher transmission-line loss, limited source power, reduced
antenna efficiency, and lack of high-performance phase shifter. This paper reviews
the recent research activities related to this emerging technology with emphasis on its
applications in integrated antenna, amplifier.
An active antenna with simultaneous transmit and receive integrates an active
device into a printed antenna to improve its performance or combine functions within
the antenna itself. Such antennas are of increasing interest [3] as system designers
require more complex functions to be implemented in reduced space. New highvolume millimeter applications such as vehicle collision avoidance radar, wireless
local-area networks (WLAN), and electronic tagging are driving costs lower and
putting further constraints on size and weight. This paper hopes to take further steps
in the integration of active antennas by combining both transmit and receive
functions into a single antenna. This paper uses a circular polarized microstrip-patch
antenna resonant at 2.4 GHz, with two metal–semiconductor field-effect transistor
3
(MESFET) to form two amplifiers used in transmit and receive channel separately,
one Rat-Race Coupler isolated the two channels and one T-junction Power Divider
connected the two channels to the input and output port. The channels are of the
same frequency.
1.2 Problem Statement
The problem statement of this project is stated in the follow: An antenna with
good gain and integration of transmit and receive for the indoor WLAN in 2.4 GHz
band is required. Theoretical results are obtained to satisfy good return loss and gain
requirements and isolation between transmit and receive channels for the active
antenna with simultaneous transmit and receive, but the practical result is still big
challenge to be verified, this project will prove that down-to-earth.
1.3 Objective
The objective of this project is to design, simulate, fabricate and test a
simultaneous transmit–receive active antenna at the frequency of 2.4 GHz, which is
attractive for potential uses in WLAN communication.
1.4 Scope of Research
1. To design a circular polarized microstrip patch antenna at 2.4 GHz;
To design a Rat-race Coupler at 2.4 GHz;
To design an Amplifier’s Biasing circuit at 2.4 GHz;
To design a T-junction Power Divider at 2.4 GHz;
To combine all these parts above and design an Active Integrated Antenna with
Simultaneous Transmit and Receive at 2.4 GHz.
4
2
The antenna specifications include parameters such as frequency,
bandwidth, and all theoretical investigations.
3
To simulate all these designs until reach the best result using Microwave
Office and Agilent ADS software.
4
The prototype will be fabricated with the available microstrip materials
(FR4) based on the simulation and using wet etching technique.
5 All the fabrication will be tested in Lab and test bed as a field trial to measure
their performance.
6
The comparison between measurement results and simulation results.
7 Analyzing the results and writing up thesis.
1.5 Research Methodology
A theoretical and experimental design approach was utilized to optimize the
antenna structure, the strategy implemented for simplifying the design and
development procedures in this research work can be divided into the following
points:
1. Initial concept
•
Literature review
•
Problem statement
•
Design conceptual understanding
2. Design and simulation stage
•
Design consideration based on previous research results
•
Decide the input parameters of the antenna
•
Design the passive part of the antenna using antenna design software
(Microwave Office and Agilent ADS).
•
Do simulation for the whole antenna design using antenna design
software (Microwave Office and Agilent ADS).
3. Prototype stage
•
Fabrication of the passive part of the designed antenna
5
•
Combining the passive and active part of the proposed antenna
4. Measurement stage
•
Do measurement of the properties of the fabricated antenna
5. Analysis and conclusion stage
•
Do comparison between measurement results and the simulation results
and draw a conclusion
6. Collect results and produce papers
The antenna fabrication needs to fit within the costing constraints and the
availability of materials. The design and development procedures are briefly
summarized in the following chart Figure 1.1 in particular, this methodology
provides an approximate chronological progress of the work performed to finally
complete the full design cycle.
6
1、Initial antenna design specifications:
·Frequency of the operation
·Antenna Bandwidth
·Desired Polarization
2、Design antenna dimensions and choose
feed type
No
3、Simulation using an electromagnetic analysis tool
for
·Return Loss/VSWR
·Polarization , Isolation
4、Design specification
9、Change
parameters and
redesign
the antenna
Yes
5、Prototype fabrication:
Fabricate the passive antenna and combine the passive and
active part on the same designed antenna substrate.
6、Perform experimental evaluation of the constructed
prototype (return loss, bandwidth , aperture profile)
7、Result agree with
simulation?
No
yes
8、Design complete
Figure 1.1 Flow chart representing of AIA with Tx & Rx function
7
1.6 Specification
z Antenna patch : FR4 materials εr= 4.6 ,
h (substrate thickness) = 1.6 mm
T (conductor thickness) = 0.035 mm
Patch size: W=205mm, L=160mm
z Antenna resonate frequency at 2.4 GHz
z Use Transistor : ATF-21186 (2)
z Input impedance is 50 ohm
z Antenna has Tx and Rx function
z Passive antenna Bandwidth more than 100MHz , 4% efficiency more
z Tx and Rx channel isolation is lower than -30dB
1.7 Thesis Outline
This thesis consists of six chapters describing all the work done in the project.
The thesis outline is generally described as follows.
Chapter 1: This chapter explains the introduction of the project. Brief general
background is presented. The objectives of the project are clearly phased with
detailed. The research scope implementation plan and methodology are also
presented.
Chapter 2: This chapter discusses some previous literature. It includes Bingchi Luo’s
new approach which omitted the input matching on designing an Active Integrated
Receiving Antenna, Robert Flynt’s low cost and compact Active Integrated Antenna
Transceiver, M.J.Cryan and P.S. Hall’s Integrated Active Antenna with simultaneous
transmit-receive operation, S. L. Karode’s Dual Polarized Microstrip Patch Antenna
Using Feedforward
Isolation Enhancement for Simultanious Transmit receive
8
Applications and V.B. Erturk’s Design/Analysis of an Active Integrated Antenna.
Chapter 3: This chapter discusses the theory and equation needed to design this AIA
and its components. This chapter also presents the design, simulation and fabrication
of every components and final AIA in detail and presents the simulation results.
Chapter 4: This chapter presents the setting up of different measurement separately,
it includes the setting up for biasing circuit testing, setting up for radiation pattern
and gain measurement and setting up for power strength measurement.
Chapter 5: This chapter discusses and analyzes the results of antenna prototype
measurement compared to the simulation result. The antenna application in the real
environment and comparison with monopole also presents in this chapter.
Chapter 6: This chapter presents the conclusion based on the analysis and
comparison of results in chapter 5. Recommendations for future works are also
presented.
9
1.8 Summary
Brief introduction on project and its objective have been presented. Its scopes
has also been presented to give a clear view on the direction of this project. The
methodology and outline of this thesis have also been described.
CHAPTER 2
LITERATURE REVIEW
2.1 Introduction
There are some researches have been gone into this field, and the most of them
developed the application. For example, Bingchi Luo explained the new approach which
omitted the input matching on designing an Active Integrated Receiving Antenna;
Robert Flynt developed a low cost and compact Active Integrated Antenna Transceiver;
M.J. Cryan and P.S. Hall developed an Integrated Active Antenna with simultaneous
transmit-receive operation; V.B. Erturk’s Design/Analysis of an Active Integrated
Antenna and S.
L. Karode’s Dual Polarized Microstrip Patch Antenna Using
Feedforward Isolation Enhancement for Simultaneous Transmit receive Applications
2.2 Bingchi Luo’s Active Integerated Antenna [ 3 ]
This design provides the new approach on designing a Receiving Antenna. It
concentrates on the research of the new approach by omitting the input matching of an
Active Integrated Receiving Antenna at the frequency of 2GHz.
11
Figure 2.1 normal conjugate matching
In order to improve the NF while maintaining a good gain, conjugate matching is
no longer used at the input port of the transistor. Instead, transistor is tested to see which
impedance gives a better NF and a high enough gain. Then the Antenna is designed to
have such output impedance. And it is connected directly to the transistor. As compared
to the normal approach, Matching Network A is omitted. Thus less circuit element is
required. Matching Network B still presents so that the output of the Receiving Antenna
is still the conventional 50-ohm interface.
Figure 2.2 New approach using only output conjugate matching
Procedures:
1. Bias Point
The bias point is simulated in ADS. The bias point is then chosen to be V
CE
= 2V,
12
V
BE
= 888.7mV.
2. Check for stability
Stability of the amplifier is an important factor in the design. Notice in the circuit on
the left, that a microstrip line is added between the emitter and the ground. It is to
improve the stability. This is analogous adding a resistor to the emitter. Stability must be
check at each step of the design.
Figure 2.3 Stability checking
3. Output Matching
Figure 2.4 The output matching
Notice two microstripe lines are added to the collector port of the BJT. This is
called single stub matching. Now looking from the right hand side of the Matching
Network to the left, the output impedance should be 50-ohm. Meanwhile, looking
from the left hand side of the Matching Network to the right, the load impedance
should be exactly the conjugate of the output impedance of the transistor.
4. Find Z opt
Figure 2.5 The optimization matching
14
Figure 2.6 Simulation results of impedance
The above circuit is now used to find the optimum impedance for NF. The Term
1 impedance is set to values ranged from 45 to 80 ohms. The width of the microstrip
line is adjusted each time such that its characteristic impedance is the same as the
Term 1 impedance. The value of S11, S22, S21, and NF is recorded each time. From the
graph on the left, it is noticed that NF keeps decreasing as impedance increasing, so
large Zopt is preferred for better NF; however, from the graph on the right, S22 reaches
its minimum value at 65 ohm. So Zopt is chosen to be 65 ohm as a compromise.
2.3 Robert Flynt’s Active Integrated Antenna Transceiver [ 4 ]
This design describes the novel integration of an FET and a mixer diode on the
surface of an inverted patch antenna. The FET is configured within the inverted
patch structure to oscillate at C-band . The oscillator acts as a transmitter and it also
couples a portion of the power to the mixer as a local oscillator. The optimal position
for placing the mixer diode on the patch is determined. A Schottky barrier diode is
placed within the antenna cavity and it receives a portion of the oscillator power that
mixes with an incoming RF signal. For a 5.8 GHz LO and a 6 GHz incoming RF
signal, the 200 MHz IF exhibits 5.5 dB isotropic conversion loss ( Liso ). Very good
antenna patterns were obtained. The mixer and the oscillator mounted directly onto
the patch antenna, reducing the need for interconnect lines. This structure can be
used for many applications such as wireless communications, Sensors and radar
systems.
15
Figure 2.7 Complete active integrated antenna transceiver (a) top view (b) side view.
Figure 2.8 Rectified DC voltage measured parallel to the bias cut
Figure 2.9 Available LO power and measured IF power vs position.
16
Figure 2.10 H-Plane pattern with the mixer diode in place. The cross polarization
level = -18.84 dB and the HPBW = 67.0°.
Figure 2.11 E-Plane pattern with the mixer diode in place. The cross polarization
level = -17.99 dB and the HPBW = 49.30
Figure 2.7 shows the FET and mixer integrated with the inverted stripline
antenna. For the integrated antenna, the introduction of several DC blocks for biasing
dramatically changes the performance with respect to the original circular patch.
17
Gaps are etched to isolate the source, gate and drain for DC biasing. Chip capacitors
were used to provide some RF continuity between the two halves of the patch. DC
bias is achieved with voltage across the drain-to-gate(VDC) and a 2 Ω chip resistor
from the source to the gate. A similar integrated antenna with only an FET device [8]
has previously demonstrated good oscillation and excellent radiation performance.
Due to the physical considerations for placing the mixer diode onto the patch, the
cavity depth had to be changed This resulted in a gain of 6 dB which is lower than
the value reported in [8].
The best possible position for the mixer diode was determined by trying to
obtain a good impedance match between the diode and the patch at a location where
there is sufficient LO power. When the diode is placed at a location when there is
sufficient LO power, the diode will be positively biased due to the rectified DC
voltage from the LO source. The relative received power as a function of position
can be approximated by measuring the rectified DC voltage that results from power
applied from an external source while the FET is turned off. This is plotted as a
function of position in Figure 2.8. It was determined that the FET produced
sufficient LO power for any practical position for the diode. This is shown in Figure
2.9 along with the measured IF power plotted as a function of position. The final
position was a compromise between the impedance matching, relative received
power, and the physical constraints of mounting the diode. The position was also
chosen so that the active antenna radiating pattern was not adversely affected. The
optimized position .was determined to be 8 mm from the center of the patch on the
gate side of the bias cut, as shown in Figure 2.7. A low pass filter was designed and
placed behind the cavity to filter out the LO and RF signals.
2.4 M.J. Cryan and P.S. Hall’s AIA with Simultaneous Tx & Rx Operation [ 5 ]
The authors present results for a novel two element active transmit-receive array
using dual linear polarization and sequential rotation. Each element includes an
integrated oscillator and amplifier mounted on orthogonal edges of a square patch,
such that transmit and receive paths are isolated and polarization duplexed. The array
18
gives in excess of 55dB transmit-receive isolation at 3.77 GHz.
This design uses a square microstrip-patch antenna resonant at 4.0 GHz, with a
metal–semiconductor field-effect transistor(MESFET) centrally mounted on the edge
of the patch to form an oscillator and another MESFET, configured as an amplifier
on the orthogonal edge to act as the first stage in a receiver. The inherent isolation of
the center points of orthogonal edges of a square patch is used as the basis for the
transmit–receive isolation. This transceiver is linearly polarized with transmit and
receive channels on orthogonal polarizations. The channels are of the same
frequency, but could be offset depending on the application. A method to improve
the isolation of the single patch is that of sequential rotation. Here, the receiver
outputs are taken from opposite edges of the two patches and the phases of the direct
feed through signals from transmit to receive are adjusted so that they are 1800 out of
phase—which upon combining, will cancel. The received signals are forced to be
1800 out of phase by the positions of the receiver outputs; thus, when combined they
will add in-phase. This method can increase the isolation by 20–30 dB .The oscillator
and the final antenna are shown below.
Figure2.12 MDS circuit schematic for HB simulation of a microstrip-patch oscillator
19
Figure 2.13 Physical layout for a 4-GHz patch oscillator. Patch length
( Lp ) = 24 mm, width ( Wp ) = 20 mm, εr = 2.33, substrate height = 0.508 mm.
Figure 2.14 Sequentially rotated two-element active array. Patch length = 24 mm,
width = 24 mm, εr = 2.33, substrate height = 0.508 mm.
20
(a)
(b)
Figure 2.15 Measured and modeled performance of a 4-GHz patch oscillator.
(a) Frequency and (b) output power at fundamental and first harmonic.
21
(a)
Figure 2.16 Results for array shown in Figure 2.14 (a) Transmit. (b) Receive.
Figure2.12 shows the schematic layout for the patch oscillator model; the patch
is represented by five transmission lines, which allows for the connection of bias
22
circuits and active device. The transistor is connected close to the 50- point on the
nonradiating edge of the patch, and short-circuited transmission lines are connected
to the source and gate terminals of the FET. Drain bias is applied to the patch via a
radial-stub bias circuit. The oscillator free-running frequency can be tuned by
altering the drain-bias voltage; this allows the possibility of employing frequency or
phase modulation techniques. Figure2.13 shows a schematic generated from
Figure2.12 for the physical layout of the patch oscillator circuit. Measured and
modeled results for frequency and output power have been compared, and are shown
in Figure2.15. The modeled output power is defined as the sum of the powers in
the radiation resistances, and the measured output power has been calculated from
the effective isotropic radiated power (EIRP) and the gain of an identical passive
array. Figure2.15 (a) shows the free-running frequency-tuning response—the trend
between measured and modeled results shows reasonable agreement. However,
absolute agreement is less good. This will be very dependent on the agreement
between the large signal model and the actual device used, and due to device
the output power, and good agreement is shown for the fundamental and reasonable
agreement for the first harmonic. These results suggest that a voltage-controlled
patch oscillator can be designed and implemented to a reasonable level of accuracy
by using currently available simulation packages. A tuning range of 20 MHz is
achieved, which would be sufficient for many system applications.
A schematic of the array is shown in Figure2.14, The measured results are
shown in Figure2.16. Figure 2.16(a) shows the oscillator output power together
with the isolation signal after cancellation. The output power of the oscillator is
calculated from the EIRP and measurements of the gain of an identical passive array.
Frequency tuning is performed by adjusting both drain and gate voltages of both
oscillators. The tuning bandwidth was found to be 28 MHz centered on 4.04 GHz.
An array output power of 5.4 dBm was obtained with an isolation of better than 45
dB at 4.05 GHz. The isolation is better than 43 dB across the whole band, and this
could be improved by using broad-band constant phase-shift networks, such as
Schiffman phase shifters. Figure 2.16(b) shows the receive performance. A gain of
8.2 dB was obtained at 4.05 GHz—the maximum being 16.5 dBi. The output return
loss is better than 10 dB from 4.04 to 4.20 GHz. The transmit and receive bands are
reasonably well aligned, and with minor modifications to the matching networks, an
23
optimum design could be achieved. Initially, it was felt that greater isolation could be
obtained since the passive isolation with wire-bond connections was 40 dB, thus with
sequential rotation this might be expected to be 70 dB. However, since the amplifier
is matched to the patch edge impedance, and the oscillator as a result of the
oscillation conditions is delivering all its output power to the patch, then a more
realistic estimate for the passive isolation was that of the quarter-wave matched
patch. This agrees with the results of Figure 2.16, since with a passive isolation of
25 dB and an extra 30 dB from sequential rotation, the overall isolation would be
approximately 55 dB—similar to that shown in Figure 2.16. These results suggest
that by mismatching the amplifier, extra passive isolation could be achieved at the
expense of receiver gain.
2.5 V.B. Erturk’s Design/Analysis of an Active Integrated Antenna [ 6 ]
In this design, an active integrated antenna design and analysis approach
based on a hybrid combination of full wave/nonlinear circuit solvers has been
introduced. A prototype antenna working at 2.01 GHz is designed and fabricated
using a low cost medium power Silicon Bipolar Transistor. Radiation patterns are
calculated with two different FDTD models of the antenna and compared with
measurements. Numerical and experimental results for the radiation patterns are
found to be in good agreement and the cross-polarizations in the main beam region
are 30dB and 25dB down in the
H- and E-planes, respectively.
Figure 2.18, Figure 2.19 show both the measured and calculated E- and Hplane radiation patterns, respectively. The calculated results are for the case where
the substrate is assumed to be infinite in size. Therefore, these results do not include
the effect of the edges. Additional computed results will be presented where the
effect of the edges of the ground plane are taken into account and compared with the
measured data. In conclusion, the design shows good oscillator characteristics and
good agreement between the numerical and experimental results.
24
Figure 2.17 Configuration of active integrated microstrip antenna
Figure 2.18 Experimental and numerical results for the E-plane radiation pattern of
the active integrated antenna. solid line : measurement result; dashed line : numerical
result for antenna with infinite ground plane.
25
Figure 2.19 Experimental and numerical results for the H-plane radiation pattern
of the active integrated antenna. solid line: experimental result, dashed line:
numerical result for antenna with infinite ground plane.
2.6 S.
L. Karode’s Dual Polarized Microstrip Patch Antenna Using
Feedforward
Isolation Enhancement for Simultaneous Transmit receive
Applications [ 7 ]
This paper presents a method for increasing the isolation between transmit
and receive ports of a dual polarized microstrip patch antenna in order to enhance its
performance in single frequency simultaneous transmit receive applications. A
feedforward circuit signal cancellation scheme is demonstrated which is shown to
improve the dual polarized antenna port isolation performance to be better than 60dB while leaving the antenna radiation pattern largely unaltered when compared to
a uncompensated dual port microstrip patch antenna with -35dB port isolation.
26
In the dual polarized antenna developed in this work the TMlo and TMol
modes are used simultaneously. Here, for the TMol mode, the length of the patch acts
as the
radiating edge whereas in the TMlo mode the width is the radiating edge resulting in
orthogonal polarization allocation for the two ports. It has been reported that the
maximum isolation between the ports in a dual polarized patch can be obtained by
selecting the position of these ports to lie at the centre of the orthogonal edges of the
patch. For proper matching the feed point can be moved to the inside of the patch
along the cross hairs of the orthogonal edges of the patch OX, OY to an appropriate
location Xi, Yi from the patch center which offers the desired real impedance part
and, to facilitate matching, preferably with minimum reactive part. This is shown in
Figure 2.20. The patch impedance is maximum at its edge and falls rapidly as it is
moved inside towards the center of the patch, position ‘0’ in Figure 2.20. Figure
2.21 shows impedance profile for the patch from edge to the center of the patch
antenna, W=L=70mm constructed on FR4 material. Any residual reactive portion
can then be nullified using matching components. Such an arrangement will offer
the best possible inherent isolation available from a single patch antenna. The
isolation of the patch can be further improved by using a feedforward isolation
circuit. The feedforward isolation circuit works on the principle of cancellation of the
leakage signal by summing it with an equal amplitude sampled signal from the input,
but with a destructive phase shift introduced. These two signals, when added together
at antenna port, interfere destructively thus improving the isolation considerably.
The feedforward high isolation circuit was then added to the patch as shown
in Figure 2.21. A signal sampled from the input using a coupler is attenuated and
phase shifted such that the amplitude of the sampled signal is same as that of leakage
signal at the receive port but it is 1800 out of phase with respect to the leaked signal.
This is
then added to the signal coming from the receive port of the patch so that
cancellation occurs. The addition of the sampled signal with the signal at the
received port does not affect the received signal from the antenna since only the
leakage signal is cancelled.
Figure 2.22 (a) and (b) shows the s-parameters and the port isolation for the
27
dual polarized patch with high isolation circuit in place. Here, the port isolation for
this
arrangement is now >60dB at the lGHz design frequency, a 30dB improvement on
Figure 2.20 Dual polarized microstrip patch antenna
Figure 2.21 Dual polarized patch antenna with feedforward isolation circuit
28
(a) Input matching
(b) Port isolation
Figure 2.22 S-parameters for the dual polarized microstrip patch antenna with
feedforward isolation circuit [ 7 ]
29
Figure 2.23 Patch radiation patterns with feedforward network in position [ 7 ]
the inherent port isolation of the antenna element. Over the entire measured
bandwidth 800-1 200MHz the port isolation is better than 28dB. Under these
condition the input reflection coefficient at both ports is at least -26dB. The
bandwidth of dual port microstrip patch antenna element does not degrade after the
addition of the feedforward isolation circuit. For a VSWR of 1.2 the bandwidth
before and after inclusion of the feedforward networks are 4.1 and 4.2 MHz
respectively. Far field E and H field co-polar and cross-polar radiation patterns for
the transmit and receive ports are shown in Figure 2.22 a and b respectively,
obstinately these are same as the patch element without the feedforward network in
position.
2.7 Summury
In this chapter, some literature reviews have been presented which presented
some techniques of active integrated antenna. Bingchi Luo’s design explained the
new approach which omitted the input matching on designing an Active Integrated
Receiving Antenna; Robert Flynt developed a low cost and compact Active
Integrated Antenna Transceiver; M.J. Cryan and P.S. Hall developed an Integrated
Active Antenna with simultaneous transmit-receive operation; V.B. Erturk presented
30
an active integrated antenna design and analysis approach based on a hybrid
combination of full wave/nonlinear circuit solvers; L. Karode developed a Dual
Polarized Microstrip Patch Antenna Using Feedforward Isolation Enhancement for
Simultaneous Transmit receive Applications. All these designs have been presented
and results have been analyzed.
CHAPTER 3
SYSTEM DESIGN AND FABRICATION
3.1 Introduction
This AIA with simultaneous Tx & Rx is normally made up of several parts: a
Microstrip Circular Polarized Patch Antenna, a Rat-race Coupler, two Transistors (as
Amplifiers), Amplifier’s Biasing Circuit and T-junction Power Divider. A Rat-race
Coupler is commonly used to connect the radiating element to the transmitter and
receiver and isolate transmit and receive channel. Transistors are used in transmit and
receive channel separately as Amplifiers. Amplifier’s Biasing circuit is used to bias the
Amplifier. T-junction Power Divider is used to connect transmit and receive channel to
the Input/Output port.
The design is made using the Agilent ADS tool. The design process is mainly
carried out in two steps.
* The first step is a schematic representation of circuits and components. This
implies that component values and behavior are close to ideal. This will be
32
referred to as the Schematic representation or simply Schematic.
* When the initial reference design in Schematic is satisfactory the second step of
the design process can begin. By converting the Schematic design into a Layout
design, physical simulations on the actual transmission lines and traces can be
made using the Momentum tool in ADS. These simulations will give a result that
is closer to the "truth" than the Schematic simulations since the electromagnetic
simulation take adjacent components into account.
This whole system design block diagram is shown below.
Antenna
LNA
Coupler
I/O
Amplifier
Divider
Figure 3.1 Block diagram of AIA with Simultaneous Transmit & Receive
3.2 Microstrip Patch Antenna
Microstrip antennas are planar resonant cavities that leak from their edges and
radiate. Printed circuit techniques can be utilized to etch the antennas on soft substrates
to produce low-cost and repeatable antennas in a low profile. The antennas fabricated on
compliant substrates withstand tremendous shock and vibration environments.
Manufacturers for mobile communication base stations often fabricate these antennas
33
directly in sheet metal and mount them on dielectric posts or foam in a variety of ways
to eliminate the cost of substrates and etching. This also eliminates the problem of
radiation from surface waves excited in a thick dielectric substrate used to increase
bandwidth.
Microstrip antennas have attractive features such as light weight, small volume low
profile and low production cost which widely have been researched and developed in the
recent twenty years. However, microstrip antenna has a limitation which is the narrow
bandwidth of the basic element. The bandwidth of the basic patch antenna is usually 1–
3%. The bandwidth of the antenna is defined as the range of the frequencies, over which
the performance of the antenna with respect to some characteristic conforms to a specific
standard. The bandwidth of the antenna depends on the patch shape, dielectric constant,
the thickness of the substrate and the resonant frequency.
The theory of a single microstrip patch is presented in this section. The equations
shown are only approximations on how to find the input impedance, bandwidth and
resonant length of the patch.
3.2.1 Antenna Polarization
Antenna polarization is a very important consideration when choosing and installing
an antenna. Most communications systems use either vertical, horizontal or circular
polarization. Knowing the difference between polarizations and how to maximize their
benefit is very important to the antenna user.
An antenna is a transducer that converts radio frequency electric current to
electromagnetic waves that are then radiated into space. The electric field or "E" plane
determines the polarization or orientation of the radio wave. In general, most antennas
radiate either linear or circular polarization.
34
A linear polarized antenna radiates wholly in one plane containing the direction of
propagation. In a circular polarized antenna, the plane of polarization rotates in a circle
making one complete revolution during one period of the wave. If the rotation is
clockwise looking in the direction of propagation, the sense is called right hand circular
(RHC). If the rotation is counterclockwise, the sense is called left hand circular (LHC).
An antenna is said to be vertically polarized (linear) when its electric field is
perpendicular to the Earth's surface. An example of a vertical antenna is a broadcast
tower for AM radio or the "whip" antenna on an automobile.
Horizontally polarized (linear) antennas have their electric field parallel to the
Earth's surface. Television transmissions in the USA use horizontal polarization.
A circular polarized wave radiates energy in both the horizontal and vertical planes
and all planes in between. The difference, if any, between the maximum and the
minimum peaks as the antenna is rotated through all angles, is called the axial ratio or
ellipticity and is usually specified in decibels (dB). If the axial ratio is near 0 dB, the
antenna is said to be circular polarized. If the axial ratio is greater than 1-2 dB, the
polarization is often referred to as elliptical. Circular polarization is most often use on
satellite communications. This is particularly desired since the polarization of a linear
polarized radio wave may be rotated as the signal passes through any anomalies (such as
Faraday rotation) in the ionosphere. Furthermore, due to the position of the Earth with
respect to the satellite, geometric differences may vary especially if the satellite appears
to move with respect to the fixed Earth bound station. Circular polarization will keep
the signal constant regardless of these anomalies. Circularly polarized antennas are
normally more costly than linear polarized types since true circular polarization is
difficult to attain. An example of a true circularly polarized antenna is the helix.
Polarization is an important design consideration.
The polarization of each
antenna in a system should be properly aligned. Maximum signal strength between
stations occurs when both stations are using identical polarization. When choosing an
antenna, it is an important consideration as to whether the polarization is linear or
35
elliptical. If the polarization is linear, is it vertical or horizontal? If circular, is it RHC
or LHC?
On line-of-sight (LOS) paths, it is most important that the polarization of the
antennas at both ends of the path use the same polarization. In a linearly polarized
system, a misalignment of polarization of 45 degrees will degrade the signal up to 3 dB
and if misaligned 90 degrees the attenuation can be 20 dB or more. Likewise, in a
circular polarized system, both antennas must have the same sense. If not, an additional
loss of 20 dB or more will be incurred.
Also note that linearly polarized antennas will work with circularly polarized
antennas and vice versa. However, there will be up to a 3 dB loss in signal strength. In
weak signal situations, this loss of signal may impair communications.
Cross polarization is another consideration. It happens when unwanted radiation is
present from a polarization which is different from the polarization in which the antenna
was intended to radiate. For example, a vertical antenna may radiate some horizontal
polarization and vice versa. However, this is seldom a problem unless there is noise or
strong signals nearby.
3.2.2 Radiating Microstrip Patch
Microstrip antenna patch elements are the most common form of printed
antennas. They are popular for their low profile, geometry and low cost. A microstrip
device in its simplest form is a layered structure with two parallel conductors separated
by a thin dielectric substrate. The lower conductor acts as a ground plane. The device
becomes a radiating microstrip antenna when the upper conductor is a patch with a
length that is an appreciable fraction of a wavelength, approximately half a wavelength
(Figures 3.2). Conventional patch designs yield bandwidths as low as a few percent.
36
Figure 3.2a Geometry for analyzing the Edge-Fed Microstrip Patch Antenna
Figure 3.2b Side view showing the electric fields
37
Figure 3.2c Top view showing the fringing electric fields that are responsible for
radiation. The equivalent magnetic surface MS currents are also shown
3.2.3 Rectangular Microstrip Antenna Design Way
Rectangular patch antennas can be designed by using a transmission-line model [9]
suitable for moderate bandwidth antennas. Patches with bandwidths of less than 1% or
greater than 4% require a cavity analysis for accurate results, but the transmission line
model covers most designs. The lowest-order mode, TM10, resonates when the effective
length across the patch is a half-wavelength. Figure 3.2 demonstrates the patch fed
below from a coax along the resonant length. Radiation occurs from the fringing fields.
These fields extend the effective open circuit (magnetic wall) beyond the edge. The
extension is given by [11]
(3.1)
where H is the substrate thickness, W is the patch nonresonant width, and εeff is the
effective dielectric constant of a microstrip transmission line the same width as the
patch. A suitable approximation for εeff is given by [5]:
38
(3.2)
For W, assume at first that the width is λ/2.
(3.3)
By Equation. (3.2), get εeff. On substituting that value into Equation. (3.1), we obtain the
effective cutback on each edge Δ. The resonant length is
(3.4)
Figure 3.3 Coax-fed microstrip patch antenna
where εr is the substrate dielectric constant. The transmission-line model represents
the patch as a low-impedance microstrip line whose width determines the impedance
and effective dielectric constant. A combination of parallel-plate radiation conductance
and capacitive susceptance loads both radiating edges of the patch. Harrington [11] gives
the radiation conductance for a parallel-plate radiator as
39
(3.5)
where λ0 is the free-space wavelength. The capacitive susceptance relates to the effective
strip extension:
(3.6)
3.2.4 Circular Polarized Patch Antenna
Figure 3.4 show methods of achieving circular polarization with square patches fed
with two inputs. The patches are fed by equal signals 90◦ out of phase. The branch line
hybrid (Figure 3.4 a) consists of four transmission lines connected in a square. The
hybrid shown(100-Ω system) produces equal outputs 90◦ out of phase at center
frequency. The two inputs produce patterns with opposite senses of circular polarization.
Both the VSWR and axial ratio bandwidths far exceed the singly fed patch bandwidth.
Reflections due to the patch mismatch are routed to the opposite input. Patch input
reflections, undetected at the input, reduce the efficiency of the antenna by the same
amount as the singly fed patch mismatches. The antenna can be fed from below in two
places by using a coupled line hybrid, but it suffers from the same efficiency problem.
The cross-fed antenna (Figure 3.4 b) splits the signal to feed both edges. A quarter wave
length-longer line provides the extra 90◦ phase shift to give circular polarization.
Shifting the impedance from one input through a quarter-wavelength line before
adding the two in shunt cancels some of the reflection from the second line and increases
the impedance bandwidth. The impedance bandwidth approximately doubles compared
to the singly fed patch. The 6-dB axial ratio bandwidth roughly equals the singly fed
square-patch bandwidth. The polarization loss (0.5 dB) of a 6-dB axial ratio equals the 2
: 1 VSWR mismatch loss. The antennas in Figure 3.5 use asymmetries to perturb the
resonance frequencies of two possible modes and achieve circular polarization [15]. The
approximately square patches have been divided into two groups: type A, fed along the
40
centerline, and type B, fed along the diagonal. All these antennas radiate RHC.
Figure 3.4 Dual-fed circularly polarized patch antennas: (a) branch line hybrid fed;
(b)cross-fed patch
41
Figure 3.5 Classes of perturbed microstrip patches to generate circular polarization from
a single feed[11]
3.2.5 Microstrip Patch Antenna Design
The design of this antenna starts from a square patch linear polarized antenna.
After getting the antenna and matching at 2.4 GHz, then it will be changed to a circular
polarized antenna.
3.2.6 Liner Polarized Square Patch Antenna Design
Step 1: Calculation of the width ( W ): The width of the Microstrip patch antenna is
given by equation (3.3) as:
(3.3)
Substituting c = 3e8 m/s, εr = 4.6 and f = 2.4 GHz, get:
W = 0.02914m = 29.14mm
Step 2: Calculation of effective dielectric constant (εreff ): Equation (3.2) gives the
effective dielectric constant as:
42
(3.2)
Substituting εr = 4.6, W = 29.14 mm and h = 1.6mm, get:
εreff = 4.246
Step 3: Calculation of the effective cutback on each edge Δ: Equation (3.1) gives the
effective length as:
(3.1)
Substituting εreff = 4.246, W=29.12mm and H=1.6mm, get:
Δ=0.73mm
Step 4: Calculation of the length L: Equation (3.4) gives the length extension as:
(3.4)
Substituting εreff = 4.246, Δ=0.73mm, c = 3e8 m/s and f = 2.4 GHz, get:
L=28.87mm
Using the above initial number, antenna can be drawn and simulated using Agilent
ADS. During simulation, the width and the length should be tuned a little.
The patch antenna is shown in Figure 3.6:
43
Figure 3.6 Rectangular patch antenna (W=L=28.85mm)
The simulation process and results are shown in Figure 3.7 and 3.8. Firstly,
antenna is designed in schematic model (Figure 3.7a), simulation result is shown in
Figure 3.7b. From the graph, S11 is -3.2 dB and matching at 2.4 GHz. Secondly,
antenna is designed in layout model (Figure 3.8a), simulation result is shown in
Figure 3.8b. From the graph, S11 is - 4.893 dB and matching at 2.4 GHz. The
antenna dimension tuning during the layout simulation is shown in Table 3.1.
a. Schematic circuit
b. Simulation result (S11)
Figure 3.7 Schematic design and result of square antenna in ADS
44
a. Square antenna layout
b. Simulation result (S11)
Figure 3.8 Layout design and result of square antenna in ADS
Table 3.1 Square antenna dimension tuning and results
3.2.7 Circular Polarized Patch Antenna Design
Circular polarized antenna design is based on the liner polarized antenna
designed in 3.2.5. The square patch of the above is cut on one corner, this cut width is
3mm,
45
a. Square patch cut on one corner
b. Layout of antenna cut on one corner
Figure 3.9 Square patch cut on one corner and simulation layout
which is shown in Figure 3.9a. The simulation of this patch antenna (Figure 3.9b )
gives the S11 is -3.78 dB(magnitude: 0.419), this patch impedance ZL can be calculated
using Equation (3.7), ZL=122 ohm. This S11 is not good enough, in order
S11 =( ZL－Z0)/ (ZL＋Z0)
(3.7)
to get a good return loss and wide bandwidth, a quarter wave transformer is added
between the patch and 50 ohm transmission line to match them. ZL and Z0 are known
and substituted in Equation (3.12), the quarter wave transformer impedance ZT can be
got, ZT =78 ohm. Using this number in the LineCalc tool of ADS, the approximate
width (W2) and length (L2) of this transformer can be got: W2=1.22mm, L2 =17.3mm
This circular polarized antenna layout and simulation results (S11) are shown in
Figure 3.10, radiation pattern is shown in Figure 3.11. During the simulation, the patch
width (W), length (L) and the quarter wave transformer width (W2) and length (L2) were
tuned in order to get a good return loss and wide bandwidth and also match at 2.4 GHz.
Final Simulation Result:
W = 30mm, L = 28.5mm
W2=1.5mm, L2= 17mm,
Bandwidth: 135 MHz
46
a. Circular polarized antenna layout
b. Simulation result (S11)
Figure 3.10 Circular polarized antenna layout and simulation result
a. E-co side view
c. E-co and E-cross side view
b. E-cross side view
d. E-co and E-cross top view
Figure 3.11 Circular polarized antenna radiation pattern
After getting a good results in the simulation, this antenna is fabricated with
available microstrip materials (FR4) using wet etching technique. Antenna prototype are
shown in Appendix A.
47
3.3 Rat-race Coupler
Couplers and hybrids are components used in systems to combine or divide signals.
They are commonly used in antenna feeds, frequency discriminators, balanced mixers,
modulators, balanced amplifiers, phase shifters, monopulse comparators, automatic
signal level control, signal monitoring, and many other applications. A good coupler or
hybrid should have a good VSWR, low insertion loss, good isolation and directivity, and
constant coupling over a wide bandwidth.
Figure 3.13 A 180_ hybrid coupler. For a -3dB hybrid, ZR = √2Z0
In general, the performance of the coupler is specified by its coupling factor,
directivity, and terminating impedance. The isolated port is usually terminated by a
matched load. Low insertion loss and high directivity are desired features of the coupler.
Multisection designs are normally used to increase the bandwidth.
The 1800 hybrid has characteristics similar to the 900 hybrid except that the two
output signals are 1800 out of phase. As shown in Figure 3.13, a hybrid ring or rat-race
circuit can be used as a 1800 hybrid. For a 3-dB hybrid, the signal input at port 1 is split
into ports 2 and 3 equally but 1800 out of phase. Ports 1 and 4 are isolated. Similarly,
ports 2 and 3 are isolated. The input signal at port 4 is split into ports 2 and 3 equally,
48
but in phase. The characteristic impedance of the ring ZR = √2Z0 for a -3dB hybrid [11],
where Z0 is the characteristic impedance of the input and output ports.
3.3.1 Rat-race Coupler Design
The purpose of this rat-race coupler (Figure 3.14) is to connect the antenna with
transmit channel and receive channel and at the same time isolate these two channels. Its
working theory and structure are introduced in detail in 3.3. According to equation ZR =
√2Z0, ZR=70.71 ohm. Using this number in the LineCalc tool of ADS
a. Rat-race coupler with short
transmission line
b. Rat-race coupler with long
transmission line
Figure 3.14 Rat-race coupler
the width (W3) of the ring is 1.5 mm and quarter wave length is 17.24 mm. Radius =
6*17.24/2∏ = 16.46 mm.
After getting all the initial numbers, rat-race coupler is designed in schematic
model which is shown in Figure 3.15 and designed in layout which is shown in Figure
3.16.
49
a. Schematic circuit
b. Simulation result
Figure 3.15 Schematic design and result of rat-race coupler in ADS
a. Rat-race coupler layout
b. Simulation result
Figure 3.16 Schematic design and result of rat-race coupler in ADS
In order to match this coupler with the whole antenna circuit, the transmission
line length of 3 ports should be increased which is shown in Figure 3.14b, the tuning
options are L3 , L4 ,L5 ,L6 ,W3 and Radius. Simulation result is shown in Figure 3.17.
Final Simulation Results:
W3 = 1.53 mm, L3 = 52.26 mm
L4 = 31.18 mm, L5 = 20mm
L6 = 25.85 mm, Radius = 16.98 mm
50
a. Rat-race coupler S parameters
b. Rat-race coupler phase
Figure 3.17 Final rat-race coupler simulation result
Rat-race coupler is then fabricated with available microstrip materials (FR4)
using wet etching technique, measured using Network Analyzer. Appendix B shows the
fabrication prototype.
3.4 Amplifier
Generally, an amplifier is any device that uses a small amount of energy to control a
larger amount of energy. In popular use, the term today usually refers to an electronic
amplifier, often as applied to audio applications. The relationship of the input to the
output of an amplifier — usually expressed as a function of the input frequency — is
called the transfer function of the amplifier, and the magnitude of the transfer function is
termed the gain.
51
3.4.1 Amplifier Classes ( Power amplifier for Tx )
Amplifiers are commonly classified by the conduction angle (sometimes known as
'angle of flow') of the input signal through the amplifying device; see electronic
amplifier.
Class A:
Where efficiency is not a consideration, most small signal linear amplifiers are
designed as Class A, which means that the output devices are always in the conduction
region. Class A amplifiers are typically more linear and less complex than other types,
but are very inefficient. This type of amplifier is most commonly used in small-signal
stages or for low-power applications (such as driving headphones).
Class B:
In Class B, there are two output devices (or sets of output devices), each of which
conducts alternately for exactly 180 deg (or half cycle) of the input signal.
Class AB:
Class AB amplifiers are a compromise between Class A and B, which improves
small signal output linearity; conduction angles vary from 180 degrees upwards, selected
by the amplifier designer. Usually found in low frequency amplifiers (such as audio and
hi-fi) owing to their relatively high efficiency, or other designs where both linearity and
efficiency are important (cell phones, TV transmitters).
Class C:
Popular for high power RF amplifiers, Class C is defined by conduction for less
than 180° of the input signal. Linearity is not good, but this is of no significance for
single frequency power amplifiers. The signal is restored to near sinusoidal shape by a
tuned circuit, and efficiency is much higher than A, AB, or B classes of amplification.
Class D:
Class D amplifiers use switching to achieve a very high power efficiency (more than
90% in modern designs). By allowing each output device to be either fully on or off,
losses are minimized. A simple approach such as pulse-width modulation is sometimes
still used; however, high-performance switching amplifiers use digital techniques, such
52
as sigma-delta modulation, to achieve superior performance. Formerly used only for
subwoofers due to their limited bandwidth and relatively high distortion, the evolution of
semiconductor devices has made possible the development of high fidelity, full audio
range Class D amplifiers, with S/N and distortion levels similar to their linear
counterparts.
Other classes:
There are several other amplifier classes, although they are mainly variations of the
previous classes. For example, Class H and Class G amplifiers are marked by variation
of the supply rails (in discrete steps or in a continuous fashion, respectively) following
the input signal. Wasted heat on the output devices can be reduced as excess voltage is
kept to a minimum. The amplifier that is fed with these rails itself can be of any class.
These kinds of amplifiers are more complex, and are mainly used for specialized
applications, such as very high-power units. Also, Class E and Class F amplifiers are
commonly described in literature for radio frequencies applications where efficiency of
the traditional classes deviate substantially from their ideal values. These classes use
harmonic tuning of their output networks to achieve higher efficiency and can be
considered a subset of Class C due to their conduction angle characteristics.
Power Amplifier:
The term "power amplifier" is a relative term with respect to the amount of power
delivered to the load and/or sourced by the supply circuit. In general a power amplifier is
designated as the last amplifier in a transmission chain and is the amplifier stage that
typically requires most attention to power efficiency. For these reasons, a power
amplifier is typically any of the above-mentioned classes except Class A.
53
3.4.2 Transistor Amplifiers
The essential role of this active element is to magnify an input signal to yield a
significantly larger output signal. The amount of magnification (the "forward gain") is
determined by the external circuit design as well as the active device.
Many common active devices in transistor amplifiers are bipolar junction transistors
(BJTs) and metal oxide semiconductor field-effect transistors (MOSFETs).
Applications are numerous, some common examples are audio amplifiers in a home
stereo or PA system, RF high power generation for semiconductor equipment, to RF and
Microwave applications such as radio transmitters.
3.4.3 FET Transistor
What’s a FET? In microwaves we are almost always referring to a MESFET, which
stands for metal-semiconductor field effect transistor. A FET is a three terminal device
capable of both microwave amplification and switching. The FET’s three terminals are
denoted as gate, source and drain. With respect to a bipolar transistor (BJT), the gate of
a FET corresponds to the base of a BJT, the drain corresponds to the collector and the
source corresponds to the emitter terminal. This is useful knowledge since every curve
tracer we’ve ever seen in a lab has its three terminals labeled collector, base, and emitter,
not drain, gate, and source.
Used as an amplifier, the gate is most often configured as the input terminal, the
source is grounded and the drain is the output. The output current (IDS) is controlled by
the input voltage (VGS). This configuration is called common source since the source is
common to the input and output ground connections. It is also possible (but unusual) to
ground the gate and create a common-gate amplifier. Such an amplifier does not
provide the voltage gain of the common-source amplifier, but it has the interesting
property of being easier to impedance match than a "normal" common-source amplifier.
54
The figure below shows a cross-section of the channel of a field-effect transistor and
explains some FET terminology. The drain and source are connected by the FET
channel, which is formed by creating a mesa of N-type semiconductor (for an N-channel
FET) on top of a semi-insulating substrate (typically GaAs). In microwaves we are
almost often dealing with N-channel FETs. P-channel FETs are possible but are never
used at microwave frequencies, because they would have far worse performance
compared to N-channel FETs. Go ask a device guy why that is and he will explain to
you something about the electron mobility of the device, but who really cares? The
drain and source contacts are connected to the channel with ohmic metal contacts that
form low-resistance connections to these terminals. The gate connection to the channel
is formed between the drain and source by a Schottky metal contact to the channel. The
rectifying property of the gate contact means that when it is reverse biased with respect
to the channel it conducts almost zero DC current (IGS) to the channel, but its electric
field can be used to effectively displace the electrons within the channel. Thus an AC
voltage incident on the gate terminal causes a variable resistance between the source and
drain of the FET. When the gate reaches pinch-off voltage the electrons below the gate
are depleted to the point where essentially no current can flow from drain to source.
The source connection is the "source" of electrons in the channel, and the drain is
where they are "drained off". Remember that we are talking about electrons flowing
here, and you will see that the direction of current flow is positive from drain to source.
55
Figure 3.18 FET channel cross-section
The FET is built on top of a semi-insulating substrate, most often GaAs. When we
say “semi-insulating” this is perhaps misleading. In its pure form, GaAs is remarkable
insulator, which is what makes monolithic microwave integrated circuits (MMICs)
practical. Here is one advantage GaAs has over silicon. Pure silicon is a better
conductor than pure GaAs, so it tends to dissipate electrical fields that are needed to
support transmission modes and hence needs some "help" to be used as a MMIC.
The operation of any three-terminal device is well described on 2-D plots of IV
curves. For an FET, the output characteristics VDS and IDS are shown to be a function
of the input voltage VGS. A typical FET response is shown below.
56
Figure 3.19 FET IV characteristics and definitions
IMAX: the drain-source current when the gate is forward biased for maximum channel
current. This is typically measured at up to 1.0 volts on the gate (higher potentials will
conduct tons of current across the gate Schottky contact which tends to roast your FET)
and perhaps 1.5 or 2 Volts drain-to-source. To get to IMAX the gate must be raised to its
Schottky barrier height (voltage), which is approximately 0.7 volts. This is the intrinsic
gate bias. The other 0.3 volts will drop across the intrinsic source resistance RS. Still,
you might want to limit the measurement current with a current-limiting resistor....
IDSS: the saturated drain-source current when the gate is biased at zero volts (grounded
to the source). This is typically measured at 1.5 or 2 Volts drain-to-source.
VPO: pinch-off voltage. This is where the drain-source terminals start to look like an
open circuit, and no appreciable current flows even at high drain-source potentials. In
practice there is always some residual current and the actual VPO measurement must
make an allowance for this. For example, the pinch-off voltage could be measured at
2.5% of IDSS and VDS=2 volts.
VBR: the gate-drain breakdown voltage, which is indirectly measured on the IV
curves. At high drain-source potential and near pinch-off, the IV curves tend to bend
57
up. As shown in the picture the breakdown voltage VDS is approximately 10 volts
(VGS=-4 volts and VDS=6 volts combined). Stay away from this bias region if you
want your FET to have a long and happy life!
Knee voltage: the voltage at which the curves transition from "linear" to "saturation". In
the linear region, IDS depends on both VGS and VDS (from VDS=0 Volts to
approximately VDS=2 Volts). In the saturation region, IDS depends mainly on VGS and
not VDS. This is the right side of the curve, beyond VDS=2 volts.
Where did we get the nice transfer curve shown in the above plot? We have
developed a model that allows the user to fit a continuous transfer curve to measured
data, with separate coefficients to fit the regions above and below VGS=0 Volts. The
equations are shown below. By using two different exponent terms, is possible to control
the ratio of IMAX/IDSS, which is impossible in simpler models.
(3.8)
(3.9)
for 0<VGS≤VMAX
(3.10)
(3.11)
58
3.4.4 Low Noise Amplifier ( Rx )
LNA is typically used in receiving and transmitting applications because it has very
low noise figure (NF). NF is a measurement of noise in two-port network. LNA is made
up of a transistor and Matching Networks. Matching Networks are used in microwave
circuits to transform impedance from one value to another in order to eliminate
reflection or improve circuit performance such as gain or noise figure. For example, in a
regular Receiving Antenna, Matching Networks are inserted between the output of the
antenna and the input of the transistor, and also at the output of the transistor, as shown
below:
Figure 3.20 LNA conjugate matching
Not only transforming impedance, Matching Network also provides gain. For the
above Receiving Antenna, in order to get the highest gain, Matching Network A must
transform the output impedance of Antenna to the conjugate of the input impedance of
transistor, and Matching Network B transform the 50-ohm load impedance to the
conjugate of the output impedance of transistor. This is called conjugate matching.
Although the LNA has the highest gain, its NF is not the best.
3.5 Amplifier’s Biasing Circuit
RF/Microwave transistors/FET requires some form of circuit to set the correct bias
conditions for a particular RF performance. What is required is a low DC resistance but
a high RF resistance to ensure that the RF circuit is not loaded and RF signals do not
59
flow onto the supply lines. One probable and commonly used method is to place a radial
stub immediately after l/4 high impedance bias line [14]. This helps to achieve proper
isolation at desired RF frequency, no matter what component is added after l/4 long bias
line.
3.5.1 Quarter-wavelength
Referring to a Smith chart, if you are already at a matched impedance condition, any
length of transmission line at the system characteristic impedance Z0 does nothing to
your input match. But if the reflection coefficient of your network (S11 for example) is at
some non-ideal impedance, adding transmission line between the network and the
reference plane rotates the observed reflection coefficient clockwise about the center of
the Smith chart. Further, the rotation occurs at a fixed radius (and VSWR or return loss
magnitude) if the transmission line has the same characteristic impedance as the source
impedance Z0. By the time you have added a quarter-wavelength, you have gone 180
degrees about the center of the Smith chart.
Suppose your network was a short circuit, the left "corner" of the Smith Chart.
Adding one quarter-wavelength moves you 180 degrees to the right side of the chart, to
an open circuit. If a quarter-wave is 90 degrees in phase length, why does it transform
you 180 degrees on a Smith chart? Consider that here we are plotting reflection
coefficients on the Smith chart. Thus an imaginary signal that you send through a
transmission line of one quarter wavelength must travel fully half a wavelength, since it
travels down the quarter-wave line, gets reflected, then returns down the quarter-wave
line. So when you are plotting reflection coefficients, moving in a complete "circle"
means only adding 180 degrees!
60
3.5.2 Quarter-wave stubs
Starting with an open circuit, one quarter wavelength away you will "see" a short
circuit. Starting from a short circuit, one quarter wave away you have an open circuit.
Thus you can create an "RF open circuit" that is a DC short circuit, and vice versa.
These two properties are used to create DC and/or RF grounds for circuits, bias tees, and
much more.
Below are ADS models of a open circuit and a short circuit stub. The electrical
length of each stub is 90 degrees.
Figure 3.21 ADS models of open and short circuit stub
The following figure shows the reflection coefficient of the above stubs, versus
frequency, starting at DC and sweeping up to the quarter-wave frequency (10 GHz,
denoted by markers M1 and M2.) The open circuit S11, plotted on the left sweeps from
an open to a short, while the short circuit (S22, on the right) sweeps from a short to an
open. Note that both responses sweep clockwise with frequency, and would keep going
around and around the Smith chart if you kept increasing frequency.
61
Figure 3.22 Reflection coefficient
3.5.3 Quarter-wave transformers
Suppose you want to match a device of a different impedance than your system
impedance. A simple quarter-wave transformer can do this for you, with bandwidth
somewhat inversely proportional to the relative mismatch you are trying to overcome.
For a single-stage quarter-wave transformer, the correct transformer impedance is the
geometric mean between the impedances of the load and the source:
ZT=(ZL*ZS)^0.5
(3.12)
We will deal only with purely real impedances here, but transformers can be used
successfully to impedance match loads with reactive components as well. The problem
is that the math gets indeterminate quickly!
62
3.5.4 Microstrip radial stub
The open circuit stub trick (creating an RF short circuit that is DC-open) is often
done with lower impedance lines than Z0. This effectively gives a wider bandwidth. An
even better trick is to us a microstrip radial stub; the radial stub is widely used as a
bypass capacitor in the bias circuit of microstrip amplifiers. It provides a low impedance,
it doesn't suffer from a large distributed tee junction that a constant-width lowimpedance stub would, and it just works better.
What we are looking for with an open-circuit stub is a high return loss, like -1 dB.
The plot below shows that. Why is a radial stub better? We think it's because it has
better bandwidth and a larger fringing capacitance at the open end, which adds a
"lumped-element" quality to it.
Keywords:
- At connection point, Width should be small
- At the open end, Width should be wide
- At open end, corners should be avoided (reduce charge accumulation due to point
effect), so here's why it's rounded.
Figure 3.23 The layout of a radial stub
The radial stub is formed by a ring section with the inner radius ri, outer radius ro and
the angle a (see Figure 3.23). At the reference plane, which is placed in the distance
63
d = ri cosａ/2 from the ring center, width of the microstrip line is w = 2ri sinａ/2. The
model of the radial stub, used in the ADS system, is based on a stair-case approximation
of the stub shape with short sections of the microstrip line.
3.5.6 Amplifier’s Biasing Circuit Design
The purpose of this amplifier biasing circuit is to bias the amplifier DC input,
make the amplifier work and match with the whole antenna circuit. Its working theory
and structure has been introduced above. From Figure 3.24, the radial stub angle is
chosen by 700, radius and the transmission line between the radial stub end to the main
circuit both are quarter wave and the width also both are chosen by 0.7 mm. Using this
number in the LineCalc tool of ADS, the length of the quarter wave transformer is 17.96
mm.
Figure 3.24 Amplifier biasing circuit layout
Amplifier biasing circuit passive part is then fabricated with available microstrip
materials (FR4) using wet etching technique, active part is soldered on the passive
substrate and then measured whole circuit using Power Supply and Multimeter.
Appendix D shows the fabrication prototype.
64
3.6 T-junction Power Divider
Power dividers are passive devices that are designed to combine multiple antennas
in stacked antenna systems, while providing a constant 50 ohm impedance over the
bandwidth chosen. The advantages of this system are wider bandwidth and lower loss
than that of odd multiple quarter wave cable trimming solutions. Custom trimming for
best VSWR is time consuming. Power dividers provide simple, repeatable performance
improvements for stacked antenna systems.
Power dividers and combiners are used in microwave frequencies because of the
different behavior of circuits. In microwave frequencies, the modeling of the circuits are
different from the expected lumped model. Therefore, power dividers in microwave
frequencies are not just resistive power dividers as in the low frequency or DC model.
The important issues while using power dividers, are the specifications or properties of
the dividers. In microwave frequency applications, for example power can be reflected
from the output ports. Then to avoid the reflected power expected power divider should
satisfy the maximum power transfer theory, which says the impedance of the load
should be the complex conjugate of the impedance seen from the load end when load is
not connected. This is called "matching" of the ports. This leads to transferring
maximum power to the load and because of no reflection, avoid to burn out the power
supply. Also, again for maximally transfer the power to the output ports, the network
should not dissipate power and this property is called "losslessness".
Figure 3.25 The Power Divider
To efficiently divide power, the input port (port 1) must first be matched:S11 = 0
65
Likewise, the divided power must be efficiently (i.e, without loss) delivered to the
output ports of the divider:
S21 2+S31 2= 1
(3.13)
In addition, it is desirable that ports 2 and 3 be matched: S 22 = S 33 = 0
And also desirable that ports 2 and 3 be isolated:
S 23 = S 32 = 0
This ensures that no signal incident on port 2 will “leak” into port 3—and vice versa.
These conditions completely describe an ideal power divider conditions that can be met.
Power and Impedance calculation:
Yin=1/Z1 +1/Z2=1/Z0 (3.14)
Pin=1/2*V02/Z0
(3.15)
P1=1/2*V02/Z1
(3.16)
P2=1/2*V02/Z2
(3.17)
P1
Z1
Pin Z0
Z2
Yin
P2
Figure 3.26 T-junction Power Divider
66
3.6.1 T-junction Power Divider Design
The purpose of this T-junction power divider (Figure 3.27) is to connect transmit
channel and receive channel to the input & output port. Its working theory and structure
are introduced in detail in chapter 3. According to equation Z = √2Z0, Z=70.71 ohm.
Using this number in the LineCalc tool of ADS, the width (W4) of the quarter wave
transformer is 1.5 mm and quarter wave length is 17.24 mm.
After getting all the initial numbers, T-junction power divider is designed in
schematic model which is shown in Figure 3.28a and designed in layout which is shown
in Figure 3.29a. In order to match it with the whole antenna circuit, the transmission
line length should be increased which is shown in Figure 3.27b, the tuning options are
L7, L8, L9, W4 (Table 3.2 and Table 3.3 show the parameters changing while tuning)
Simulation result is shown in Figure 3.29b.
Table 3.2 S21 and S31 changing while L8 tuning
Table 3.3 S21 and S31 changing while L7, L8, L9 tuning
67
Final Simulation Results:
W4 =1.51 mm, L7 = 14.52 mm, L8 = 8.43 mm, L9 = 17.56 mm
a. Power divider with short
transmission line
b. Power divider with long
transmission line
Figure 3.27 T-junction power divider
a. T-junction power divider schematic circuit
b. T-junction power divider simulation result
Figure 3.28 Schematic design and result of T-junction power divider in ADS
68
a. Power divider layout
b. Simulation result
Figure 3.29 Layout design and result of T-junction power divider in ADS
T-junction power divider is then fabricated with available microstrip materials
(FR4) using wet etching technique, measured using Network Analyzer. Appendix C
shows the fabrication prototype.
3.7 Combination Design
The combination is consisted of combination of coupler and amplifier and
combination of the whole circuit except antenna. The purpose of the first combination is
to design and simulate the isolation of the transmit channel and receive channel. The
purpose of the second combination is to design and simulate the transmit channel Gain
and receive channel Gain.
3.7.1 Combination of Coupler and Amplifier
This combination can only be designed and simulated in schematic model. From
Figure 3.30, the rat-race coupler designed before is represented by a 4 ports Data Item,
the amplifier is represented by a 2 ports Data Item. The coupler S parameter is
69
imported into Data Item from the simulation graph file. The amplifier S parameters is
input into a Mini-circuit file first and then imported into Data Item from this file.
Simulation result is shown in Figure 3.31.
Figure 3.30 Schematic combination of coupler and amplifier in ADS
Figure 3.31 Transmit and receive channel isolation simulation result
3.7.2 Combination of Whole Circuit except Antenna
This combination also can only be designed and simulated in schematic model.
From Figure 3.32, it is based on the first combination and just adds the power divider.
This power divider is represented by a 3 ports Data Item and S parameter import is same
70
with coupler. Simulation result is shown in Figure 3.33. Table 3.4 and Figure 3.34
show the transmit and receive channel gain changing with the tuning of L9.
Figure 3.32 Schematic combination of whole circuit except antenna in ADS
Table 3.4 S21 and S12 changing
While L9 tuning
Figure 3.33 Graph showing of Table 3.4
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Figure 3.34 Simulation result of transmit and receive channel gain
The final whole antenna layout is shown in Figure 3.35, resistor and capacitor are
added. After fabricating the passive patch of this antenna, the transistor, resistor,
capacitor, port and conduction line are soldered on this patch. Final fabrication prototype
is shown in Appendix E.
72
Figure 3.35 Final antenna layout
3.8 Summary
This chapter discussed the theory of all the parts of this design system and some
relevant field knowledge and also presented the design of single elements of AIA and
the whole AIA design. The ADS simulations give a good simulation results. After the
simulation, all these parts have been fabricated using FR4 materials. The antenna is
tested to study the performance of the prototype antenna. The results of the elements and
AIA are presented in chapter 5.
CHAPTER 4
MEASUREMENT SETUP
4.1 Introduction
All system elements and whole antenna have been fabricated. Next is the
measurement of different parts. In this chapter, the setup of different measurement will
be presented separately, it includes the setup for biasing circuit testing, setup for
radiation pattern and gain measurement and setup for power strength measurement.
4.2 Biasing Circuit Testing Setup
This biasing circuit has been fabricated, after the resistor (68 ohm), capacitor
(100 pF), transistor (ATF-21186) and conduction line are soldered with the passive
patch, this circuit is measured using Power Supply and Multimeter (Figure 4.1).
1. Switching off the Power Supply (putting voltage on 0) and Multimeter;.
74
2. Using 2 conduction lines to connect the drain and ground of biasing circuit to one
Power supply (Vds) which is positive. Using another 2 conduction lines to connect
the gate and ground of biasing circuit to another Power supply (Vgs) which is
negative.
3. Switching on the Power Supply, putting Vgs a certain number then changing Vds
little by little. Changing Vgs a little, doing the changing of Vds again until biasing
current is 60 mA.
4. Using multimeter to measure different end to end voltage in this circuit during the
testing.
Figure 4.1 Measurement of amplifier biasing circuit
4.3 Radiation Pattern and Gain Measurement Setup
This active antenna radiation pattern has been measured in the anechoic Chamber
( Figure 4.2). As shown in Figure 4.3 and Figure 4.4, the antenna under test will be
placed on the rotating machine. For transmit operation, this AIA will be the transmitting
75
antenna while the reference antenna which is a horn antenna will be the receiver. While
for receive operation, this AIA will be the receiving and the horn antenna will be
transmitting.
Both the AIA and the reference antenna will be aligned in the chamber and will be
rotated according to the polarization that is going to be measured. The data collection
can be eased by reducing the angle interval of rotation at 20.
(a) Power Supply of AIA
76
(b) Setup of AIA on the rotator
Figure 4.2 Measurement of antenna radiation pattern setup
Antenna(Rx)
Reference
Antenna
Signal
Generator
Rotator
Power Supply
Figure 4.3 Radiation pattern measurement setup (Rx)
Spectrum
Analyzer
77
Antenna (Tx)
Reference
Antenna
Rotator
Spectrum
Analyzer
Power Supply
Signal
Generator
Figure 4.4 Radiation pattern measurement setup (Tx)
4.4 Power Strength Measurement Setup
The power strength measurement is using Network Stumbler, the setup for the
measurement and comparison follows this below procedure.
1. Choosing the distance between the transmit antenna and the receive computer.
2. Opening the monopole antenna and measuring the power strength by running
the software.
3. Taking monopole antenna out of the base device, running software for a certain
time in which the graph is becoming stable.
4. Connecting AIA without power supply to the base device, running software for a
certain time in which the graph is becoming stable.
5. Adding power to this AIA, running software for a certain time in which the graph
78
is becoming stable.
Figure 4.5 and Figure 4.6 show the measurement setup of different distance (
5m and 15m ).
Figure 4.5 Active antenna power strength comparison with monopole antenna (5m)
Figure 4.6 Active antenna power strength comparison with monopole antenna (15m)
79
4.5 Summary
This chapter presented the setup of different measurement. The setup for biasing
circuit testing, setup for radiation pattern and gain measurement and setup for power
strength measurement all have been presented in detail. These setup have been done to
do good preparation for the measurement.
CHAPTER 5
RESULTS COMPARISON AND ANALYSIS
5.1 Introduction
Antenna design, simulation and fabrication have been completed, chapter 3 gives a
detailed description of this process. The simulation results for different parts have been
obtained through using ADS software, in this chapter, the measurement of different
elements will be done separately first, next the whole antenna measurement and
application in the real environment will be presented and in terms of the simulation
results and the measurement results the comparison and analysis will be done.
5.2 Passive Antenna Analysis
The passive circular polarized patch antenna return loss has been measured using
Network Analyzer which is shown in Figure 5.2. The antenna radiation pattern is
measured in the Chamber, Figure 5.1 shows the results.
81
Radiation Pattern
0
30
330
-35
-40
-45
-50
60
300
-55
-60
-65
90
-70
-35 -40 -45 -50 -55 -60 -65 -70 -65 -60 -55 -50 -45 -40 -35
-65
27
-60
-55
120
240
-50
-45
-40
150
-35
210
180
E-Copolarization vs Degree
E-Crosspolarization vs Degree
Radiation Pattern
0
30
330
-35
-40
-45
-50
60
300
-55
-60
-65
90
-70
-35 -40 -45 -50 -55 -60 -65 -70 -65 -60 -55 -50 -45 -40 -35
-65
27
-60
-55
120
240
-50
-45
-40
150
-35
210
180
H-Copolarization vs Degree
H-crosspolarization vs Degree
Figure 5.1 Measurement result of passive antenna radiation pattern
82
From these simulation results and measurement results of radiation pattern, Eplane and H-plane pattern are similar, and co-polarization and cross-polarization in both
plane are also similar, like a half circle from -900 to 900. So this antenna is a circular
polarized antenna. The half power beam width (HPBW) for this antenna at E plane is
about 720 with cross isolation at 00 is about 0 dB.The half power beam width (HPBW)
for this antenna at H plane is about 700 with cross isolation at 00 is about 0 dB.
Figure 5.2 Circular polarized single antenna measurement result (S11)
0
- 2
S11( dB)
- 4
- 6
- 8
- 10
- 12
- 14
- 16
- 18
1. 6
1. 8
2. 0
2. 2
2. 4
2. 6
2. 8
Fr equenc y
Mo me n t u m S u mu l a t i o n
Me a s u r e me n t
Figure 5.3Comparison of simulation and measurement return loss result
3. 0
83
Figure 5.3 shows the comparison between simulation and measurement return loss
result. From this graph, the simulation and measurement results S11 both are lower than 10 dB around 2.4 GHz. Simulation bandwidth is 135 MHz, bandwidth percentage is
about 5.6%, measurement bandwidth is 160 MHz, bandwidth percentage is about 6.7%,
they are all more than 100MHz and percentage is more than 4%. But measurement result
shifts right around 0.04 GHz because of the material, and measurement result is better
than simulation result, this is because in design process simulation is not good enough,
the best point was not got in the simulation.
Above all, this passive circular polarized antenna has been designed, simulated,
fabricated and measured successfully. There is an advice to all the designer: during the
design process, the experiment of simulation should be done numerously and carefully
until the best result is got.
5.3 Rat-race Coupler Analysis
The rat-race coupler has been measured using Network Analyzer which is shown
in Figure 5.4. Comparing the simulation and measurement results, ADS give a good
simulation result, return loss (S11) is -22 dB at 2.4 GHz, bandwidth is wider and phase
difference between port 2 to port 4 (-90.3910) and port 2 to port 1 (90.2840) is 1800. The
measurement return loss is -24 dB at 2.4 GHz, bandwidth is narrower than the
simulation but still wide enough, the phase difference between port 2 to port 4 (-107.520)
and port 2 to port 1 (77.760) is 1850 which is approximate to 1800.
According to the above analysis, this coupler design has been achieved.
84
a. Coupler S11(-24 dB at 2.4 GHz)
b. Coupler phase of S12 (770)
c. Coupler phase of S42 (-1070)
Figure 5.4 Rat-race coupler measurement results
5.4 T-junction Power Divider Analysis
The T-junction power divider has been measured which is shown in Figure 5.5.
Comparing the simulation and measurement results, ADS give a good simulation result,
return loss (S11) is lower than -40 dB at 2.4 GHz, bandwidth is wider and the gain from
port 1 to port 2 and port 1 to port 3 is -3.502 dB (44.6%). The measurement return loss is
-15.5 dB at 2.4 GHz, bandwidth is similar with the simulation but this graph shifts left
85
0.34 GHz and the best matching point at 2.06 GHz which is because of the material and
the fabrication error. The gain from port 1 to port 2 and port 1 to port 3 is -4.6 dB
(34.6%), it is worse than the simulation because the loss of material and the port.
a. Measurement result S11
b. Measurement result (S21)
(matching point at 2.06 GHz)
Figure 5.5 Power divider measurement results
This T-junction power divider has been designed, simulated, fabricated and
measured. ADS give a good simulation result, but the measurement result is not good
enough, and there is -4.6 dB loss in this component, it is a problem of this design, this
component should be replaced by another component in the future work.
Figure 5.6 Measurement of T-junction power divider
86
5.5 Amplifier’s Biasing Circuit Analysis
This biasing circuit has been fabricated which is shown in Appendix C, after the
resistor (68 ohm), capacitor (100 pF), transistor (ATF-21186) and conduction line are
soldered with the passive patch, this circuit is measured using Power Supply and
Multimeter.
Amplifier biasing circuit is supplied by two power, one is drain voltage (Vd) which
is positive and another is gate voltage (Vg) which is negative. When Vg is fixed at a
certain number, the current (Ids) from drain to source will change while tuning Vd, and
when Vd is fixed at a certain number, Ids also will change while tuning Vg.
According to the above testing analysis, this biasing circuit is functioning correctly
in terms of the transistor theory and Datasheet of the transistor ATF-21186.
.
5.6 Whole Antenna Analysis
After all the single part of this antenna has been designed, fabricated and
measured, this whole antenna is fabricated which is shown in Appendix E, and the
measurement and comparison and application of this antenna are presented in this
section.
5.6.1 Antenna Radiation Pattern Measurement and Analysis
This antenna radiation pattern has been measured in the Chamber, and the result
is shown in Figure 5.6, Figure 5.7.
87
Radiation Pattern
0
30
330
-35
-40
-45
60
300
-50
-55
90
-60
-35
-40
-45
-50
-55
27
-60
-55
-50
-45
-40
-35
-55
-50
120
240
-45
-40
-35
150
H_Co vs Degree
H_Cross vs Degree
210
180
Radiation Pattern
0
30
330
-35
-40
-45
60
300
-50
-55
90
27
-60
-35
-40
-45
-50
-55
-60
-55
-50
-45
-40
-35
-55
-50
120
240
-45
-40
150
E_Co vs Degree
E_Cross vs Degree
-35
210
180
Figure 5.6 Antenna radiation pattern measurement result (Tx)
88
Radiation Pattern
0
-30
30
330
-35
-40
60
300
-45
-50
-55
90
-30
-35
-40
-45
-50
-60
-55
-60
27
-55
-50
-45
-40
-35
-30
-55
-50
-45
120
240
-40
-35
150
210
-30
H_Co vs Degree
H_Cross vs Degree
180
Radiation Pattern
0
-30
30
330
-35
-40
60
300
-45
-50
-55
90
-30
-35
-40
-45
-50
-60
-55 -60
-55
27
-55
-50
-45
-40
-35
-30
-50
-45
120
240
-40
-35
150
E_Co vs Degree
E_Cross vs Degree
-30
210
180
Figure 5.7 Antenna radiation pattern measurement result (Rx)
89
From the graph, in transmit and receive channel E-plane and H-plane pattern are
similar and co-polarization and cross-polarization in both plane are also similar, like a
half circle from -900 to 900. This active result compares with passive result, they are also
similar, but the active antenna result is worse than the passive antenna, in some degree
the radiate power is very low, this is because the transistor is unstable. During the
measurement, Vgs and Vds can not keep a constant number, they are unstable around
biasing voltage (Vgs = -1.2 V, Vds = 2.0 V, Ids = 60mA). The half power beam width
(HPBW) for this antenna (Tx) at E plane is about 680 with cross isolation at 00 is about 0
dB.The half power beam width (HPBW) for this antenna (Tx) at H plane is about 650
with cross isolation at 00 is about 0 dB. The half power beam width (HPBW) for this
antenna (Rx) at E plane is about 660 with cross isolation at 00 is about 0 dB.The half
power beam width (HPBW) for this antenna (Tx) at H plane is about 630 with cross
isolation at 00 is about 1 dB.
Above all, the active antenna is working both the transmit channel and the
receive channel, but this transistor is not stable during the operation.
5.6.2 Antenna Comparison with Monopole Antenna and Analysis
Antenna gain has been measured in the Chamber, and the result comparison with
the monopole antenna gain is shown in Figure 5.8, the result comparison with passive
antenna is shown in Figure 5.9. From the first graph, the transmit and receive gain is
lower than monopole antenna from 2 GHz to3 GHz, and both of them are around 4 dB
lower than the monopole around 2.4 GHz. From the second graph, active and passive
antenna gain are all lower than monopole antenna, and active antenna gain is about 2 dB
higher than passive antenna around 2.4GHz.
90
dBm
- 25
- 30
- 35
- 40
- 45
- 50
- 55
- 60
- 65
2
2. 1
Tr ans mi t
Rec ei ve
Monopol e
2. 2
2. 3
2. 4 2. 5 2. 6
Fr equenc y
2. 7
2. 8
2. 9
3
Figure 5.8 Active antenna gain comparison with monopole antenna
-25
-30
dBm
-35
-40
-45
-50
2
2. 1 2. 2 2. 3 2. 4 2. 5 2. 6 2. 7 2. 8 2. 9
Frequency
Act i ve ant enna
Monopol e ant enna
Passi ve ant enna
3
Figure 5.9 Active and passive antenna gain comparison with monopole antenna
91
Another comparison between active antenna and monopole antenna has been
done inside and outside the laboratory, results are shown in Figure 5.10 and Figure
5.11.
Figure 5.10 Active antenna power strength comparison with monopole (5m)
92
Figure 5.11 Active antenna power strength comparison with monopole (15m)
From the graph, no matter inside (5m) or outside (15m) the monople antenna
power strength is the first part; and if there is no antenna the power strength decreases to
the second part; after a certain time passive antenna is connected with the device, so the
graph increases to the third part; when the power strength is stable the power supply is
added to the passive antenna, the power strength increases to the same level like
monopole antenna.
So from the above measurement and comparison, the active and passive antenna
gain is not better than the monopole antenna, the active antenna gain is higher than
passive antenna at resonating frequency, the active antenna gain is amplified, it is
working but not good enough because of the power loss of the power divider, unstable
of the active transistor and this transistor is not matched, so it is not working at the
optimization point.
5.7 Summary
This chapter has presented the results of all the elements and final antenna
fabricated that serve the purpose of the project. The active antenna shows a similar result
compared to the passive antenna in terms of radiation pattern. And active antenna does
improve the gain of this antenna compared to the passive antenna.
CHAPTER 6
CONCLUSION AND FUTURE WORK
6.1 Conclusion
The goal of this project is to produce design of Microstrip Antenna (Passive and
active), Rat-race Coupler ,Amplifier’s Biasing circuit and T-junction Power Divider
using ADS, and use these design to build prototype. These prototypes should then be
measured in order to determine if these designs can be successfully produced. All these
designs have been produced using ADS and functional prototypes have been built,
measured and analyzed in the facility in the laboratory of UTM. The active integrated
antenna with transmit and receive function has been designed, fabricated and measured.
Some conclusions that can be drawn from the obtained results in this project are the
following. The integration of the active device on to the passive antenna can improve the
gain of this antenna, the transmit and receive function can be integrated onto one single
patch. The antenna that has been tested are proven to be operational. Although the ADS
simulation gives good results, the measurement results are not good enough. A main
problem is that the active part is not stable and matching well with the passive part,
94
another problem is the power loss of the Power Divider, there is much power loss both
in the transmit and receive channel. In the future work, these problems should be solved.
As an overall conclusion, all the planned works and objectives of this project have
been successfully implemented and achieved.
6.2 Future Work
Suggested future work in this design is to do the matching of transistor first,
since transistor is not working at the optimization point. After doing the matching of
transistor, result is still not good enough, this transistor (ATF-21186) should be replaced
by another kind and redesign this AIA. Another future work is using other component to
replace T-junction Power Divider, since there is much power loss in this component.
95
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97
APPENDIX A
Circular polarized single antenna prototype
98
APPENDIX B
Rat-race coupler prototype
99
APPENDIX C
T-junction power divider prototype
100
APPENDIX D
Amplifier biasing circuit prototype
101
APPENDIX E
AIA with Tx & Rx prototype