PERFORMANCE STUDY OF PROXIMITY COUPLED STACKED
CONFIGURATION FOR WIDEBAND MICROSTRIP ANTENNA
ZULHANI BIN RASIN
UNIVERSITI TEKNOLOGI MALAYSIA
iii
To my loves
Noor Azilah
Muhammad Azhan Hakimi
Muhammad Azhan Wafi
iv
ACKNOWLEDGEMENT
First of all, syukur to Allah s.w.t for his blessing, I was able to complete this
project successfully.
I wish to express my sincere appreciation to my project supervisor, Dr. Nor
Hisham Bin Hj Khamis for his advice, guidance and help during the course of this
project. The comfort level that he created while supervising me, helped me a lot in
completing the project.
I also would like to thank several individuals at Wireless Communication
Centre, UTM for the help and guidance given while working for my project. Special
thanks also to colleagues at the Faculty of Electrical Engineering, Universiti
Teknikal Malaysia Melaka for their assistance and guidance.
My fellow postgraduate students should also be recognized for their support.
Their views and tips are useful indeed.
Last but not the least, my beloved family for all the support and
encouragement given. Without them, all this would not be achieved.
v
ABSTRACT
This project started by identifying two main disadvantages of the typical
microstrip antenna that are the low gain and narrow bandwidth. These two major
drawbacks have limited its application despite of other advantages as compared to
the conventional antenna. With the purpose of designing a wideband microstrip
antenna, the two already proven bandwidth enhancement techniques; the patch stack
configuration and coplanar parasitic patch was studied. Several antenna
configurations were proposed and from the simulation result, the antenna bandwidth
was improved from the typical 8 ~ 9 % up to 36 % by using these two techniques
using a simple coaxial probe feeding without any matching network. Actual
fabrication was also carried out and several measurements were conducts to verify its
performance. The measurement results, even not fully conform to the simulation
result, has proven the effectiveness of the above mentioned bandwidth enhancement
techniques.
vi
ABSTRAK
Projek ini bermula dengan mengenal pasti dua kekurangan utama yang
terdapat pada antena mikrostrip, iaitu gandaan yang rendah dan jalur lebar operasi
yang kecil. Kedua-dua kekurangan yang utama ini telah menghadkan aplikasi antena
mikrostrip walaupun terdapat banyak kelebihan-kelebihan lainnya berbanding antena
konvensional. Dengan objektif untuk menghasilkan antenna mikrostrip dengan jalur
lebar operasi yang luas, kajian terhadap dua teknik yang telah pun teruji mampu
untuk meningkatkan jalur lebar operasi, iaitu susunan secara bertingkat dan susunan
parasitik di atas satah yang sama telah dijalankan. Beberapa contoh konfigurasi
antena telah pun dihasilkan, dan berdasarkan keputusan daripada proses simulasi
yang telah dijalankan, jalur lebar operasi mampu ditingkatkan daripada hanya sekitar
8 ~ 9 % kepada lebih 36 % dengan menggunakan dua teknik tersebut. Antena yang
dihasilkan menggunakan “coaxial probe feeding” untuk memasukkan signal tanpa
menggunakan sebarang litar penyuai. Proses fabrikasi sebenar antena juga telah
dijalankan, dan beberapa pengukuran telah dilakukan untuk memastikan keupayaan
sebenarnya. Keputusan pengukuran yang telah dijalankan, walaupun tidak
sepenuhnya selaras dengan keputusan simulasi, telah membuktikan keberkesanan
dua teknik ini untuk meningkatkan jalur lebar operasi.
vii
TABLE OF CONTENTS
CHAPTER
1
2
TITLE
PAGE
DECLARATION
ii
DEDICATION
iii
ACKNOWLEDGEMENT
iv
ABSTRACT
v
ABSTRAK
vi
TABLE OF CONTENTS
vii
LIST OF TABLES
x
LIST OF FIGURES
xi
LIST OF SYMBOLS
xiv
LIST OF ABBREVIATIONS
xvii
INTRODUCTION
1
1.1
Thesis motivation
1
1.2
Thesis outline
2
BACKGROUND AND RELATED WORK
3
2.1
The Basic of Microstrip Antenna
3
2.2
Advantages/Disadvantages of Microstrip Antennas
4
viii
2.3
Radiation mechanism
6
2.4
TM Modes
8
2.5
Antenna polarization type
9
2.6
Feeding technique
11
2.6.1 Coaxial Probe Feeding
11
2.6.2 Microstrip coplanar feeding
12
2.6.3 Aperture coupled feeding
13
2.6.4 Proximity coupled feeding
13
Analysis Method
14
2.7.1 Transmission line model
15
2.7.2 Cavity model
19
2.7.3 Method of Moments (MoM)
22
2.8
Impedance bandwidth of the antenna
26
2.9
Effect of substrate parameters on bandwidth
27
2.10
Bandwidth enhancement technique
28
2.10.1 Stacked patches configuration
28
2.10.2 Coplanar parasitic patches
30
Application of microstrip patch antenna
31
2.7
2.11
3
THE DESIGN AND CONFIGURATION
32
3.1
Design Methodology
32
3.2
Basic configuration and parameter of the antenna
33
3.2.1 Patch shape
33
3.2.2 Feeding technique : Coaxial probe feeding
34
3.2.3 Determine the operating frequency
34
3.2.4 Input impedance matching and Smith chart
35
3.2.5 Voltage Standing Wave Ratio (VSWR)
37
3.2.6 Return Loss
39
3.2.7 Determine the bandwidth based on return
39
loss graph
3.2.8 Determine the probe feeding location
40
3.2.9 Determine the gain of the antenna
41
3.2.10 Antenna Radiation pattern
42
ix
4
3.2.11 Antenna directivity
43
3.2.12 Antenna efficiency
44
3.3
Method of Analysis
45
3.4
Effects of Finite Size Ground Plane
45
SIMULATION AND FABRICATION
47
4.1
Simulation process
47
4.1.1 Design 1 :single patch microstrip antenna
47
4.1.2 Design 2 : the use of stacked patch technique
50
4.1.3 Design 3 : the use of coplanar parasitic patch
52
4.1.4 Design 4 : combination of stacked patch with
54
coplanar parasitic patch configuration (case 1)
4.1.5 Design 5 : combination of stacked patch with
56
coplanar parasitic patch configuration (case 2)
4.1.6 Design 6 : combination of stacked patch with
58
coplanar parasitic patch configuration (case 3)
5
4.1.7 Summary of simulation results
60
4.2
Fabrication process
61
4.3
Antenna Measurement
64
4.3.1 Return loss measurement
64
4.3.2 Radiation pattern measurement
66
CONCLUSION AND FUTURE WORK
REFERENCES
69
70
x
LIST OF TABLES
TABLE NO.
2.1
TITLE
Typical microstrip patch antenna application
PAGE
31
xi
LIST OF FIGURES
FIGURE NO.
TITLE
PAGE
2.1
Typical microstrip patch antenna
4
2.2
Charge distribution and current density
6
2.3
Electric field and magnetic current distribution
8
2.4
Linearly (vertically) polarized wave
10
2.5
Linear and circular polarization of microstrip antenna
10
2.6
Coaxial probe feeding
12
2.7
Microstrip coplanar feeding
12
2.8
Aperture coupled feeding
13
2.9
Proximity coupled feeding
14
2.10
Microstrip line and electric field lines
16
2.11
Microstrip patch antenna
17
2.12
Top view and side view of patch antenna
18
2.13
Charge distribution and current density
20
2.14
Variation of radiation Q over substrate’s dielectric
27
constant
2.15
Variation of radiation Q over substrate’s thickness
28
2.16
Multilayer stacked patches configuration
29
2.17
Coplanar multi-resonator configuration
30
3.1
Methodology flowchart
33
3.2
Circular patch microstrip antenna with coaxial feeding
34
xii
3.3
Equivalent circuit of transmitting antenna
36
3.4
Smith chart diagram
37
3.5
Bandwidth determination from the return loss graph
40
3.6
Determination of probe feeding location
41
3.7
Spherical coordinate system for antenna radiation
43
4.1
Structure of Design 1
48
4.2
Return loss simulation result for Design 1
49
4.3
Radiation pattern simulation result for Design 1
49
4.4
Structure of Design 2
50
4.5
Return loss simulation result for Design 2
51
4.6
Input impedance and radiation pattern simulation result
51
for Design 2
4.7
Structure of Design 3
52
4.8
Return loss simulation result for Design 3
53
4.9
Input impedance and radiation pattern simulation result
53
for Design 3
4.10
Structure of Design 4
54
4.11
Return loss simulation result for Design 4
55
4.12
Input impedance and radiation pattern simulation result
55
for Design 4
4.13
Structure of Design 5
56
4.14
Return loss simulation result for Design 5
57
4.15
Input impedance and radiation pattern simulation result
57
for Design 5
4.16
Structure of Design 6
58
4.17
Return loss simulation result for Design 6
59
4.18
Input impedance and radiation pattern simulation result
59
for Design 6
4.19
Mask generation on transparency
61
4.20
FR4 material used for fabrication
62
4.21
Process of exposing the FR4 material under UV light
62
4.22
Soaking the FR4 material in the developer
63
4.23
Etching process using the ferrite chloride
63
4.24
Fabricated structure of the antenna
63
xiii
4.25
Return loss measurement using the microwave analyzer
64
4.26
Return loss measurement result for Design 4
65
4.27
Return loss measurement result for Design 5
66
4.28
Radiation pattern measurement result for Design 4
67
4.29
Radiation pattern measurement result for Design 5
68
xiv
LIST OF SYMBOLS
1.
Uppercase
D
-
directivity of the antenna
E
-
electric field
G
-
gain of antenna
H
-
magnetic field
Jb
-
current density at bottom surface of patch
Jt
-
current density at top surface of patch
L
-
length dimension of patch
∆L
-
extendend length of patch caused by fringing field effect
Ms
-
magnetic current density along the periphery of the patch
P
-
total power radiated from the antenna
Pc
-
conductor loss
Pd
-
dielectric loss
Pr
-
power radiated from the patch
Qc
-
quality factor of the conductor
Qd
-
quality factor of the dielectric
Qr
-
quality factor for radiation
QT
-
total antenna quality factor
Rin
-
antenna resistance at the terminals
RL
-
loss resistance
Rr
-
radiation resistance
xv
2.
3.
Rs
-
antenna source resistance
U
-
radiation intensity of the antenna
Ui
-
radiation intensity of an isotropic source
Vi
-
amplitude of the incident wave
Vr
-
amplitude of the reflected wave
W
-
width dimension of patch
WT
-
total energy stored during resonance
Xin
-
antenna reactance at the terminals
Xs
-
antenna source reactance
Zin
-
antenna impedance at the terminals
Zo
-
characteristic impedance along the transmission line
Zs
-
antenna source impedance
Lowercase
a
-
radius of the patch
ae
-
effective radius of patch caused by fringing field
c
-
speed of light 3 x 10 ms-1
ec
-
conduction efficiency
ecd
-
radiation efficiency of the antenna
ed
-
dielectric efficiency
er
-
reflection (mismatch) efficiency
et
-
total antenna efficiency
f
-
frequency
fH
-
upper frequency of the frequency range
fL
-
lower frequency of the frequency range
fo
-
resonance frequency
h
-
height of dielectric substrate
vo
-
speed of light 3 x 108 ms-1
xo
-
feeding point along the radius of the patch
Greek Symbol
εr
-
dielectric constant of substrate
xvi
εreff
-
effective dielectric constant
λ
-
wavelength in the dielectric medium
λo
-
wavelength in the free space
δ
-
loss tangent of the dielectric
δeff
-
effective loss tangent
ωr
-
angular resonance frequency
∆
-
skin depth of the conductor
π
-
phi = 3.142
Γ
-
reflection coefficient
θ
-
elevation angle of antenna radiation
φ
-
azimuthal angle of antenna radiation
xvii
LIST OF ABBREVIATIONS
3D
-
three dimension
BW
-
Bandwidth
dB
-
Desible
GHz
-
Giga Hertz
ISM band
-
Industrial, Science, Medical frequency band
LOS
-
Line of sight
MMIC
-
Monolithic Microwave Integrated Circuit
MoM
-
Method of Moments
RF
-
Radio Frequency
RHC
-
Right hand circular
RL
-
Return loss
TEM
-
Transverse Electric Magnetic
TM
-
Transversal Magnetic
UV
-
Ultra Violet
VSWR
-
Voltage Standing Wave Ratio
CHAPTER 1
INTRODUCTION
1.1
Thesis motivation
Despite of its advantages; light weight, low profile, easy fabrication and
conformability to mounting post etc as compared to the conventional microwave
antennas, narrow bandwidth and low gain are two major disadvantages that limit its
application. The compact configuration of microstrip antenna is the main factor to
these limitations. The smaller the antenna, either the operation bandwidth or the
antenna efficiency (gain) will be decreased. For that, the size reduction together with
gain and bandwidth enhancement has become a major consideration in the microstrip
antenna design. Many studies have been carried out and several proposed techniques
are proven to be able to improve the bandwidth performance and gain of the
microstrip antenna.
Several techniques such as stack configuration and co-planar parasitic patch
were proposed [1] and able to improve the bandwidth up to 20 %. Using a right
parameter configuration, further improvement is expected.
2
This thesis describes the theory, implementation and discusses the
performance of using the bandwidth enhancement techniques; the proximity coupled
stack configuration and the coplanar parasitic multi-resonator in the microstrip
antenna in order to improve the bandwidth performance.
1.2
Thesis outline
This thesis project starts with the literature study of the microstrip antenna in
order to get its basic fundamental and they are all concluded in chapter 2. Here, all
the main aspects of the microstrip antenna such as its structure configuration,
radiation mechanism, polarization, feeding techniques, method of analysis etc are
covered. Several techniques used in the enhancement of the bandwidth are also
included in this chapter.
Chapter 3 covers the necessary fundamental aspects for the implementation
of the antenna design. This chapter discusses about the configuration of the design
including the specification and parameter setting necessary before validation process
is carried out.
Validation process including the simulation and fabrication of the proposed
antenna is detailed in chapter 4. Here, different variables effects on the performance
of the antenna are described.
In chapter 5, based on the result obtained in previous chapter, the overall
performance of the proposed designs is concluded. Last but not the least, possible
improvement for future work is also outlined.
CHAPTER 2
BACKGROUND AND RELATED WORK
The concept of microstrip antenna was first proposed by Deschamps in 1953
[2]. The fundamental of this antenna is a metallic patch, usually copper printed on a
thin, grounded dielectric substrate. In the early stage, signal is fed to the antenna
either by a coaxial line through the bottom of the substrate, or by a coplanar
microstrip line. Here the metallic patch acts as a transitional structure radiating and
receiving the electromagnetic waves. Since the original configuration was proposed,
various kind of shape, feeding technique, substrate configuration has been proposed.
2.1
The Basic of Microstrip Antenna
Below Figure 2.1 shows the basic form of a microstrip patch antenna, a flat
plate over a ground plane. The center conductor of a coax serves as the feed probe to
couple electromagnetic energy in and/or out of the patch. The electric field
distribution of a rectangular patch excited in its fundamental mode is also indicated.
4
Figure 2.1
Typical microstrip patch antenna
The electric field is zero at the center of the patch, maximum (positive) at one
side, and minimum (negative) on the opposite side. The minimum and maximum
electric fields continuously change side according to the instantaneous phase of the
applied signal. The electric field shown above does not stop abruptly at the patch’s
periphery as in cavity, but the fields extend the outer periphery to some degree.
These fields extension are known as fringing fields and cause the patch to radiate.
The dielectric substrate act as an electrical insulator is a substance that is
highly resistant to the flow of electric current. Ideally, the dielectric constant εr of the
substrate should be low (εr < 2.5) to enhance the fringing field that account for the
radiation. However in some cases, other performance requirements may dictate the
use of dielectric material whose dielectric constants can be greater for εr > 4.
2.2
The Advantages & Disadvantages of Microstrip Antenna
Microstrip antennas have several advantages compared to the conventional
microwave antennas, and many applications cover the broad frequency range from
∼100 MHz to ∼100 GHz. Below are some advantages of them compared to the
conventional antennas [3].
5
-
Light weight, low volume, thin profile configurations
-
Low fabrication cost, readily amenable to mass production
-
Both linear and circular polarization are possible with simple feed
-
Dual frequency and dual polarization antennas can be easily made
-
No cavity backing is required
-
Easily can be integrated with microwave integrated circuits
-
Feed
lines
and
matching
networks
can
be
fabricated
simultaneously with the antenna structure
On the other side, there are also some limitations compared to the
conventional antennas as follow.
-
Narrow bandwidth and associated tolerance problems
-
lower gain (∼6 dB)
-
Large ohmic loss in the feed structure of arrays
-
Most microstrip antennas radiate into half-space
-
Complex feed structures required for high performance arrays
-
Polarization purity is difficult to achieve
-
Poor end-fire radiator, except tapered slot antenna
-
Extraneous radiation from feed and junctions
-
Lower power handling capability
-
Excitation of surface waves
-
Microstrip antenna fabricated on a substrate with high dielectric
constant are strongly preferred for easy integration with MMIC
RF front-end circuitry. However, use of high dielectric constant
substrate leads to poor efficiency and narrow bandwidth.
However many studies have been made to overcome the above limitations.
6
2.3
Radiation Mechanism
The radiated field from a microstrip antenna can be determined from either
the field distribution between the patch and the ground plane or the surface current
distribution on the patch [4].
A microstrip antenna that is fed from a source is considered. At microwave
frequencies, the transported energy is regarded as electromagnetic waves at
microwave frequencies, rather than current or voltage as in the conventional
electronics. A charge distribution will be established on the upper and lower surfaces
of the patch as well as on the ground plane, much as in the capacitor. The patch is
half a wave long so that the distribution at an arbitrary point in time is positive on
one side and negative on the other side of the patch as shown in below figure 2.2.
The repulsive force between charges of the same sign tends to push some of the
charges around the patch edge. This movement creates the current densities Jb and Jt.
The currents create a magnetic field tangential to the patch edges.
Figure 2.2
Charge distribution and current density on cavity model of a λ/2
mirostrip antenna
7
The attractive force between charges of different sign on the patch and the
ground plane is dominant though and the magnetic field is weak. Here approximation
is made that the tangential magnetic field is zero and place a magnetic wall around
the periphery of the patch. Moreover, since the substrate is thin compared to the
wavelength, the variations of the electric field over height can be considered to be
constant, means it is normal to the patch. This creates a cavity model situation with
electric field at top and bottom, and magnetic field wall along the edge. The
sidewalls can be considered as narrow apertures through which radiation takes place.
Using the Huygen [4] field equivalence principle, the microstrip patch can be
represented by a current density Jt at the top surface to account for patch
metallization. The four side slots are represented by equivalent current densities Js
and Ms, corresponding to the electric and magnetic fields Ea and Ha in the slots,
defined as,
→
∧
→
Js = n× Ha
→
∧
(2.1)
→
Ms = − n× Ea
(2.2)
As Jt is very much smaller than Jb in thin substrates, it is set to zero to indicate
negligible radiation from the patch current. Also the tangential magnetic fields are set
to zero, means that the current densities Js are zero. The only non-zero current
density is the equivalent magnetic current density Ms along the periphery of the patch.
The presence of the ground plane is taken into account via the image method which
doubles the Ms. Here, the radiation can now be ascribed to four ribbons of magnetic
current radiating into the free space and the new current density is given by
→
∧
→
Ms = −2 n× Ea
(2.3)
where the slot electric field in the dominant mode is defined as
→
∧
Ea = z E 0
(2.5)
8
for the slots of length W and height h. For the other two slots of length L and height h,
→
∧
E a = − z E 0 sin(πx / L)
(2.6)
As shown in below figure 2.3, each slot radiates the same field as a magnetic
dipole with current density Ms. The slots along the x-axis have equal and opposite
current distribution so the radiation from them can be set to zero. Hence the radiation
from the patch can be described in terms of two vertical slots. Vertical slots are
difficult to analyze and they are replaced with two horizontal slots. This model is
easy to understand and is the basis for many analytical models.
Figure 2.3
2.4
Electric field and magnetic current distribution
Transversal Magnetics Modes
In the transversal magnetics modes or TM modes means that only three field
components are considered. The three fields are electric field in the z direction that is
perpendicular to the ground plane and x-y direction that are parallel to the ground
plane. In general the modes are designated as TMxyz. The z is mostly omitted since
the electric field variation is considered negligible in the z-axis. Hence TMxy remains
with x and y the field variation in x and y direction. The field variation in the y
9
direction (impedance width direction) is negligible, thus y is 0. Lastly the field has
one minimum-to-maximum variation in the x direction (resonance length direction),
thus x is 1 in a fundamental case. Hence the notation is TM10.
2.5
Antenna polarization type
Basically an antenna is a transducer that converts the radio frequency electric
current to the electromagnetic waves that are then radiated into space. The electric
field or “E” plane determines the polarization or orientation of the radio wave. In
most communication applications, antenna normally uses either linear or circular
polarization.
A linearly polarized antenna radiates wholly in one plane containing the
direction of propagation. In linear polarization, the antenna is said to be vertically
polarized when the electric field is perpendicular to the earth’s surface. If the electric
filed is parallel to the earth’s surface, then the antenna is said to be horizontally
polarized.
In a circular polarized antenna, the plane of polarization rotates in a circle
making one complete revolution during one period of the wave. Right-hand-circular
(RHC) is when the rotation is clockwise looking in the direction of propagation while
the left-hand-circular is when the rotation is counterclockwise looking in the
direction of propagation. Main advantage of circular polarization is that regardless of
the receiver orientation, it will always receive a component of the signal due to the
incoming signal wave has an angular variation.
10
Figure 2.4
A linearly (vertically) polarized wave
(a) Linear polarization
Figure 2.5
(b) circular polarization
Linear and circular polarization for microstrip antenna
Knowing the polarization type of the antenna is important in order to
maximize its performance. Maximum signal strength between two antennas occurs
when both are using identical polarization. On line-of-sight (LOS) path, in a linearly
polarized system, a misalignment of polarization of 45˚ will degrade the signal up to
3 dB and a 90˚ misalignment can cause an attenuation of 20 dB or more. In a
circular polarized system, an additional loss of 20 dB will be incurred if both
antennas do not have the same polarization.
Cross polarization is another aspect need to be considered. Cross polarization
occurs when unwanted radiation is present from a polarization which is different
from the polarization in which the antenna was intended to radiate. For example, a
vertical antenna may radiate some horizontal polarization and vice versa.
11
2.6
Feeding techniques
Selection of the feeding technique is based on some factors.
The most
important consideration is the efficient transfer of power between the radiating
structure and the feed structure that is the impedance matching between them.
Basically they can be classified into two categories; the contacting and noncontacting method.
In the contacting method, the RF source is fed directly to the radiating patch
using a connecting element such as a microstrip line. In the non-contacting method,
electromagnetic field coupling is done to transfer power between the microstrip line
and the radiating patch. Below are some most popular feeding technique.
2.6.1 Coaxial probe feeding
Coaxial probe feeding is the most common feeding technique for the
microstrip antenna. As shown below in figure 2.6, the inner conductor of the coaxial
connector is extended through the dielectric substrate and directly attached to the
patch while the outer conductor is connected to the ground plane. Coaxial feeding
has several advantages such as easiness in term of fabrication process, flexibility to
be located at any desired location at the patch in or order to get matched input
impedance and low spurious radiation. However in case of application for the
thicker dielectric substrate, longer probe length makes the input impedance more
inductive that leads to the matching problem.
12
Figure 2.6
Coaxial probe feeding
2.6.2 Microstrip coplanar feeding
In this kind of feeding technique, a conducting strip line is directly connected
to the edge of the patch to create a planar structure. The width and the inset position
of the strip line can be optimized to match the input impedance without the need for
any additional matching element. This provides an ease in term of fabrication,
modeling and the impedance matching. However as the thickness of the dielectric
substrate is increased, surface waves and spurious feed radiation also increases,
creating a limit to the bandwidth of the antenna.
Figure 2.7
Microstrip coplanar feeding
13
2.6.3 Aperture coupled feeding
The radiating patch and the microstrip feed line are separated by the ground
plane as show below in figure 2.8. Coupling between the patch and the feed line is
made through a slot or an aperture in the ground plane. The coupling aperture is
normally centered under the patch, leading to lower cross-polarization due to
symmetry of the configuration. The shape, size, and the location of the aperture
determine the amount of coupling from the feed line to the patch. Spurious radiation
is minimized due to separation between the patch and the feed line by the ground
plane. Generally, a high dielectric material is used for the bottom substrate and a
thick, low dielectric material is used for the top substrate to optimize the radiation
from the patch. Difficulty in fabrication due to multiple layers which also increase
the antenna thickness is the major advantage of this feeding technique.
Figure 2.8
Aperture coupled feeding
2.6.4 Proximity coupled feeding
As shown in below Figure 2.9, the feed line is sandwiched between two
dielectric substrates and the radiating patch is on top of the upper substrate. The
14
advantages of this kind of feeding technique are elimination of spurious feed
radiation and high bandwidth due to overall increase in the thickness of the
microstrip patch antenna. Impedance matching is achieved by controlling the length
of the feed line and the width-to-line ratio of the patch. Difficulty in fabrication due
to the need of proper alignment between the two substrates is the main disadvantage
of this technique.
Figure 2.9
2.7
Proximity coupled feeding
Analysis Methods
Analysis methods for the microstrip antenna can basically be divided into two
groups; methods that are based on the equivalent magnetic current distribution
around the patch edges and also methods that are based on the electric current
distribution on the patch conductor and the ground plane. For this kind of analysis
methods, the radiation from the microstrip antenna is calculated from the equivalent
magnetic current distribution around the periphery of the radiating patch, which is
obtained from the corresponding voltage distribution. Means the analysis problem is
concentrated on finding the edge voltage distribution for a given excitation and for a
specified mode. Two main methods based on this kind of analysis are transmission
line model and the cavity model.
15
For the electric current distribution based method, the Method of Moments
(MoM) is the most common. This method is considered as a full wave model which
includes primarily integral equations or Moment Method.
In overall, the transmission line model is the simplest of all and it gives good
physical insight but it is less accurate. The cavity model is more accurate and gives
good physical insight but is complex in nature. The full wave models such as the
Method of Moments is extremely accurate, versatile and can treat single elements,
finite and infinite arrays, stacked elements, arbitrary shaped elements and coupling.
These characters give less insight as compared to the two models mentioned
previously and it is far more complex in nature. In the next sections, the three
methods mentioned above are discussed briefly.
2.7.1 Transmission line model
This model is considered as the simplest and normally used in understanding
the basic performance of the microstrip antenna.
Originally this model was
developed for rectangular patch but has been extended for generalized patch shapes.
In this modeling the microstrip antenna is represented by two slots of width W and
height h, separated by a transmission line with length L. The microstrip is a nonhomogeneous line of two dielectrics, typically the substrate and air.
Looking at the figure 2.10 shown below, most of the electric field lines reside
in the substrate and parts of some lines are in the air. For that, this transmission line
cannot support the pure transverse-electric-magnetic (TEM) mode of transmission
due to the fact that the phase velocities would be different in the air and the substrate.
Instead, the dominant mode of the propagation would be the quasi-TEM mode,
16
means that the effective dielectric constant εreff must be obtained in order to account
for the fringing and the wave propagation in the line.
(b)
(a)
Figure 2.10
(a) microstrip line (b) Electric field lines
The value of εreff is slightly smaller than εr since the fringing fields around
the periphery of the patch are not confined in the dielectric substrate, but also spread
in the as shown in figure 2.10(b). Based on Balanis [5], the expression for εreff is
given by
εreff
h
εr + 1 εr − 1 
1 + 12 
=
+

2
2 
W
−
1
2
(2.7)
where εreff : effective dielectric costant
εr : dielectric constant of substrate
h : height of dielectric substrate
W : width of the patch
Below figure 2.11 shows a rectangular microstrip patch antenna of length L,
width W layered on a substrate with height of h. The coordinate axis is selected such
that the length is along x direction, width is along the y direction and the height is
along the z direction.
17
Figure 2.11
Microstrip patch antenna
In order to operate in the fundamental TM10 mode, the length of the patch
must be slightly less than λ/2 where λ is the wavelength in the dielectric medium and
is equal to λ / εreff where λ is the free space wavelength. The TM10 implies that
the field varies one λ / 2 cycle along the length and there is no variation along the
width of the patch.
In figure 2.12(a) shown below, the microstrip patch antenna is represented by
two slots, separated by a transmission line of length L and open circuited at both the
ends. Along the width of the patch, the voltage is maximum and the current is
minimum due to the open ends. The fields at edges can be resolved into normal and
tangential components with respect to the ground plane.
It is seen from figure 2.12(b), that the normal components of the electric field
at the two edges along the width are in opposite directions and thus out of phase
since the patch is λ / 2 long and hence they cancel each other in the broadside
direction. The tangential components in figure 2.12(b) which are in phase, means that
the resulting field combines to give maximum radiated field normal to the surface of
the structure. Hence the edges along the width can be represented as two radiating
slots, which are λ / 2 apart and excited in phase and radiating in the half space above
the ground plane.
18
(a)
Figure 2.12
(b)
(a) Top view of patch antenna (b) Side view of patch antenna
The fringing fields along the width can be modeled as radiating slots and
electrically the patch of the microstrip antenna looks greater than its physical
dimensions. The dimension of the patch along its length have now been extended on
each end by a distance ∆L , which is given empirically by [3] as,
∆L = 0.412h
(εreff + 0.3) W
(εreff

+ 0.264 
h

W


− 0.258) + 0.8 
h


(2.8)
The effective length of the patch Leff now becomes,
Leff = L + 2∆L
(2.9)
For a given resonance frequency fo, the effective length is given by [3] as,
Leff =
c
(2.10)
2 f εreff
For a rectangular microstrip patch antenna, the resonance frequency for any TMmn
mode is given by [3] as,
1
f =
c
2 εreff
 m  2  n  2  2
  +   
 L   W  
(2.11)
19
where m and n are modes along L and W respectively. For efficient radiation, the
width W is given by [3] as,
c
W =
2 f
(εr + 1)
(2.12)
2
2.7.2 Cavity model
The transmission line model discussed previously, even it is easy to use, it
has some disadvantages. Specifically it is useful for patches of rectangular design
and it ignores field variations along the radiating edges. These advantages can be
overcome by using the cavity model.
In this model, the interior region of the dielectric substrate is modeled as a
cavity bounded by electric walls on the top and bottom. The basis of this assumption
is the following observations for thin substrates where h << λ.
-
Since the substrate is thin, the fields in the interior region do not vary
much in the z direction, i.e. normal to the patch.
-
The electric field is z directed only, and the magnetic field has only the
transverse components Hx and Hy in the region bounded by the patch
metallization and the ground plane. This observation provides for the
electric walls at the top and the bottom.
From the below figure 2.13, when the microstrip patch is provided with
power, a charge distribution is seen on the upper and lower surfaces of the patch and
20
at the bottom of the ground plane. This charge distribution is controlled by two
mechanisms; an attractive mechanism and a repulsive mechanism. The attractive
mechanism is between the opposite charges on the bottom side of the patch and the
ground plane, which helps in keeping the charge concentration intact at the bottom of
the patch. The repulsive mechanism is between the same charges on the bottom
surface of the patch, which causes pushing of some charges from the bottom, to the
top of the patch. As a result of this charges movement, currents flow at the top and
bottom surface of the patch.
Figure 2.13
Charge distribution and current density on the microstrip patch
antenna
The cavity model assumes that the height to width ratio (height of substrate
and width of the patch) is very small and as a result of this, the attractive mechanism
dominates and causes most of the charge concentration and the current to be at the
below of the patch surface. Much less current would flow on the top surface of the
patch and as the height to width ratio decreases, the current on the top of the patch
would be almost equal to zero, which would not allow the creation of any tangential
magnetic field components to the patch edges. Hence, the four sidewalls could be
modeled as perfectly magnetic conducting surfaces. This implies that the magnetic
fields and electric fields distribution beneath the patch would not be disturbed.
21
However in practice, a finite width to height ratio would be there and this
would not make the tangential magnetic field to be completely zero, but they are
being very small, the sidewalls could be approximated to be perfectly magnetic
conducting.
Since the walls of the cavity, as well as the material within it are lossless, the
cavity would not radiate and its input impedance would be purely reactive. Hence, in
order to account for radiation and a loss mechanism, one must introduce a radiation
resistance Rr and a loss resistance RL. A lossy cavity would now represent an antenna
and the loss is taken into account by the effective loss tangent δeff which is given by
[3] as,
δeff =
1
QT
(2.13)
QT is the total antenna quality factor and has been expressed in the form,
1
1
1
1
=
+
+
QT Qd Qc Qr
(2.14)
Qd represents the quality factor of the dielectric and is given as,
Qd =
ω rWT
1
=
Pd
tan δ
(2.15)
where ωr is the angular resonant frequency
WT is the total energy stored in the patch at resonance
Pd is the dielectric loss
tan δ is the loss tangent of the dielectric.
Qc represents the quality factor of the conductor and is given as,
Qc =
ωrWT h
=
Pc
∆
(2.16)
22
where Pc is the conductor loss
∆ is the skin depth of the conductor
h is the height of the substrate.
Qr represents the quality factor for radiation and is given as,
Qr =
ωrWT
Pr
(2.17)
where Pr is the power radiated from the patch.
Then by substituting above equation (2.14), (2.15), (2.16) and (2.17) into equation
(2.13),
δeff = tan δ +
P
∆
+ r
h ωrWT
(2.18)
which describes the total effective loss tangent for the microstrip patch antenna.
2.7.3 Method of Moments (MoM)
Method of Moments provides the full wave analysis for the microstrip patch
antenna. Using this method, the surface current is used to model the microstrip patch
and the volume polarization currents are used to model the fields in the dielectric
slab.
It has been shown by Newman and Tulyathan [6] how the integral equation is
obtained for these unknown currents and using the Method of Moments, these
electric field integral equations are converted into matrix equations which can then
be solved by various techniques of algebra to provide the result.
23
Overview of Moment Method described by Harrington [7] is given as follow.
The basic form of the equation to be solved by the Method of Moments is,
F (g) = h
(2.19)
where F is a known linear operator
g is an unknown function
h is the source or an excitation function.
The objective here is to find g, when F and h are known. The unknown function g
can be expanded as a linear combination of N terms to give,
N
g = ∑ a n g n = a1 g1 + a 2 g 2 + ...a N g N
(2.20)
n =1
where an is an unknown constant
gn is a known function normally called as a basis or expansion function.
By substituting equation (2.10) in (2.19) and using the linear property of the
operator F, below equation is obtained.
N
∑a
n =1
n
F(gn ) = h
(2.21)
The basis function gn must be selected in such a way, that each F(gn) in the above
equation can be calculated. The unknown constants an cannot be determined directly
because there are N unknowns, but only one equation. One method of finding these
constants is the method of weighted residuals. In this method, a set of trial solutions
is established with one or more variable parameters. The residuals are a measure of
the difference between the trial solution and the true solution. The variable
parameters are selected in a way which guarantees a best fit of the trial functions
based on the minimization of the residuals. This is done by defining a set of N
weighting or testing function {wm} = w1, w2, …wN in the domain of the operator F.
Taking the inner product of these functions, equation (3.21) becomes,
24
N
∑a
n =1
n
wm , F ( g n ) = wm , h
(2.22)
where m = 1, 2, …N. By writing in matrix form,
[Fmn ][a n ] = [hm ]
(2.23)
where
 w1 , F ( g1 ) w1 , F ( g 2 ) ........ 


w2 , F ( g 2 ) w2 , F ( g 2 ) ........

[Fmn ] = 





a1 
a 
 2
[a n ] = a3 
 
 
a N 
 
 w1 , h

 w2 , h
[hm ] =  w3 , h


 w N , h








The unknown constants an can now be found using algebraic techniques such
as LU decomposition of Gaussian elimination. It must be remembered that the
weighting functions must be selected appropriately so that elements of {wm} are not
only linearly independent but they also minimize the computations required to
evaluate the inner product. One such choice of the weighting functions may to let the
weighting and the basis function be the same, that is wn = gn.
From the antenna point of view, the electric field integral equation can be
written as,
E = fe( J )
where E is the known incident electric field
J is the unknown induced current
fe is the linear operator
(2.24)
25
The first step in the moment of method solution process would be to expand J
as a finite sum of basis function given as,
M
J = ∑ J i bi
(2.25)
i =1
where bi is the basis function
Ji is an unknown coefficient
The second step is to define a set of M linearly independent weighting
functions, wj. Taking the inner product on both sides and substituting equation (2.25)
in (2.24),
M
w j , E = ∑ w j , f e ( J i , bi )
(2.26)
i =1
where j = 1, 2, …M. Writing in the matrix form,
[Z ][J ] = [E ]
ij
j
(2.27)
where Zij = w j , f e (bi )
Ej = w j , H
J is the current vector containing the unknown quantities.
The vector E contains the known incident field quantities and the terms of Z
matrix are functions of geometry. The unknown coefficients of the induced current
are the terms of the J vector. Using any of the algebraic schemes mentioned earlier,
these equations can be solved to give the current and then the other parameters such
as the scattered electric and magnetic fields can be calculated directly from the
induced currents.
26
2.8
Impedance bandwidth of the antenna
The impedance bandwidth of a microstrip antenna is defined as the frequency
range over which it is match to a specified standard. In case of microstrip antenna, it
is proportional to its quality factor Q and given by [3] as
BW =
VSWR − 1
(2.28)
Q VSWR
The expression for approximately calculating the percentage bandwidth of the
rectangular patch microstrip antenna in terms of patch dimensions and substrates
parameters is given as follow [3].
% BW =
where A = 180 for
h
λo εr
h
λo εr
λo εr
W
L
(2.29)
≤ 0.045
A = 200 for 0.045 ≤
A = 220 for
Ah
h
≤ 0.075
λo εr
≥ 0.075
where h is the substrate thickness
λo is the wavelength in the substrate
εr is the dielectric constant of substrate
W, L are the width and length of patch dimension
In case of circular patch microstrip antenna, L = 2a (a is the radius of the patch), and
W= πa/2.
27
2.9
Effect of substrate parameters on bandwidth
Based on equation (2.28) in section 2.8, the impedance bandwidth of a patch
antenna varies inversely as Q of the patch antenna. Therefore, substrate parameters
such as dielectric constant εr and thickness h can be varied to obtain different Q thus
increasing the impedance bandwidth. Q (quality factor) of a resonator is defined as
follow.
Q=
Energy _ stored
Power _ lost
(2.30)
Below graphs [3] in Figure 2.14 and 2.15 show the relation between quality
factor Q with the substrate’s dielectric constant εr and thickness h for a rectangular
patch microstrip antenna. By assuming that the graphs are applicable for the case of
circular patch microstrip antenna, it can be concluded to that get a good impedance
bandwidth, the use of thick, low dielectric constant substrate should be considered.
Figure 2.14
Variation of radiation Q for a rectangular patch antenna as a function
of substrate dielectric constant; h = 1.59mm, W = 0.9L, f = 3 GHz
28
Figure 2.15
Variation of radiation Q for a rectangular patch antenna as a function
of substrate thickness; εr = 2.2, W = 0.9L, f = 3 Ghz
2.10
Bandwidth enhancement technique
As a technique for the purpose to increase the impedance bandwidth, two
techniques, stacked patches and coplanar parasitic patches are proposed.
2.10.1 Stacked patches configuration
As a technique or the purpose of broadbanding (to increase the impedance
bandwidth), two or more patches on different layer of the dielectric substrates are
stacked
on
each
other.
This
kind
of
configuration
is
categorized
as
electromagnetically (proximity) coupled microstrip antenna.
The basic configuration of electromagnetically coupled microstrip antenna is
shown below.
29
Figure 2.16
Multilayer electromagnetically (proximity) coupled stacked patch
microstrip antenna
The bottom patch is fed with a coaxial line, and the top parasitic patch is
excited due to the electromagnetic coupling with the bottom patch. The patches can
be fabricated on different substrates, and air gap or foam material can be placed
between these layers to create height thus increasing the bandwidth. Here, the
patches dimensions are optimized so that the resonance frequencies of the two
patches are closed to each other to yield a broad bandwidth. It was studied that the
electromagnetically coupling microstrip antenna with two or three layers of patches
provides an impedance bandwidth of up to 20 % for VSWR ≤ 2 [1].
The increase in the bandwidth is obtained due to the increase in the overall
height of the antenna and also a decrease in the effective dielectric constant in the
case where an air or foam substrate is inserted in between the two patches.
30
2.10.2 Coplanar parasitic patches
As mentioned in the previous section, the bandwidth of the microstrip
antenna improves with an increase in substrate thickness. However an increase in
thickness will cause other problems such as surface wave propagation and an
increase in probe inductance, causing a mismatch in the input impedance. In this
technique, parasitic patch is placed in coplanar close to the feeding patch and it gets
exited through the coupling with fringing field between the two patches. Same as the
previous bandwidth enhancement technique, the patches dimension is tuned so that
the resonance frequencies of these two patches are close to each other to obtain a
broad bandwidth.
(a)
Figure 2.17
(b)
(a) typical example of coplanar parasitic configuration (b) individual
response and overall response of parasitic coupled configuration
31
2.11 Application of microstrip antenna
Below table 2.1 summarize the typical application of microstrip antenna. The
wide range of application from transportation, communication to biomedical can be
seen.
System
Application
1. Aircraft and ship antennas
Communication and
navigation,altimeters, blind landing
systems
2. Missiles
Radar, proximity fuses and telemetry
3. Satellite communications
Domestic direct broadcast TV,
vehicle-based antennas,
communication
4. Mobile radio
Pagers and handphones, man pack
systems, mobile vehicle
5. Remote sensing
Large, lightweight apertures
6. Biomedical
Applicators in microwave
hyperthermia
7. Others
Intruder alarms, personal
communication etc
Table 2.1
Typical application of microstrip antenna
CHAPTER 3
THE DESIGN AND CONFIGURATION
Before the validation process of design, it is important to identify the
fundamental aspects, specification and parameter of the proposed antenna
configuration. This is important to make sure that the simulation and fabrication
process is carried out correctly.
3.1
Design Methodology
During the progress of the project, the design methodology used will be as
shown in below figure 3.1. The project starts with the study and literature review
related to get the fundamental knowledge about microstrip antenna.
Then the
configuration to be used for the antenna as well as the related basic parameters and
specification is set as desired. The validation starts with the simulation process by
using the Microwave Office software in order to confirm that the set parameters can
produce a result as desired. Finally, actual fabrication and field measurement is
carried out and comparison with the simulation result is done.
33
Figure 3.1
3.2
Methodology flowchart
Basic configuration and parameter of the antenna
In this section, the detail configuration used in the proposed wideband
microstrip antenna will be explained.
3.2.1 Patch shape
For this project, a circular shape patch is chosen. A circular patch microstrip
antenna is another widely used configuration other that the most popular rectangular
patch microstrip antenna. Basically it offers a similar performance as the rectangular
geometries. Using the circular shape type, the overall size tends to be slightly smaller
than the rectangular type shape. Furthermore, it can be easily modified to produce a
range of impedance values, radiation patterns and frequencies of operation.
34
Figure 3.2
Circular microstrip patch antenna with coaxial probe feeding
3.2.2 Feeding technique : Coaxial probe feeding
Feeding technique using a coaxial probe feeding is used. Using this
technique, the inner conductor of the coax is attached to the radiating patch while the
outer conductor is connected to the ground plane. This feeding method is chosen
mainly due to its easy fabrication process. Other advantages of coaxial feeding are
its robust nature, can be placed at any desired location for input impedance matching,
good isolation between the feeding network and the radiating element that
contributes to a good front to back ratios and minimum misalignment difficulties due
to direct contact between the probe and the patch.
3.2.3 Determine the operating frequency
The proposed microstrip antenna is tuned to operate at 5 GHz (ISM band).
The material to be used is FR4 with dielectric constant of 5.4 and thickness of 1.6
35
mm. In choosing the material the constraints in term of material cost and time is
considered. The resonance frequency is calculated from below equation [3].
( fr )110 =
1.8412vo
2πae εr
(3.1)
Here, vo : speed of light 3 x 108 ms-1
εr : dielectric constant of substrate material used
ae is effective area that considering the fringing field occurred along the
circumference of the circular patch and defined as follow [2].
ae = a{1 +
πa
2h
[ln( ) + 1.7726]} 1/2
πaεr
2h
(3.2)
Here, a is the actual radius of the circular patch. Using the above equation
(3.1) and (3.2), the actual radius of the patch can be determined.
3.2.4 Input impedance matching and Smith chart
The input impedance of an antenna is defined as the impedance presented by
an antenna at its terminal or the ratio of voltage to the current at the pair of terminals
or the ratio of the appropriate components of the electric to magnetic fields at a point.
From that, the impedance of the antenna can be written as
Zin = Rin + jXin
where Zin is the antenna impedance at the terminals
Rin is the antenna resistance at the terminals
(3.3)
36
Xin is the antenna reactance at the terminals
The imaginary part, Xin of the input impedance represents the power stored in
the near field of the antenna. The resistive part, Rin of the input impedance consists of
two components, the radiation resistance Rr and the loss resistance RL. The power
associated with the radiation resistance is the power actually radiated by the antenna,
while the power dissipated in the loss resistance is lost as heat in the antenna itself
due to dielectric or conducting losses.
Figure 3.3
An equivalent circuit of transmitting antenna
The Smith chart is a polar plot of the complex reflection coefficient, Γ that is
the ratio between the reflected voltage wave and the incident voltage wave. How
much of the incoming signal that is reflected depends on the match or mismatch
between the source impedance and the load impedance.
Below figure 3.4 is a smith chart plot, the circles are lines of constant
resistance, the smallest circle or the point to the right is the infinite resistance, a short,
and the outmost circle is the total reflection or an open circuit. The circle crossing in
37
the centre is the unit resistance or the characteristic impedance Zo. The lines crossing
the circles are the lines of constant reactance, with the horizontal line as zero. The
point of interest is where the line of zero reactance crosses the circle of unity
resistance where the system is matched.
A desirable design is one that causes the Smith chart diagram to do a loop
around the centre point as can be seen in below figure 3.4. This is created when
resonance occurs at the centre frequency. The distance from the loop to the centre of
the Smith chart indicates the reflection coefficient.
Figure 3.4
Smith chart diagram
3.2.5 Voltage Standing Wave Ratio (VSWR)
In order for the antenna to operate efficiently, maximum transfer of power
must take place between the transmitter and the antenna. Maximum power transfer is
achieved only when the input impedance of the antenna Zin is matched to that of the
transmitter Zs. According to the maximum power transfer theorem, maximum power
can be transferred only if the impedance of the transmitter is a complex conjugate of
38
the impedance of the antenna under consideration and vice versa. Means, the
condition for matching is
Zin = Zs ∗
(3.4)
where Zin = Rin + jXin
Zs = Rs + jXs as shown in the above figure 3.3.
If the condition for matching is not satisfied, then some of the power may be
reflected back and this leads to the creation of standing waves, which can be
characterized by a parameter called as the Voltage Standing Wave Ratio (VSWR).
By definition, VSWR is given by [8] as
VSWR =
1+ Γ
1− Γ
(3.5)
where Γ is given as
Γ=
Vr Zin − Zs
=
Vi Zin + Zs
(3.6)
where Γ is the reflection coefficient
Vr is the amplitude of the reflected wave
Vi is the amplitude of the incident wave
The VSWR is basically a measure of the impedance mismatch between the
transmitter and the antenna. The higher the VSWR, the greater is the mismatch. The
minimum VSWR which corresponds to a perfect match is unity.
39
3.2.6 Return Loss
The return loss (RL) is a parameter which indicates the amount of power that
is lost to the load and does not return as a reflection. As already known, waves are
reflected leading to the formation of standing waves, when the transmitter and
antenna impedance do not match. Hence the RL is a parameter similar to the VSWR
to indicate how well the matching between the transmitter and the antenna has taken
place. The RL is defined as
RL = −20 log 10 Γ
(3.7)
For perfect matching between the transmitter and the antenna, Γ = 0 and
RL = ∞ which means no power is reflected back, while Γ = 1 has an RL = 0 , which
implies that all the incident power is reflected. In practical application, a VSWR of 2
is acceptable, corresponds to an RL of -9.5 dB or 11% power reflection.
3.2.7 Determine the bandwidth based on return loss graph
In general, bandwidth of an antenna can be defined as the range of usable
frequencies within which performance of the antenna conforms to the specified
standard. For the broadband (wideband) antenna, the bandwidth is defined as the
ratio of the upper to lower frequencies of acceptable operation. The bandwidth of a
narrowband antenna can be defined as the percentage of the frequency difference
over the center frequency.
BWbroadband =
fH
=2
fL
 f − fL
BWnarrowband (% ) =  H
 f H × f L
(3.8)

 × 100

(3.9)
40
where fH is the upper frequency
fL is the lower frequency
An antenna is said to be broadband (wideband) if f H / f L = 2 . One method
that can be used to judge the performance of an antenna operating over the required
range of frequencies is by measuring its VSWR. A VSWR ≤ 2 (RL ≥ -9.5 dB)
Return loss dB
ensures good performance.
Figure 3.5
Determine the bandwidth from the return loss graph
3.2.8 Determine the probe feeding location
After selecting the radius of the circular patch, the next step is to determine
the feed point in order to get a good match between the input impedance of the patch
41
and the generator (source) impedance. Below equation [3] is used to determine the
feeding point along the radius.
xo = 0.5819[ Rin / Rr ] 1/2
(3.10)
where Rin : input resistance
Rr : radiation resistance
Figure 3.6
Determine the probe feeding location
3.2.9 Determine the gain of the antenna
Antenna gain is a parameter which is closely related to the directivity of the
antenna. Directivity is defined as how much an antenna concentrates energy in one
direction in preference to radiation in other directions. Means, if the antenna is
100 % efficient, then the directivity would be equal to the antenna gain and the
antenna would be an isotropic radiator. Since all antennas will radiate more in some
direction than in others, therefore the gain is the power that can be achieved in one
direction at the expense of the power lost in the others. The gain is always related to
42
the main lobe and is specified in the direction of maximum radiation unless indicated.
It is defined by [3] as,
G (θ , φ ) = ecd D(θ , φ )
(dBi)
(3.11)
where ecd : radiation efficiency
D : directivity
The expression for approximately calculating the directivity D is given by [3],
D ≅ 0.2W + 6.6 + 10 log(1.6 / εr ) dB
(3.12)
where W=πa/2 for the case of circular patch antenna.
3.2.10 Antenna radiation pattern
Radiation pattern from the antenna can be easily explained by using the
spherical coordinate system as shown in below figure 3.7. Here the z-axis is taken to
be the vertical direction and the xy-plane is horizontal. θ denotes the elevation angle
and φ denotes the azimuthal angle. The xz-plane is the elevation plane (φ = 0) or the
E-plane which is the plane containing the electric field vector and the direction of
maximum radiation. The xy-plane is the azimuthal plane (θ = π / 2 ) or the H-plane
which is the plane containing the magnetic field vector and the direction of
maximum radiation. The maximum radiation is normally obtained at these two
planes.
43
Figure 3.7
Spherical coordinate system for antenna radiation
3.2.11 Antenna Directivity
Antenna directivity can be defined as the ratio of the radiation intensity in a
given direction from the antenna to the radiation intensity averaged over all direction.
In other words, the directivity of a non-isotropic antenna is equal to the ratio of its
radiation intensity in a given direction, over that of an isotropic antenna.
D=
U 4πU
=
P
Ui
where D is the directivity of the antenna
U is the radiation intensity of the antenna
Ui is the radiation intensity of an isotropic source
P is the total power radiated
(3.13)
44
In some cases, the direction of the directivity is not specified. In this case, the
direction of the maximum radiation intensity is implied and the maximum directivity
is given as
Dmax =
U max 4πU max
Ui
P
(3.14)
where Dmax is the maximum directivity
Umax is the maximum radiation intensity
Directivity is a dimensionless quantity since it is the ratio of two radiation
intensities. Hence, it is generally expressed in dBi. The directivity of an antenna can
be easily estimated from the radiation pattern of the antenna. An antenna that has a
narrow main lobe would have better directivity, then one which has a broad main
lobe, means it is more directive.
3.2.12 Antenna Efficiency
Antenna efficiency is a parameter reflecting the amount of losses at the
terminals of the antenna and within the structure of the antenna. These losses
normally caused by the mismatch between the transmitter and the antenna and also
loss through conduction and dielectric I 2 R . Hence the total antenna efficiency can
be written as
et = e r e c e d
where et is the total antenna efficiency
er is reflection (mismatch) efficiency
(3.15)
45
ec is the conduction efficiency
ed is the dielectric efficiency
Since ec and ed are difficult to separate, they are lumped together to form ecd,
efficiency which is given as,
ecd = ec ed =
Rr
Rr + R L
(3.16)
where ecd is called as the antenna radiation efficiency and is defined as the ratio of
the power delivered to the radiation resistance Rr, to the power delivered to Rr and RL.
3.3
Method of Analysis
The method of analysis used is based on the electric current distribution on
the patch conductor and the ground plane known as Method of Moments (MoM).
The basic of this method already explained in section 2.7.3.
The software
Microwave Office used for the design and simulation purpose is using this MoM
analysis.
3.4
Effects of Finite Size Ground Plane
In the analysis and design of microstrip antenna, it is assumed that the size of
the ground plane is infinite. However in the actual application, only a finite size
ground plan is available. It has been shown that similar result for the infinite and
finite ground plane could be obtained if the size of the ground plane is greater than
46
the patch dimensions by approximately six times of the substrates thickness all
around the periphery [3].
CHAPTER 4
SIMULATION AND FABRICATION
4.1
Simulation process
Simulation by using the Microwave Office software was carried out to
evaluate the performance of the proposed antenna configuration. Several
configurations are proposed starting from the basic single patch design before the
actual implementation of the bandwidth enhancement techniques is done. By this
way the effect of the proposed bandwidth enhancement technique can be clearly seen.
Measurements done during the simulation are the return loss performance, smith
chart impedance variation, radiation pattern and gain of the antenna.
4.1.1 Design 1 : single patch microstrip antenna
First of all, simulation was started with the design that consists of a single
patch. The patch dimension was determined based on equation (3.1), set to resonate
at 3 different frequencies; 4.9 GHz, 5.0 GHz and 5.1 GHz. The position of the
48
coaxial was calculated to get the best input impedance matching. Dielectric substrate
material to be used was the FR4 with εr = 5.4, loss tangent δe = 0.02 and height h =
1.6 mm. Top of the structure was set to be an open air and to include the environment
factor in the simulation, the thickness of the air space above the top layer was set to
be 10 times of the overall structure height. In order to consider the finite ground
plane effect, the ground plane dimension was set to be greater than six times of the
substrates thickness all around the periphery Below figure 4.1 shows the structure of
Design 1.
(a)
Figure 4.1
(b)
Structure of Design 1 (a) 3D view (b) side view
Figure 4.2 below shows the return loss graph generated from the simulation. As
expected the three different dimension of patches resonated at the desired frequencies.
At this point, the bandwidth for VSWR ≤ 2 or equal to about 9.5 dB return loss is
around 9 ~ 10%. Maximum radiation gain is shown in below in figure 4.3 for the
case of Design 1(a) and 5.18 dB was obtained at maximum radiation angle.
49
Figure 4.2
Figure 4.3
Return loss simulation result for Design 1
Radiation pattern simulation result for Design 1
50
4.1.2 Design 2 : the use of stacked patch technique
In Design 2, two patches with different dimension were arranged in stack
configuration. The lower patch was set to resonate at 5G Hz using the structure
Design 1(a) created previously, while the top layer was set to resonate at 4.9 GHz by
using
the structure in Design 1(c). The air gap with height of 1 mm was created
between the two dielectric materials with same εr = 5.4. The purpose of this air gap is
to increase the overall height of the structure. Below figure 4.4 shows the proposed
structure.
(a)
Figure 4.4
(b)
Structure of Design 2 (a) 3D view (b) side view
From the below simulation result in figure 4.5, the stack configuration
enhanced the bandwidth up to 36.5 % as compared to 9 ~ 10 % for the single patch
configuration. Looking at the smith chart in figure 4.6 (a), the loop was created at
almost center of the chart, reflecting a resonance occurred between the two stacked
patches with an acceptable impedance matching. Maximum radiation gain is 6.1 dB
as shown in figure 4.6 (b).
51
Figure 4.5
Figure 4.6
pattern
Return loss simulation result for Design 2
(a)
(b)
Simulation result of Design 2 (a) Input impedance (b) radiation
52
4.1.3 Design 3 : the use of coplanar parasitic patch
In Design 3, bandwidth enhancement technique by using a coplanar parasitic
patch that also acts as a multi-resonator is done. Four patches were arranged 45˚ to
the x-axis around the center feeding patch and their dimension was set to resonate at
three different frequencies. Below figure 4.7 shows the configuration structure.
resonates at 4.6 GHz
resonates at 5.4 GHz
resonates at 4.75 GHz
(a)
Figure 4.7
(b)
Structure of Design 3 (a) top view (b) 3D view
The simulation result is shown in below figure 4.8. A balance stretched return
loss response was achieved from the patches combination, producing a
20 %
bandwidth. A multiple resonance can be seen from the smith chart at below figure
4.9(a), while the maximum radiation gain is 3.54 dB at figure 4.9(b).
53
VSWR ≤ 2 : 4.6 ~ 5.6 GHz
(20 %)
Figure 4.8
Return loss simulation result for Design 3
points of resonances
(a)
Figure 4.9
pattern
(b)
Simulation result of Design 3 (a) Input impedance (b) radiation
54
4.1.4 Design 4 : combination of stacked patch with coplanar parasitic patch
configuration (case 1)
In Design 4, combination of using both the stacked patches with coplanar
parasitic patches for the purpose of bandwidth enhancement was implemented. Both
the dielectric substrate layer are same er = 5.4 with a 1mm air gap between them. The
patches at the second dielectric layer were placed at the bottom of the substrate.
Dimension of the patches used are optimized to get the best bandwidth performance.
As shown in below figure 4.11, the return loss simulation result gave a
26.3 % well–balanced bandwidth. The resonance occurred can be seen from the
figure 4.12(a), with a small loop created at the center of the chart. Maximum
radiation gain of 6.73 dB was obtained as shown in figure 4.12(b).
1st layer
open air layer
dielectric layer 2
2nd layer
dielectric layer 1
1 mm air gap
(a)
Figure 4.10
(b)
Structure of Design 4 (a) top view (b) 3D view
55
VSWR ≤ 2 : 4.27 ~ 5.55 GHz
(26.3 %)
Figure 4.11
(a)
Figure 4.12
pattern
Return loss simulation result for Design 4
(b)
Simulation result of Design 4 (a) input impedance (b) radiation
56
4.1.5 Design 5 : combination of stacked patch with coplanar parasitic patch
configuration (case 2)
Design 5 shown in figure 4.13 is actually same as Design 4 except for the
upper patches that were placed at the top of the second dielectric layer. Means that
the distance from the first layer patches was higher compared to the Design 4
structure.
1st layer
open air layer
dielectric layer 2
2nd layer
1 mm air gap
(a)
Figure 4.13
dielectric layer 1
(b)
Structure of Design 5 (a) top view (b) 3D view
As shown in figure 4.14, the return loss simulation result gave a 30.7 % , 4 %
improvement compared to previous Design 4. The resonance occurred can be seen
from the figure 4.15(a), with small loop created almost at the center of the chart.
Maximum radiation gain of 6.68 dB was obtained, shown in figure 4.15 (b).
57
VSWR ≤ 2 : 4.27 ~ 5.8 GHz
(30.7 %)
Figure 4.14
(a)
Figure 4.15
pattern
Return loss simulation result for Design 5
(b)
Simulation result of Design 5 (a) input impedance (b) radiation
58
4.1.6 Design 6 : combination of stacked patch with coplanar parasitic patch
configuration (case 3)
A totally different configuration was designed in Design 6. Several different
dimensions of patches are arranged in size order to further stretch the bandwidth to
its maximum. Both the dielectric substrate layer are same er = 5.4. The patches at the
second dielectric layer were placed at the top of the substrate and a 1.6 mm air gap
was created between the dielectric substrates. Dimension of the patches used are
optimized to get the best bandwidth performance. The structure is shown in figure
4.16.
As shown in below figure 4.17, the result gave a further improved result
compared to the previous designs with a 34.7 % bandwidth. From figure 4.18 (a) and
(b), resonance can be seen produced at the center of the smith chart and a maximum
radiation gain of 6.26 db was obtained.
2nd layer
open air layer
1st layer
2nd dielectric layer
1st layer
1st dielectric
layer
1.6 mm air gap
(a)
Figure 4.16
(b)
Structure of Design 6 (a) top view (b) 3D view
59
VSWR ≤ 2 : 4.36 ~ 6.16
(34.7 %)
Figure 4.17
(a)
Figure 4.18
pattern
Return loss simulation result for Design 6
(b)
Simulation result of Design 6 (a) input impedance (b) radiation
60
4.1.7 Summary of simulation results
In the previous sections from section 4.1.1 to section 4.1.6, simulation for six
different configurations was carried out and as it can be said that the results almost
followed the theory as expected.
Bandwidth enhancement could be seen by using either the stack or coplanar
parasitic coupling but the dimensions of the patches need to be carefully determined
in order to get the maximum result. If the resonance frequency of every patch is set
too far with each other, the coupling could not be realized hence no effect could be
seen in term of bandwidth enhancement.
As comparison, stacked configuration technique gave a better bandwidth
enhancement as compared to coplanar parasitic technique. This might be caused by a
more fringing field coupling was realized in stack configuration as compared to
coplanar parasitic technique. This can be seen in the simulation result of Design 2
and Design 3.
Increasing the thickness of the overall structure also helped in increasing the
bandwidth where this was seen in the simulation of Design 4 and design 5 where
about 4% of improvement was obtained.
61
4.2 Fabrication process
As a validation of the results obtained from the simulation carried out, several
designs proposed in previous section were fabricated. For this purpose, Design 4 and
Design 5 were chosen.
The flow of the fabrication process started with the mask generation on the
transparency. Autocad software was used to draw the actual dimension of the
antenna based on the configuration in the simulation.
Figure 4.19
Mask generation on transparency
The transparency is then is attached to the FR4 material and place under the
UV light for about 50 ~ 90 seconds. Here the mask unprotected area is exposed to the
UV photo leaving the silicon dioxide layer on the top.
62
Figure 4.20
Figure 4.21
FR4 material used for fabrication
Process of exposing the FR4 material under UV light
Then silicon oxide etch process is carried out to remove the areas of silicon
dioxide unprotected by the photo-resist to remove the SiO2 and expose the silicon
underneath. This is done by soaking the FR4 material in the developer solution for
about 10 minutes as shown in figure 4.22.
Lastly to etch the silicon layer, the material is again soaked in the ferrite
chloride for half an hour as in figure 4.23. Figure 4.24 shows the desired copper
patch on the FR4 dielectric material after completing this process.
63
Figure 4.22 Soaking the FR4 material in the developer
Figure 4.23
Etching process using the ferrite chloride
(a)
Figure 4.24
(b)
(c)
Fabricated structure (a) first layer substrate (b) second layer substrate
(c) stacked structure of the antenna
64
Figure 4.24(c) shows the completed structure of the stacked patches antenna
with the SMA connector to be connected to the coaxial line. Form is used to support
the upper layer positioned on the lower layer. This is to create a 1mm air gap
between them.
4.3
Antenna Measurement
After fabrication process was finished, measurement was carried out to
evaluate the antenna performance and the result was compared with the simulation
result.
4.3.1 Return loss measurement
Figure 4.25
Return loss measurement using the microwave analyzer
65
Based on the return loss graph shown below in figure 4.26, the behavior of
the measurement result follows the simulation. In term of the bandwidth for VSWR
≤ 2, the measurement gives a 25.19 % bandwidth compared to 26.5 % through the
simulation. However a frequency shift can be seen from the graph, where the
measurement result moves to the right side of the simulation result. The difference is
mainly caused by the inaccuracy, defects and imperfectness occurred during the
fabrication process. During the simulation, the dimension plays with the figure of 0.1
mm (high precision) between the patches, which is very hard to achieve if the
fabrication process used is not precise enough.
Frequency vs Return Loss
0
Return loss (dB)
4
5
6
7
-10
-20
Simulation
-30
Measurement
-40
-50
Frequency (GHz)
Figure 4.26
Return loss measurement result for Design 4
Below graph in figure 4.27 shows the return loss measurement result for the
case of Design 5. Same as the result for Design 4, even the same behavior can be
seen between the simulation and the measurement result, the frequency position is
shifted to the right. The same cause is considered for this case too. The bandwidth for
the measurement is 32.2 % while the simulation result is at 30.7 %.
66
Frequency vs Return loss
0
Return loss (dB)
-5 4
5
6
7
8
-10
-15
simulation
-20
measurement
-25
-30
-35
Frequency (GHz)
Figure 4.27
Return loss measurement result for Design 5
4.3.2 Radiation pattern measurement
Measurement of the radiation pattern was carried out in the anechoic chamber.
A horn antenna was used as a transmitter, transmitting a signal at strength of 25 dBm.
The measured antenna acted as a receiver and its radiation pattern was measured at
5.5 GHz.
The results are shown in below figure 4.28 and figure 4.29. In general, a
balance radiation pattern was obtained along the -90˚ ~ 90˚ direction. For both
Design 4 and Design 5, the maximum value of around 20 dBm was achieved
between the co-polarization and the cross-polarization pattern.
Overall, from the two measurements carried out, the result seemed to be not
so convincing as compared to the simulation results. The main cause behind this was
the dimension precision level of the fabricated antenna that was very hard to achieve
67
same as the one in simulation. However, the results still follow the rules which it was
expected to show.
Radiation pattern E-plane
0
-40
magnitude (dBm)
-90
-10
10
60
Design 4
-20
-30
co-polar
-40
cross-polar
-50
-60
-70
-80
theta/degree
(a)
magnitude (dBm)
Radiation pattern H-plane
-90
-40
0
-10
-20
-30
-40
-50
-60
-70
-80
10
60
Design 4
co-polar
cross-polar
theta/degree
(b)
Figure 4.28
Radiation pattern measurement of Design 4 (a) E-plane (b) H-plane
68
magnitude (dBm)
Radiation pattern E-plane
-90
-40
0
-10
-20
-30
-40
-50
-60
-70
-80
10
60
Design 5
co-polar
cross-polar
theta/degree
(a)
Radiation pattern H-pane
0
magnitude (dBm)
-90
-40
-10
10
60
Design 5
-20
-30
co-polar
-40
cross-polar
-50
-60
-70
-80
theta/degree
(b)
Figure 4.29
Radiation pattern measurement of Design 5 (a) E-plane (b) H-plane
CHAPTER 5
CONCLUSIONS AND FUTURE WORKS
Simulation and actual hardware measurement was carried out in order to
study the use of the proximity coupled stacked configuration that combined with the
coplanar parasitic multiresonators in the microstrip antenna design. The purpose was
to enhance the bandwidth performance up to the wideband level.
The result showed that the bandwidth can be increased from the typical 8 ~
9 % up to 36 % and maintain the gain at around 6 dB. However this is still far from
the objective, considering the definition of wideband
f H = 2 f L , where fH is the
higher frequency and fL is the lower frequency for the VSWR ≤ 2 frequency range.
The current design has showed its limitation, where the bandwidth could not be
further stretched anymore even more patches with different dimensions was tested.
Further studies still need to be continued by using a different structure
configuration, the use of different dielectric substrate material as well as combination
of different substrate in one structure, the use of different feeding technique, the use
of different shape of patch with larger radiating area and the use of impedance
matching network to enhance the impedance bandwidth.
70
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