Integrated Cellular Transceivers: Challenging Traditional Test Philosophies
Edwin Lowery
Agilent Technologies
12401 Research Blvd, Suite 100 Austin, TX 78759
[email protected], (512)-257-5825
Abstract
In today's high volume manufacturing of cellular
transceivers, cost of test is paramount. With more and more
integration of mixed signal and RF components onto the same
die, traditional test philosophies are being challenged. Should
this highly integrated transceiver be considered as a series of
blocks, which need to be tested individually? Would it make
more sense to treat the whole die as a radio and test it
accordingly? In general Mixed Signal engineering techniques
treat the chip as a series of blocks, and RF engineering
techniques treat the chip as a radio. The optimal test flow lies
somewhere in between.
This paper will illustrate, with examples, the tradeoffs
between treating a system on a chip or SOC as a radio versus
a series of functional blocks. Both approaches have their
merits and this paper will give an example test list breakdown
for each case. Two real world block diagrams of cellular
receivers will be used as examples and the test breakdown
needed to cover functionality of all the blocks using both
philosophies will be discussed. Each approach will be
analyzed to determine its effectiveness and efficiency in final
test.
To illustrate the differences, two optimized test flows will
be presented which contain elements of both approaches and
the reasoning behind the tests chosen will be discussed,
providing the reader with more insight into writing efficient
system tests for radio transceivers. This paper will also
highlight the difference between characterization and final
test, as well as provide some insight into the differences
between mixed signal and RF test.
Introduction to Digital Radio Receivers
As cost pressures continue to rise on the integrated circuit
industry, the trend towards higher and higher integration
continues. In the RF world, this manifests itself as more and
more digital and mixed signal components showing up on
traditionally RFICs. The industry goal is to integrate as much
onto a single IC as possible.
In the consumer cellular market, it is well established that
digital radios can save a lot of battery power as well as
conserve precious and finite frequency bandwidth. Let’s start
by analyzing the contents of a generic digital radio for cellular
phone receivers (Figure1.)
Figure 1 Superhet Digital Radio Block Diagram
The entire spectrum available to the antenna arrives at the
Preselecting Filter. This filter limits the spectrum to a few
specific frequencies, namely the RX or receiver band of the
cell phone. Since this input channel has a very low
amplitude, it is then boosted in power by the Low Noise
Amplifier or LNA. This signal is then downconverted
through the first mixer, and then further filtered to ensure that
only one RX channel gets through. What remains is a single
RX channel, which needs to be demodulated. This single
channel is split into its in-phase and quadrature components
by using two additional mixers. The LO’s, or local
oscillators, for these mixers are offset by a 90 degree phase
separation. The baseband signals are then filtered another
time and digitized.
In this example, most of the components described are
traditional RF system blocks. The only ones that are
traditional mixed signal blocks are the baseband filter and the
ADCs. In fact, RF engineers typically get the signals directly
from the baseband filters and perform their analysis from
there.
In the semiconductor test world, things are a bit different.
The block diagram changes a bit, which has a huge impact on
the test list and test methodology. There is no preselecting
filter or baseband analog to digital converters (ADCs.)
Consider a real world example of a block diagram of a code
division multiple access (CDMA) receiver as shown in Figure
2.
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AMPS
820MHz
30KHz
FM
Figure 3 shows another example of a real world cellular
receiver. It is also a ZIF receiver, but this one is designed to
work in the GSM or global system for mobile
communications band.
Figure 2 CDMA/AMPS Receiver Block Diagram
The first thing that should be apparent is that this is a
slightly different architecture. In fact, this is a zero IF or ZIF
design. This means that there is a single LO and hence a
single down conversion used for the device. In this case, the
entire RX input band is incident on the first LNA, and then
taken off-chip for further band pass filtering. The signal is
then sent through an automatic gain control amplifier or
AGC. This AGC amplifies the signal so that it can be detected
by the demodulator circuit. Next the signal is directly down
converted so that its center frequency is at 0Hz (hence zero
IF.) The 90 degree phase shifter block from the LO causes
the incoming signal to be demodulated and broken down into
its corresponding I and Q components. Finally the signal is
sent through a final low pass filter and buffer before being
sent off chip.
A few new components are visible in this block diagram
as well. Notice there are two DAC’s at the I and Q buffers.
In ZIF architectures, converting a signal from RF to baseband
will inevitably have a DC component. If the component is
too high, the signal needed will not be detectable. These two
DAC’s will compensate for the DC component and make sure
the signal stays around 0V. Also notice that the AGC block
needs some sort of control to manage the level of
amplification for the receiver. These kinds of DAC blocks
are quite common for mixed signal test engineers, but may be
new for RF test engineers.
This device is a CDMA/AMPS receiver, which also
supports PCS.
Table 1 includes some of the basic
specifications for these standards.
Table 1 PCS, CDMA, AMPS Summary
RX
Frequency
Channel spacing
Modulation
PCS
1.9GHz
1.25MHz
QPSK
CDMA
820MHz
1.25MHz
QPSK
Figure 3 GSM/DCS Receiver Block Diagram
From a block diagram point of view, this is very similar to
the CDMA chip. It is also a ZIF receiver, except that it works
at two different bands. This is why it has two LNA’s before
the AGC. Each LNA is tuned for either the DCS or GSM
frequency band. Also, it is not necessary to have off chip
filtering as GSM adjacent channel filtering is less rigorous.
Note that this chip has a voltage controlled oscillator or VCO
integrated right onto the chip. This adds more complexity to
the chip test list. This device is described as a DCS,
GSM850, GSM and PCS compatible receiver. It is also
specified as being EDGE ready. Table 2 summarizes what
this means from a high level point of view.
DCS
GSM850
GSM
PCS
EDGE
Table 2 DCS/GSM/PCS/EDGE Summary
RX Frequency
Channel
Modulation
Spacing
1805-1880MHz
200KHz
GMSK
869-894MHz
200KHz
GMSK
925-960MHz
200KHz
GMSK
1930-1990MHz
200KHz
GMSK
Any Above
200KHz
8PSK
Mixed Signal vs. RF Test Engineering
As integration increases, it is becoming more and more
necessary for RF engineers to learn mixed signal test and for
mixed signal engineers to learn RF techniques. The main
difference between RF engineers and mixed signal engineers
is that RF engineers do a frequency conversion before their
digitizing. RF engineers also play more tricks with incident
and reflective waves by using reflectometers. For this
discussion though, the device under test or DUT acts as the
frequency translator, and the digitizing is done using
traditional mixed signal techniques. Therefore, all RF blocks
will be treated as cascaded RF elements, and the mixed signal
blocks will be tested with mixed signal techniques. For
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consistency, keep in mind that dBm for a 50 ohm system is
defined as:
dBm = 10log[((vout)2/50)/1mW]
This is an absolute measurement of signal level compared
to 1mW. All baseband power and gain measurements need to
be converted to this unit for consistency with other radio
specifications.
Production vs. Characterization
Traditionally, test engineers start with a very complete test
program, and then simply remove some of the tests when
production starts. As the product matures, additional tests are
removed until a final test flow is achieved. This process can
take years to achieve a final test flow. Looking at the
different modes of operation, there is a strong temptation to
test all permutations of types of modulation, frequencies, and
channels. There is also a great deal of pressure from the
designers to add additional tests to the test flow for their “pet”
module.
Test engineers should strongly resist the temptation to put
all these tests into the production test flow. Keep in mind that
each device should be characterized before going into
production, and characterization should be as complete as
possible. For production, however, the main goal is to
determine that the DUT is good and that it is functional. It is
not necessary to characterize every device that ships from the
test floor. However, understanding the need to minimize risk,
there is a very easy way to get the best of both worlds.
One approach could be to minimize the test flow at the
beginning of the ramp up, and characterize every Nth device
(see Figure 4) putting all results into a datalog.
Figure 4 Alternative Production Flow
This approach can get the best of both worlds. It allows a
much faster production test time, and still allows visibility
into overall device performance. This approach also has the
benefit of keeping track of production corners and also
keeping costs low. Defining P as production test time, and C
as characterization test time, overall test time will be:
Teff= {P*(N-1)+C}/N.
Based on this production flow approach, it becomes
extremely important to reduce the production tests needed to
show that the DUT is working and meets specifications. The
best way to do this is start by analyzing each block of the
DUT.
SOC Test Generation Block by Block Approach
A good approach to developing a test list for these devices
is to understand the kinds of measurements needed for each of
the blocks of the radio. Once all blocks are understood, the
tests are then cascaded together to come up with a complete
test list, which covers all of the individual blocks.
Amplifiers
When testing amplifiers, the primary measurements of
interest are:
•
Gain
•
Noise Figure
•
Impedance Match
•
Saturation/Compression
•
Flatness across channel
•
Intermodulation Distortion
To describe any amplifier’s performance, gain and
saturation, will show the block’s amplification and dynamic
range. The flatness across the channel will make sure that the
gain is constant across the channel frequency range. Since
each block in any radio design will add noise to the signal,
noise figure will determine exactly how much noise is added.
For DUT test, impedance match of internally connected
amplifiers is probably not necessary or even possible to
measure. It is assumed that if there were a problem with the
impedance match of an internal node, it would certainly show
up on one of the other measurements (gain, flatness, or noise
figure.)
The previous measurements show a variety of linear
effects. It is also crucial to measure non-linear effects on the
incident signals. Figure 5 shows the most popular non-linear
measurement, intermodulation distortion. The idea is that
when two signals are applied to the DUT, more than these
two signals appear at the output. F1 and F2 are applied very
close to each other in the frequency band. It turns out that
two intermodulation products will appear very close to the F1,
and F2 signals at 2*F1-F2 and 2*F2-F1.
For example if N=50, P=2.0 seconds and C = 60.0
seconds, Teff = 3.16 seconds. As the product matures, simply
increase N to further improve throughput. If N = 200, Teff
becomes 2.29 seconds.
This paper assumes that the characterization test flow is
pretty obvious to derive (although time consuming to code.)
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Figure 5 In Band Intermodulation Products
The good news for the test engineer is that some of these
concepts are easy to cascade. Gain and noise figure for an
amplifier coupled with a mixer, can be combined into one
measurement. In other words for both of these blocks
combined, one measurement can be made to qualify both
blocks. Once again, impedance match of an internal node is
not so crucial to measure in production as any impedance
match problems will show up in gain or noise figure.
The other measurements are graphically depicted in
Figure 7. The signal incident on the mixer is called the RF
signal at port 1. The LO is the local oscillator signal which is
incident at port 2. The IF or intermediate frequency is known
as the output port, which is visible at port 3.
RF to IF isolation is shown in blue and it is simply the
amount of unwanted RF signal which “bleeds through” to the
IF port. LO to IF isolation is shown in gray, and LO to RF
isolation is shown in pink.
An interesting property of these signals is that the
fundamentals (f1 and f2) will increase 1dB at the output for
every 1dB increase at the input. The intermodulation
products have a 3 to 1 growth. This means that if the
fundamentals are increased by 1dB on the input, the
intermodulation products will grow by 3dB. We can plot
these two lines on an input vs output chart. Finding where
these two lines intersect is called the third order intercept or
TOI, shown in Figure 6.
Figure 7 Mixer Isolation Paths
Mixer operation is simple as it is designed to take an input
RF signal and translate that up and down in the frequency
domain by the equation: FIF = FLO-FRF, and FIF=FRFFLO. In radio design, either the lower or upper IF bandwidth
is filtered out. (Figure 8)
Figure 6 Third Order Intercept
Mixers
In the ZIF radio block diagram, the next block of interest
is the mixer. In measuring mixers, the primary interest is in:
• Gain
• Noise Figure
• Impedance match
• RF to IF Isolation
• LO to IF Isolation
• LO to RF Isolation
Figure 8 Mixer Output Products
DUT verification for test engineers is simplified greatly
because most RF receivers are designed such that the LO-RF
and LO+RF are always much greater than the channel
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bandwidth. This means that there is no need to test for this
bleed through effect.
Other intermodulation products can, however, be
generated due to other frequencies and interactions, which
can indeed show up in the operating band of the radio. A
SINAD (signal to noise + distortion) test can be used to check
the entire operating bandwidth for these intermodulation
products.
Demodulators
The next block in the ZIF radio design is the demodulator
(Figure 9.) For the ZIF design, the demodulation occurs by
taking an LO frequency, and shifting it by 90 degrees. This
signal is then applied to two different mixers. The resultant
demodulation becomes highly dependent on extremely
accurate phase separation between the two mixers, and how
accurately the gain of the I and Q paths are matched. These
characteristics are described by:
• Phase Mismatch
• Amplitude Mismatch
Also, the gain of the I and Q signals must be identical.
This is measured by sending in a single tone and checking the
amplitude difference between the I and Q outputs. Figure 11
shows the graphical representation and the equation used to
make this measurement.
Figure 11 Amplitude Mismatch Plot
Baseband Filters
The next major functional block common to both ZIF
receivers is the baseband anti-aliasing filter. For the baseband
anti-aliasing filter the measurements of interest are:
• Amplitude ripple per channel
• Phase response per channel
• 3dB Rolloff
• Cumulated ISI
Figure 9 ZIF Demodulator
The theory behind how this works is based on the idea
that information can be sent over the RF link by
instantaneously varying the amplitude and phase of the signal.
For thorough testing of the device, phase mismatch of the
demodulator across the entire operating band of the channel
must be measured. In practice, a few points are measured and
linearity is assumed between the points (Figure 10.)
Figure 10 Phase Mismatch Plot
Amplitude ripple is measured by looking over the entire
operating channel and finding variation in the pass band.
This can be measured at the same time as amplitude mismatch
is made. The same idea applies to Phase response per
channel.
Additionally a major filter figure of merit is the frequency
point at which the amplitude drops by 3dB. This can also be
a time consuming search. One approach might be to make
several data point measurements and then do an interpolation
to find exactly where the 3dB point occurs. This 3dB point is
known as the operating bandwidth of the device.
Figure 12 Amplitude ripple and 3dB Rolloff
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Even ideal filters will affect the incident signals in some
way. Figure 13 shows the impulse response on a standard
LPF filter. If the signal is spread too far in the time domain,
inter symbol interference is created. This can cause signal
degradation.
Figure 13 LPF Impulse Response
Beyond the 3dB cutoff frequency, the ISI is the ultimate
goal of any baseband filter measurement. The test engineer
wishes to ensure that the baseband filter will not cause inter
symbol interference. If ISI is too high, the device will not be
able to demodulate the signal properly.
VCOs
For the examples used in this paper, the VCO block is
only present in GSM ZIF receiver. For VCO characterization,
major interests are:
• VCO output power,
• VCO Frequency Range
• Voltage vs. Frequency response
• Phase Noise
Figure 14
To measure phase noise, lock the VCO, and compare the
maximum output with the power over a narrow bandwidth at
a specific offset frequency. The purpose of this measurement
is to ensure that any received signal will stay within band.
This number is usually recorded with the measurement
bandwidth normalized as in dBc/Hz. See Figure 15.
The output power and frequency range, are
straightforward RF measurements. In order to test voltage vs.
frequency response, the VCO must be locked with a PLL
circuit. First the VCO is locked at the lower end of the band,
and the output frequency, and input voltage are measured.
Next the VCO is locked at the upper end of the band and
another frequency and voltage measurement is made. The
slope is then recorded as: (V1-V2)/(F1-F2).
Figure 15 Phase Noise Measurement
Overall DUT System Test List
As stated before, many of the tests that the RF blocks have
in common can be cascaded. By cascading all relevant tests,
an overall radio test list can be generated. Beginning with the
GSM radio receiver, table 3 provides a rather comprehensive
production test list.
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GSM band, or 914MHz. Remember, that as the test list is
being generated, make sure that the test frequencies show a
representative list of the operating bands of the device.
Table 3 GSM Comprehensive Test List
The list starts with the non-linear tests. The non-linearity
tests include TOI, and Noise Figure. In the case of the GSM
receiver, there will need to be two measurements for each of
these tests. One measurement will be for the DCS LNA path,
and the other for the GSM LNA path. Table 2 shows the
frequency bands of interest for GSM/DCS radios. As the
final test list gets generated, it is a good idea to use as many
corner cases of this frequency band as possible. For instance,
the NF measurement could be made at 869MHz, and
1805MHz. The TOI’s could then be measured at 960MHz
and 1990MHz. This provides coverage at various frequency
points in the operating band.
The next section, AGC linearity, will require two
measurements. One measurement is for minimum sensitivity,
and the other for maximum input power. This test will make
sure that the DUT is not in compression over the specified
dynamic range. This test could be done in the middle of the
The gain/balance tests will be the most involved. Recall
that in order for the demodulator to work properly, proper
phase and amplitude match across the operating channel must
be ensured. For this reason, three phase and amplitude
measurements must be made, and three corresponding SINAD
measurements must be made at high, med and low band. The
good news is that this can generally be done in six baseband
captures (three captures of both I and Q), but all three
quantities must be calculated and datalogged. Overall Gain
must also be checked for both the GSM and DCS LNA’s.
For the filter response, both the I and the Q channels must
be checked and at least three measurements must be made for
each in order to find exactly where the 3dB rolloff occurs.
This makes a total of 6 measurements
Since the GSM receiver has two DC offset correction
DAC’s two traditionally mixed signal tests of INL and DNL
will be needed in order to ensure that all bits are working
properly. INL stands for integral non-linearity while DNL
stands for differential non-linearity. These tests are well
defined and can be found in any mixed signal test text. In
order to perform theset tests, the part must have some sort of
test circuit to breakout the DAC output voltage. For the AGC
circuit one additional INL/DNL test will be needed.
Recall that this GSM receiver also has a VCO. This
implies that two measurements of both frequency and voltage
need to be made in order to calculate the VCO frequency
response slope. The VCO’s output power and phase noise
will also be needed, for a total of six different VCO
measurements.
Using the technique of block by block analysis, a
production test list has now been generated that checks the
major functionality of each block. Each block has been
activated and its major effects measured. The total number of
measurements needed for this GSM receiver is 32.
For the CDMA receiver, the main difference is that it has
an embedded LNA that must be tested. It does not have a
VCO so those tests can be omitted. The test summary can be
seen in Table 4.
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Table 4 CDMA Receiver Comprehensive Test List
the result is what is referred to as constellation plots. Figures
16 and 17 are constellation plots for GSM and EDGE
modulation respectively. GSM is a constant amplitude
modulation and appears as a circle with discrete points. Edge
is much more complex and appears as a series of dots on the
I-Q plane. EVM is the magnitude of error that the measured
signal has as compared to the ideal constellation plot. Since
this is a dynamic measurement, it contains both amplitude and
time components.
Figure 16 GSM Constellation Plot
For the non-linearity tests, the performance of the first
LNA by itself will be needed and then the overall
performance of the rest of the receive chain should be
measured. This works out to be 2 NF measurements and 2
TOI measurements.
For the gain/balance tests, only the receiver gain of the
chain from the output of the SAW filter will be needed, but
the rest of the list will be identical to the GSM test list.
Of course the CDMA receiver has no VCO on board, so
these tests will not be needed. However, the same techniques
as before should be employed to ensure that all major
frequencies in the 820MHz band and the 1900MHz band are
represented in the test list.
Once again, the entire DUT has been tested and each
block has been represented in the final test list. The end result
is 25 measurements needed to test this CDMA receiver.
SOC Test Breakdown A Radio Approach
In the previous example, signals used on the DUT were
limited to CW’s or discrete carrier waves at the input and
output of the device. In the real world, this is never the case.
Once the DUT is installed in a phone, it is only given
complex signals with coherent messages. Modulated signals
are measured with EVM.
EVM or error vector magnitude is a measure of
modulation quality. It is a measurement which quantifies how
several radio blocks act and interact. If a modulated signal is
applied to the receiver and a plot is made of the I and Q plane,
By using EVM, the DUT is no longer being treated as a
series of blocks. Now the DUT is being treated as a system.
The benefit here is that one test can replace several other
discrete tone tests.
Figure 17 EDGE Constellation Plot
In order for a demodulator to pass a complex
demodulation test, the phase and amplitude balance have to be
correct across the entire operating band. Also, there has to be
very little inter symbol interference (Figure 13.) What this
means is that by using EVM to test the demodulator, all of the
amplitude/phase imbalance tests, and all of the filter tests can
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be replaced with a single EVM measurement. This will
greatly reduce the test list. Of course, for the EVM
measurement to be accurate, very good test instrumentation
linearity is required, and extremely robust demodulation
algorithm must be available.
CDMA modulation uses QPSK or quadrature phase shift
keying modulation. Figure 18 shows QPSK modulation with
various filter slopes. This is intended to demonstrate the
effect of defective filters on the EVM constellation plot.
Table 6 CDMA Test List with EVM
Figure 18 CDMA’s QPSK Modulation With Various Filter
Slopes
Tables 5 and 6 show the new abbreviated test list that
would be needed to provide good coverage of these two
receivers using EVM. Note that the GSM test list was
reduced from 32 elements to 18 elements. The CDMA test
list went from 25 to 11 measurements. Keep in mind that this
did not reduce test coverage at all. In fact, EVM provides the
entire phase and amplitude response across the entire
operating band and not just three discrete points. It could
easily be argued that test coverage is significantly improved.
Table 5 GSM Test List with EVM
Conclusions
This test list breakdown has been done based off of
information found in the public domain. It is by no means
exhaustive. This is just a representation of one specific GSM
receiver and one specific CDMA receiver. Of course, there
will be many other details in the final test list not included
here.
It is very common for production test lists to contain many
redundant tests that do not improve system test coverage.
This paper is meant to demonstrate the kind of thinking
needed in order to come up with a comprehensive test list that
makes sense instead of “shot-gunning.” Keep in mind
differences between characterization and production. The
best of both worlds can be achieved without significantly
driving up cost of test.
Minimizing the number of tests will always be the greatest
throughput advantage. Take advantage of how the radio
blocks work and interact to get as much impact as possible
from each test. Keep in mind that the ability to perform EVM
measurements on cellular receivers will greatly enhance
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coverage and drive down cost of test. The ability to perform
these tests at final will ultimately determine cost of test.
References
1. “Testing and Troubleshooting Digital RF Communication
Receiver Designs”, Agilent Technologies Application
Note 1314 pp 3-6.
2. “Modern Solutions for Testing RF Communication
Amplifiers”, Hewlett Packard, 1995 RF and Microwave
Design Test Seminar. pp 35-40
3. “Measuring EDGE Signals-New and Modified Techniques
and Measurement Requirements” Agilent Technologies
Application Note 1361 pp 4-12
4. “Digital Modulation in Communication Systems- An
Introduction” Agilent Technologies Application Note
1298 pp 12-25
5 “Understanding GSM/EDGE Transmitter and Receiver
Measurments for Base Transceiver Stations and Their
Components” Agilent Technologies Application Note
1312 pp 30-33
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